LT1169 Dual Low Noise, Picoampere Bias Current, JFET Input Op Amp DESCRIPTIO U S

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1 FEATRES Input Bias Current, Warmed p: pa Max % Tested Low Voltage Noise: nv/ Hz Max S and N Package Standard Pinout Very Low Input Capacitance:.pF Voltage Gain:. Million Min Offset Voltage: mv Max Input Resistance: Ω Gain-Bandwidth Product:.MHz Typ Guaranteed Specifications with ±V Supplies Guaranteed Matching Specifications APPLICATI O S Photocurrent Amplifiers Hydrophone Amplifiers High Sensitivity Piezoelectric Accelerometers Low Voltage and Current Noise Instrumentation Amplifier Front Ends Two and Three Op Amp Instrumentation Amplifiers Active Filters, LTC and LT are registered trademarks of Linear Technology Corporation. LT9 Dual Low Noise, Picoampere Bias Current, JFET Input Op Amp DESCRIPTIO The LT9 achieves a new standard of excellence in noise performance for a dual JFET op amp. For the first time low voltage noise (nv/ Hz) is simultaneously offered with extremely low current noise (fa/ Hz), providing the lowest total noise for high impedance transducer applications. nlike most JFET op amps, the very low input bias current (pa Typ) is maintained over the entire common mode range which results in an extremely high input resistance ( Ω). When combined with a very low input capacitance (.pf) an extremely high input impedance results, making the LT9 the first choice for amplifying low level signals from high impedance transducers. The low input capacitance also assures high gain linearity when buffering AC signals from high impedance transducers. The LT9 is unconditionally stable for gains of or more, even with pf capacitive loads. Other key features are.mv V OS and a voltage gain over million. Each individual amplifier is % tested for voltage noise, slew rate (.V/µs), and gain-bandwidth product (.MHz). The LT9 is offered in the S and N packages. A full set of matching specifications are provided for precision instrumentation amplifier front ends. Specifications at ±V supply operation are also provided. For an even lower voltage noise please see the LT data sheet. TYPICAL APPLICATI HAMAMATS S-BK (9) -9 O Low Noise Light Sensor with DC Servo C D N9 V D N9 R k R k D N9 R k C pf V V R M / LT9 C.µF / LT9 RC > CR C D = PARASITIC PHOTODIODE CAPACITANCE V OT = mv/µwatt FOR nm WAVE LENGTH mv/µwatt FOR nm WAVE LENGTH R k V OT LT9 TA TOTAL khz VOLTAGE NOISE DENSITY (nv/ Hz) khz Output Voltage Noise Density vs Source Resistance k k R SORCE V N V N SORCE RESISTANCE ONLY V S = ±V k k k M M M G SORCE RESISTANCE (Ω) V N = (V OP AMP ) ktr S qi B R S LT9 TA

2 LT9 ABSOLTE AXI RATI GS W W W Supply Voltage C to C... ±V C to C... ±V Differential Input Voltage... ±V Input Voltage (Equal to Supply Voltage)... ±V Output Short-Circuit Duration... Indefinite Operating Temperature Range... C to C Storage Temperature Range... C to C Lead Temperature (Soldering, sec)... C PACKAGE/ORDER I FOR OT A IN A IN A V A N PACKAGE -LEAD PDIP TOP VIEW V OT B IN B B IN B S PACKAGE -LEAD PLASTIC SO T JMAX = C, θ JA = C/W (N) T JMAX = C, θ JA = 9 C/W (S) W Consult factory for Industrial and Military grade parts. ATIO ORDER PART NMBER LT9CN LT9CS S PART MARKING 9 ELECTRICAL CHARA CTERISTICS V S = ±V, V CM = V,, unless otherwise noted. SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS V OS Input Offset Voltage.. mv V S = ±V.. mv I OS Input Offset Current Warmed p (Note ). pa T J = C (Note ). pa I B Input Bias Current Warmed p (Note ). pa T J = C (Note ). pa e n Input Noise Voltage.Hz to Hz. µv P-P Input Noise Voltage Density f O = Hz nv/ Hz f O = Hz nv/ Hz i n Input Noise Current Density f O = Hz, f O = khz (Note ) fa/ Hz R IN Input Resistance Differential Mode Ω Common Mode V CM = V to V Ω C IN Input Capacitance. pf V S = ±V. pf V CM Input Voltage Range (Note ).. V.. V CMRR Common Mode Rejection Ratio V CM = V to V 9 db PSRR Power Supply Rejection Ratio V S = ±.V to ± V 9 db A VOL Large-Signal Voltage Gain V O = ±V, R L = k V/mV V O = ±V, R L = k V/mV V OT Output Voltage Swing R L = k ±. ±. V R L = k ±. ±. V SR Slew Rate R L k (Note ).. V/µs GBW Gain-Bandwidth Product f O = khz.. MHz Channel Separation f O = Hz, V O = ±V, R L = k db I S Supply Current per Amplifier.. ma V S = ±V.. ma V OS Offset Voltage Match.. mv I B Noninverting Bias Current Match Warmed p (Note ) pa CMRR Common Mode Rejection Match (Note ) 9 db PSRR Power Supply Rejection Match (Note ) 9 db

3 LT9 ELECTRICAL CHARA CTERISTICS V S = ±V, V CM = V, C T A C, (Note 9), unless otherwise noted. SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS V OS Input Offset Voltage.. mv V S = ± V.. mv V OS Average Input Offset Voltage Drift (Note ) µv/ C Temp I OS Input Offset Current pa I B Input Bias Current pa V CM Input Voltage Range.9. V.. V CMRR Common Mode Rejection Ratio V CM = V to.9v 9 9 db PSRR Power Supply Rejection Ratio V S = ±.V to ±V 9 db A VOL Large-Signal Voltage Gain V O = ±V, R L = k V/mV V O = ±V, R L = k V/mV V OT Output Voltage Swing R L = k ±. ±. V R L = k ±. ±. V SR Slew Rate R L k (Note ).9 V/µs GBW Gain-Bandwidth Product f O = khz. MHz I S Supply Current per Amplifier.. ma V S = ±V.. ma V OS Offset Voltage Match. mv I B Noninverting Bias Current Match. pa CMRR Common Mode Rejection Match (Note ) 9 db PSRR Power Supply Rejection Match (Note ) 9 db V S = ±V, V CM = V, C T A C, (Note ), unless otherwise noted. SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS V OS Input Offset Voltage.. mv V S = ±V.9. mv V OS Average Input Offset Voltage Drift µv/ C Temp I OS Input Offset Current pa I B Input Bias Current pa V CM Input Voltage Range.. V.. V CMRR Common Mode Rejection Ratio V CM = V to.v 9 db PSRR Power Supply Rejection Ratio V S = ±.V to ±V 9 9 db A VOL Large-Signal Voltage Gain V O = ±V, R L = k V/mV V O = ±V, R L = k V/mV V OT Output Voltage Swing R L = k ±. ±. V R L = k ±. ±. V SR Slew Rate R L k.. V/µs GBW Gain-Bandwidth Product f O = khz. MHz I S Supply Current per Amplifier.. ma V S = ±V.. ma

4 LT9 ELECTRICAL CHARA CTERISTICS V S = ±V, V CM = V, C T A C, (Note ), unless otherwise noted. SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS V OS Offset Voltage Match. mv I B Noninverting Bias Current Match pa CMRR Common Mode Rejection Match (Note ) 9 db PSRR Power Supply Rejection Match (Note ) 9 db The denotes specifications which apply over the full operating temperature range. Note : Typical parameters are defined as the % yield of parameter distributions of individual amplifiers, i.e., out of LT9s ( op amps) typically op amps will be better than the indicated specification. Note : I B and I OS readings are extrapolated to a warmed-up temperature from C measurements and C characterization data. Note : Current noise is calculated from the formula: i n = (qi B ) / where q =. 9 coulomb. The noise of source resistors up to M swamps the contribution of current noise. Note : Input voltage range functionality is assured by testing offset voltage at the input voltage range limits to a maximum of.mv. Note : This parameter is not % tested. Note : Slew rate is measured in A V = ; input signal is ±.V, output measured at ±.V. Note : The LT9 is designed, characterized and expected to meet these extended temperature limits, but is not tested at C and C. Guaranteed I grade parts are available; consult factory. Note : CMRR and PSRR are defined as follows: () CMRR and PSRR are measured in µv/v on the individual amplifiers. () The difference is calculated between the matching sides in µv/v. () The result is converted to db. Note 9: The LT9 is measured in an automated tester in less than one second after application of power. Depending on the package used, power dissipation, heat sinking, and air flow conditions, the fully warmed-up chip temperature can be C to C higher than the ambient temperature. TYPICAL PERFOR A W CE CHARA CTERISTICS VOLTAGE NOISE (µv/div).hz to Hz Voltage Noise PERCENT OF NITS (%) khz Input Noise Voltage Distribution V S = ±V OP AMPS TESTED RMS VOLTAGE NOISE (nv/ Hz) Voltage Noise vs Frequency /f CORNER Hz V S = ±V TYPICAL TIME (SEC) INPT VOLTAGE NOISE (nv/ Hz) k k FREQENCY (Hz) LT9 TPC LT9 TPC LT9 TPC

5 LT9 TYPICAL PERFOR A W CE CHARA CTERISTICS VOLTAGE NOISE (AT khz) (nv/ Hz) Voltage Noise vs Chip Temperature V S = ±V 9 INPT BIAS AND OFFSET CRRENTS (A) n n n n p p p p p p Input Bias and Offset Currents vs Chip Temperature V S = ±V V CM = TO V BIAS CRRENT OFFSET CRRENT INPT BIAS AND OFFSET CRRENTS (pa) Input Bias and Offset Currents Over the Common Mode Range V S = ±V BIAS CRRENT OFFSET CRRENT TEMPERATRE ( C).p TEMPERATRE ( C) COMMON MODE RANGE (V) COMMON MODE LIMIT (V) REFERRED TO POWER SPPLY V V. Common Mode Limit vs Temperature LT9 TPC V = V TO V V = V TO V TEMPERATRE ( C) LT9 TPC COMMON MODE REJECTION RATIO (db) LT9 TPC* Common Mode Rejection Ratio vs Frequency V S = ±V k k k M M FREQENCY (Hz) LT9 TPC POWER SPPLY REJECTION RATIO (db) Power Supply Rejection Ratio vs Frequency PSRR PSRR k k k FREQENCY (Hz) LT9 TPC M LT9 TPC9 M VOLTAGE GAIN (db). Voltage Gain vs Frequency V S = ±V k M M FREQENCY (Hz) LT9 TPC VOLTAGE GAIN (V/µV) 9 Voltage Gain vs Chip Temperature R L = k V S = ±V V O = ±V, R L = k V O = ±V, R L = k R L =k CHIP TEMPERATRE ( C) LT9 TPC VOLTAGE GAIN (db). Gain and Phase Shift vs Frequency V S = ±V C L = pf GAIN PHASE FREQENCY (MHz) LT9 TPC PHASE SHIFT (DEG)

LT9 TYPICAL PERFOR A W Small-Signal Transient Response CE CHARA CTERISTICS Large-Signal Transient Response Supply Current vs Supply Voltage mv/div A V = C L = pf V S = ±V, ±V µs/div LT9 TPC V/DIV A V

C V C. I SINK I OTPT CRRENT (ma) SORCE OVERSHOOT (%).

6 LT9 TYPICAL PERFOR A W Small-Signal Transient Response CE CHARA CTERISTICS Large-Signal Transient Response Supply Current vs Supply Voltage mv/div A V = C L = pf V S = ±V, ±V µs/div LT9 TPC V/DIV A V = C L = pf V S = ±V µs/div LT9 TPC SPPLY CRRENT PER AMPLIFIER (ma) C C C ± ± ± SPPLY VOLTAGE (V) ± LT9 TPC V OTPT VOLTAGE SWING (V) Output Voltage Swing vs Load Current C C V S = ±V TO ±V C C. C V C. I SINK I OTPT CRRENT (ma) SORCE OVERSHOOT (%). Capacitive Load Handling V S = ±V R L k V O = mv P-P A V =, R F = k, C F = pf A V = A V = CAPACITIVE LOAD (pf) SLEW RATE (V/µs) Slew Rate and Gain-Bandwidth Product vs Temperature SLEW RATE GAIN-BANDWIDTH TEMPERATRE ( C) GAIN-BANDWIDTH PRODCT (fo = khz)(mhz) LT9 TPC LT9 TPC LT9 TPC PERCENT OF NITS Distribution of Offset Voltage Drift with Temperature V S = ±V OP AMPS OFFSET VOLTAGE DRIFT WITH TEMPERATRE (µv/ C) CHANGE IN OFFSET VOLTAGE (µv) Warm-p Drift V S = ±V N PACKAGE TIME AFTER POWER ON (MIN) CHANNEL SEPARATION (db) Channel Separation vs Frequency LIMITED BY THERMAL INTERACTION V S = ±V R L = k V O = V P-P LIMITED BY PIN-TO-PIN CAPACITANCE k k k M M FREQENCY (Hz) LT9 TPC9 LT9 TPC LT9 TPC

7 LT9 TYPICAL PERFOR A W CE CHARA CTERISTICS TOTAL HARMONIC DISTORTION NOISE (%).... THD and Noise vs Frequency for Noninverting Gain Z L = k pf V O = V P-P A V =,, MEASREMENT BANDWIDTH = Hz TO khz A V = A V = A V = NOISE FLOOR k k k FREQENCY (Hz) LT9 TPC TOTAL HARMONIC DISTORTION NOISE (%).... THD and Noise vs Frequency for Inverting Gain Z L = k pf V O = V P-P A V =,, MEASREMENT BANDWIDTH = Hz TO khz A V = A V = NOISE FLOOR A V = k k k FREQENCY (Hz) LT9 TPC TOTAL HARMONIC DISTORTION NOISE (%)..... THD and Noise vs Output Amplitude for Inverting Gain Z L = k pf f O = khz A V =,, MEASREMENT BANDWIDTH = Hz TO khz A V = A V = A V = NOISE FLOOR OTPT SWING (V P-P ) LT9 TPC TOTAL HARMONIC DISTORTION NOISE (%)..... THD and Noise vs Output Amplitude for Noninverting Gain Z L = k pf f O = khz A V =,, MEASREMENT BANDWIDTH = Hz TO khz A V = A V = A V = NOISE FLOOR OTPT SWING (V P-P ) LT9 TPC INTERMODLATION DISTORTION (AT khz)(%)..... CCIF IMD Test (Equal Amplitude Tones at khz, khz)* V S = ±V R L = k A V = ±. OTPT SWING (V P-P ) LT9 TPC *SEE LT DATA SHEET FOR DEFINITION OF CCIF TESTING APPLICATI O S LT9 vs the Competition I FOR W ATIO With improved noise performance, the LT9 dual in the plastic DIP directly replaces such JFET op amps as the OPA, OPA, OP, and the AD. The combination of low current and voltage noise of the LT9 allows it to surpass most dual and single JFET op amps. The LT9 can replace many of the lowest noise bipolar amps that are used in amplifying low level signals from high impedance transducers. The best bipolar op amps will eventually lose out to the LT9 when transducer impedance increases due to higher current noise. The extremely high input impedance ( Ω) assures that the input bias current is almost constant over the entire common mode range. Figure shows how the LT9 stands up to the competition. nlike the competition, as the input voltage is swept across the entire common mode range the input bias current of the LT9 hardly changes. As a result the current noise does not degrade. This makes the LT9 the best choice in applications where an amplifier has to buffer signals from a high impedance transducer.

8 LT9 APPLICATI INPT BIAS CRRENT (pa) O S I FOR W ATIO Figure. Comparison of LT9, OP, and AD Input Bias Current vs Common Mode Range INPT NOISE VOLTAGE (nv/ H z) k k CRRENT NOISE = qi B Amplifying Signals from High Impedance Transducers The low voltage and current noise offered by the LT9 makes it useful in a wide range of applications, especially where high impedance, capacitive transducers are used such as hydrophones, precision accelerometers, and photodiodes. The total output noise in such a system is the gain times the RMS sum of the op amp s input referred voltage noise, the thermal noise of the transducer, and the op amp s input bias current noise times the transducer impedance. Figure shows total input voltage noise versus source resistance. In a low source resistance (<k) application the op amp voltage noise will dominate R S COMMON MODE RANGE (V) C S R S C S LT9 AD V O LT* LT9* LT9 OP LT9 LT LT RESISTOR NOISE ONLY k k k M M M G SORCE RESISTANCE (Ω) SORCE RESISTANCE = R S = R * PLS RESISTOR PLS RESISTOR pf CAPACITOR V n = A V V n (OP AMP) ktr qi B R LT9 F LT9 F Figure. Comparison of LT9 and LT Total Output khz Voltage Noise vs Source Resistance the total noise. This means the LT9 is superior to most dual JFET op amps. Only the lowest noise bipolar op amps have the advantage at low source resistances. As the source resistance increases from k to k, the LT9 will match the best bipolar op amps for noise performance, since the thermal noise of the transducer (ktr) begins to dominate the total noise. A further increase in source resistance, above k, is where the op amp s current noise component (qi B R ) will eventually dominate the total noise. At these high source resistances, the LT9 will out perform the lowest noise bipolar op amps due to the inherently low current noise of FET input op amps. Clearly, the LT9 will extend the range of high impedance transducers that can be used for high signalto-noise ratios. This makes the LT9 the best choice for high impedance, capacitive transducers. Optimization Techniques for Charge Amplifiers The high input impedance JFET front end makes the LT9 suitable in applications where very high charge sensitivity is required. Figure illustrates the LT9 in its inverting and noninverting modes of operation. A charge amplifier is shown in the inverting mode example; the gain depends on the principal of charge conservation at the input of the LT9. The charge across the transducer capacitance C S is transferred to the feedback capacitor C F resulting in a change in voltage dv, which is equal to dq/c F. The gain therefore is C F /C S. For unity-gain, the C F should equal the transducer capacitance plus the input capacitance of the LT9 and R F should equal R S. In the noninverting mode example, the transducer current is converted to a change in voltage by the transducer capacitance, C S. This voltage is then buffered by the LT9 with a gain of R/R. A DC path is provided by R S, which is either the transducer impedance or an external resistor. Since R S is usually several orders of magnitude greater than the parallel combination of R and R, R B is added to balance the DC offset caused by the noninverting input bias current and R S. The input bias currents, although small at room temperature, can create significant errors over increasing temperature, especially with transducer resistances of up to MΩ or more. The optimum value for R B is determined by equating the thermal noise (ktr S ) to the current noise (qi B ) times R S. Solving for R S results in R B = R S = V T /I B. A parallel

9 LT9 APPLICATI O S I FOR W ATIO R F R C F C B R B C S R S OTPT R OTPT TRANSDCER C B R B C B = C F C S R B = R F R S dq Q = CV; d t = I = C dv d t C S R S CB C S R B = R S R S > R OR R TRANSDCER LT9 F Input: ±. Sine Wave Figure. Inverting and Noninverting Gain Configurations LT9 Output OPA Output LT9 Fa LT9 Fb LT9 Fc Figure. Voltage Follower with Input Exceeding the Common Mode Range (V S = ±V) capacitor C B, is used to cancel the phase shift caused by the op amp input capacitance and R B. Reduced Power Supply Operation To take full advantage of a wide input common-mode range, the LT9 was designed to eliminate phase reversal. Referring to the photographs in Figure, the LT9 is shown operating in the follower mode (A V = ) at ±V supplies with the input swinging ±.V. The output of the LT9 clips cleanly and recovers with no phase reversal, unlike the competition as shown by the last photograph. This has the benefit of preventing lockup in servo systems and minimizing distortion components. The effect of input and output overdrive on one amplifier has no effect on the other, as each amplifier is biased independently. Advantages of Matched Dual Op Amps In many applications the performance of a system depends on the matching between two operational amplifiers rather than the individual characteristics of the two op amps. Two or three op amp instrumentation amplifiers, tracking voltage references and low drift active filters are some of the circuits requiring matching between two op amps. The well-known triple op amp configuration in Figure illustrates these concepts. Output offset is a function of the difference between the two halves of the LT9. This error cancellation principle holds for a considerable number of input referred parameters in addition to offset voltage and bias current. Input bias current will be the average of the two noninverting input currents (I B ). The difference between these two currents ( I B ) is the offset current of the instrumentation amplifier. Common-mode and power supply rejections will be dependent only on the match between the two amplifiers (assuming perfect resistor matching). The concepts of common mode and power supply rejection ratio match ( CMRR and PSRR) are best demonstrated with a numerical example: 9

10 LT9 APPLICATI IN V / LT9 IC V O S R k R Ω R k I FOR W R k R k / LT9 R IC IN k GAIN = BANDWIDTH =khz INPT REFERRED NOISE =.nv/ Hz AT khz WIDEBAND NOISE DC TO khz =.µv RMS C L.µF ATIO R k C pf LT9 F Assume CMRR A = µv/v or db, and CMRR B = 9µV/V or db, then CMRR = µv/v or 99dB; if CMRR B = 9µV/V which is still db, then CMRR = 9µV/V or db By specifying and guaranteeing all of these matching parameters, the LT9 can significantly improve the performance of matching-dependent circuits. Typical performance of the instrumentation amplifier: Input offset voltage =.mv Input bias current = pa Figure. Three Op Amp Instrumentation Amplifier / LT9 IC OTPT C L Input offset current = pa Input resistance = Ω Input noise =.µv P-P High Speed Operation The low noise performance of the LT9 was achieved by enlarging the input JFET differential pair to maximize the first stage gain. Enlarging the JFET geometry also increases the parasitic gate capacitance, which if left unchecked, can result in increased overshoot and ringing. When the feedback around the op amp is resistive (R F ), a pole will be created with R F, the source resistance and capacitance (R S,C S ), and the amplifier input capacitance (C IN =.pf). In closed-loop gain configurations with R S and R F in the MΩ range (Figure ), this pole can create excess phase shift and even oscillation. A small capacitor (C F ) in parallel with R F eliminates this problem. With R S (C S C IN ) = R F C F, the effect of the feedback pole is completely removed. C F R F R S C S C IN Figure LT9 F OTPT TYPICAL APPLICATIONS N nity-gain Buffer with Extended Load Capacitance Drive Capability V IN / LT9 R k C R Ω C L V OT LT9 TA C = C L.µF OTPT SHORT CIRCIT CRRENT ( ma) WILL LIMIT THE RATE AT WHICH THE VOLTAGE CAN CHANGE ACROSS LARGE CAPACITORS (I = C dv dt ) I I Light Balance Detection Circuit PD PD / LT9 LT9 TA V OT = M (I I ) PD, PD = HAMAMATS S-BK WHEN EQAL LIGHT ENTERS PHOTODIODES, V OT < mv. R M C pf TO pf VOT

11 LT9 TYPICAL APPLICATIONS N Low Noise Hydrophone Amplifier with DC Servo Accelerometer Amplifier with DC Servo C pf R* M R.9k V TO V R M R k R C* Ω HYDRO- PHONE C T R M R M / LT9 V TO V R k C.µF / LT9 DC OTPT.mV FOR T A < C OTPT VOLTAGE NOISE = nv/ Hz AT khz (GAIN = ) C C T pf TO pf; RC > RC T ; *OPTIONAL R M R M LT9 TA OTPT ACCELEROMETER B & K MODEL OR EQIVALENT () - R k C µf / LT9 R M V TO V C µf / LT9 OTPT RC = RC > R ( R/R) C OTPT =.mv/pc* =.mv/g** DC OTPT.9mV V TO V OTPT NOISE = nv/ H z AT khz *PICOCOLOMBS **g = EARTH S GRAVITATIONAL CONSTANT R M LT9 TA PACKAGE DESCRIPTIO.. (..) Dimensions in inches (millimeters) unless otherwise noted. N Package -Lead PDIP (Narrow.) (LTC DWG # --).. (..). ±. (. ±.).* (.) MAX.9. (.9.) ( ). (.) TYP. ±. (. ±.) *THESE DIMENSIONS DO NOT INCLDE MOLD FLASH OR PROTRSIONS. MOLD FLASH OR PROTRSIONS SHALL NOT EXCEED. INCH (.mm). (.) MIN. ±. (. ±.). (.) MIN. ±.* (. ±.) N 9.. (..).. (..) S Package -Lead Plastic Small Outline (Narrow.) (LTC DWG # --) TYP..9 (..).. (..).9.9* (..) (..) *DIMENSION DOES NOT INCLDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED." (.mm) PER SIDE ** DIMENSION DOES NOT INCLDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED." (.mm) PER SIDE. (.) TYP.. (.9.9) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights...** (..9) SO 99

12 LT9 TYPICAL APPLICATIONS N Hz Fourth Order Chebyshev Lowpass Filter (.db Ripple) V IN R k C nf R k R 9k V / LT9 V C nf R k R k R 9k C nf LT9 TA TYPICAL OFFSET.mV % TOLERANCES FOR V IN = V P-P, V OT = db AT f > Hz = db AT f =.Hz LOWER RESISTOR VALES WILL RESLT IN LOWER THERMAL NOISE AND LARGER CAPACITORS C nf / LT9 V OT Paralleling Amplifiers to Reduce Voltage Noise 9Ω A / LT9 k.k k 9Ω 9Ω A / LT9 V An / LT9 V k k.k.k V / LT9 V OTPT. ASSME VOLTAGE NOISE OF LT9 AND Ω SORCE RESISTOR =.nv/ H z. GAIN WITH n LT9s IN PARALLEL = n. OTPT NOISE = n.nv/ H z. INPT REFERRED NOISE = OTPT NOISE =. nv/ H z n n. NOISE CRRENT AT INPT INCREASES n TIMES. IF n =, GAIN =, BANDWIDTH = khz, RMS NOISE, DC TO MHz =.µv =.µv LT9 TA RELATED PARTS PART NMBER DESCRIPTION COMMENTS LT Lowest Noise Dual JFET Op Amp.nV/ Hz Voltage Noise LT Micro Power Dual JFET Op Amp.pA I B, µa I SPPLY LT Low Power Dual JFET Op Amp.pA (Max) Input Bias Current Linear Technology Corporation McCarthy Blvd., Milpitas, CA 9- () -9 FAX: () - TELEX: fa LT/TP 9 REV A K PRINTED IN SA LINEAR TECHNOLOGY CORPORATION 99

13 Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Analog Devices Inc.: LT9CN LT9CS#TR LT9CN#PBF LT9CS#TRPBF LT9CS#PBF LT9CS

DESCRIPTIO TYPICAL APPLICATIO LT1113 Dual Low Noise, Precision, JFET Input Op Amp FEATURES APPLICATIO S

DESCRIPTIO TYPICAL APPLICATIO LT1113 Dual Low Noise, Precision, JFET Input Op Amp FEATURES APPLICATIO S LT Dual Low Noise, Precision, JFET Input Op Amp FEATRES % Tested Low Voltage Noise: nv/ Hz Max SO- Package Standard Pinout Voltage Gain:. Million Min Offset Voltage:.mV Max Offset Voltage Drift: µv/ C