LT1920 Single Resistor Gain Programmable, Precision Instrumentation Amplifier DESCRIPTIO FEATURES APPLICATIO S TYPICAL APPLICATIO

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1 FEATRES Single Gain Set Resistor: G = to, Gain Error: G =,.% Max Gain Nonlinearity: G =, ppm Max Input Offset Voltage: G =, µv Max Input Offset Voltage Drift: µv/ C Max Input Bias Current: na Max PSRR at G = : db Min CMRR at G = : 7dB Min Supply Current:.mA Max Wide Supply Range: ±.V to ±V khz Voltage Noise: 7.nV/ Hz.Hz to Hz Noise:.µV P-P Available in -Pin PDIP and SO Packages Meets IEC -- Level ESD Tests with Two External k Resistors APPLICATIO S Bridge Amplifiers Strain Gauge Amplifiers Thermocouple Amplifiers Differential to Single-Ended Converters Medical Instrumentation LT9 Single Resistor Gain Programmable, Precision Instrumentation Amplifier DESCRIPTIO The LT 9 is a low power, precision instrumentation amplifier that requires only one external resistor to set gains of to,. The low voltage noise of 7.nV/ Hz (at khz) is not compromised by low power dissipation (.9mA typical for ±.V to ±V supplies). The high accuracy of ppm maximum nonlinearity and.% max gain error (G = ) is not degraded even for load resistors as low as k (previous monolithic instrumentation amps used k for their nonlinearity specifications). The LT9 is laser trimmed for very low input offset voltage (µv max), drift (µv/ C), high CMRR (7dB, G = ) and PSRR (db, G = ). Low input bias currents of na max are achieved with the use of superbeta processing. The output can handle capacitive loads up to pf in any gain configuration while the inputs are ESD protected up to kv (human body). The LT9 with two external k resistors passes the IEC -- level specification. The LT9, offered in -pin PDIP and SO packages, is a pin for pin and spec for spec improved replacement for the AD. The LT9 is the most cost effective solution for precision instrumentation amplifier applications. For even better guaranteed performance, see the LT7., LTC and LT are registered trademarks of Linear Technology Corporation. TYPICAL APPLICATIO Single Supply Barometer R 9k LTCCZ-. R k R k R k / LT9 V S R k / LT9 7 R7 k LCAS NOVA SENOR NPC---A-L k k R SET k k R Ω R Ω V S = V TO V V S LT9 G = 7 9 TA VOLTS... TO -DIGIT DVM INCHES Hg... NONLINEARITY (ppm/div) Gain Nonlinearity OTPT VOLTAGE (V/DIV) G = R L = k V OT = ±V 7 TA

2 LT9 ABSOLTE MAXIMM RATINGS W W W (Note ) Supply Voltage... ±V Differential Input Voltage (Within the Supply Voltage)... ±V Input Voltage (Equal to Supply Voltage)... ±V Input Current (Note )... ±ma Output Short-Circuit Duration... Indefinite Operating Temperature Range... C to C Specified Temperature Range LT9C (Note )... C to 7 C LT9I... C to C Storage Temperature Range... C to C Lead Temperature (Soldering, sec)... C PACKAGE/ORDER INFORMATION R G IN IN V S TOP VIEW N PACKAGE -LEAD PDIP S PACKAGE -LEAD PLASTIC SO R G 7 V S OTPT REF T JMAX = C, θ JA = C/ W (N) T JMAX = C, θ JA = 9 C/ W (S) Consult factory for Military grade parts. W ORDER PART NMBER LT9CN LT9CS LT9IN LT9IS S PART MARKING 9 9I ELECTRICAL CHARACTERISTICS, V CM = V,, R L = k, unless otherwise noted. SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS G Gain Range G = (9.k/R G ) k Gain Error G =.. % G = (Note ).. % G = (Note ).. % G = (Note ).. % G/T Gain vs Temperature G < (Note ) ppm/ C Gain Nonlinearity (Note ) V O = ±V, G = ppm V O = ±V, G = and ppm V O = ±V, G = and ppm V OST Total Input Referred Offset Voltage V OST = V OSI V OSO /G V OSI Input Offset Voltage G =, V S = ±V to ±V µv G =, V S = ±V to ±V µv V OSI /T Input Offset Drift (RTI) (Note ) µv/ C V OSO Output Offset Voltage G =, V S = ±V to ±V µv G =, V S = ±V to ±V µv V OSO /T Output Offset Drift (Note ) µv/ C I OS Input Offset Current. na I B Input Bias Current. na e n Input Noise Voltage, RTI.Hz to Hz, G =. µv P-P.Hz to Hz, G =. µv P-P.Hz to Hz, G = and. µv P-P Total RTI Noise = e ni (e no /G) e ni Input Noise Voltage Density, RTI f O = khz 7. nv/ Hz e no Output Noise Voltage Density, RTI f O = khz 7 nv/ Hz i n Input Noise Current f O =.Hz to Hz pa P-P Input Noise Current Density f O = Hz fa/ Hz R IN Input Resistance V IN = ±V GΩ C IN(DIFF) Differential Input Capacitance f O = khz. pf

3 ELECTRICAL CHARACTERISTICS, V CM = V,, R L = k, unless otherwise noted. LT9 SYMBOL PARAMETER CONDITIONS (Note ) MIN TYP MAX NITS C IN(CM) Common Mode Input Capacitance f O = khz. pf V CM Input Voltage Range G =, Other Input Grounded V S = ±.V to ±V V S.9 V S. V V S = ±V to ±V V S.9 V S. V V S = ±.V to ±V V S. V S. V V S = ±V to ±V V S. V S. V CMRR Common Mode Rejection Ratio k Source Imbalance, V CM = V to ±V G = 7 9 db G = 9 db G = db G = db PSRR Power Supply Rejection Ratio V S = ±. to ±V G = db G = db G = db G = db I S Supply Current V S = ±.V to ±V.9. ma V OT Output Voltage Swing R L = k V S = ±.V to ±V V S. V S. V V S = ±V to ±V V S. V S. V V S = ±.V to ±V V S. V S. V V S = ±V to ±V V S. V S. V I OT Output Current 7 ma BW Bandwidth G = khz G = khz G = khz G = khz SR Slew Rate G =, V OT = ±V. V/µs Settling Time to.% V Step G = to µs G = µs R REFIN Reference Input Resistance kω I REFIN Reference Input Current V REF = V µa V REF Reference Voltage Range V S. V S. V A VREF Reference Gain to Output ±. The denotes specifications that apply over the full specified temperature range. Note : Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note : Does not include the effect of the external gain resistor R G. Note : This parameter is not % tested. Note : The LT9C is designed, characterized and expected to meet the industrial temperature limits, but is not tested at C and C. I-grade parts are guaranteed. Note : This parameter is measured in a high speed automatic tester that does not measure the thermal effects with longer time constants. The magnitude of these thermal effects are dependent on the package used, heat sinking and air flow conditions. Note : Typical parameters are defined as the % of the yield parameter distribution.

LT9 TYPICAL PERFOR A CE CHARACTERISTICS W Gain Nonlinearity, G = Gain Nonlinearity, G = Gain Nonlinearity, G = NONLINEARITY (ppm/div) NONLINEARITY (ppm/div) NONLINEARITY (ppm/div) G = OTPT VOLTAGE

4 LT9 TYPICAL PERFOR A CE CHARACTERISTICS W Gain Nonlinearity, G = Gain Nonlinearity, G = Gain Nonlinearity, G = NONLINEARITY (ppm/div) NONLINEARITY (ppm/div) NONLINEARITY (ppm/div) G = OTPT VOLTAGE (V/DIV) R L = k V OT = ±V 7 G G = OTPT VOLTAGE (V/DIV) R L = k V OT = ± V 7 G G = OTPT VOLTAGE (V/DIV) R L = k V OT = ±V 7 G NONLINEARITY (ppm/div) Gain Nonlinearity, G = G = OTPT VOLTAGE (V/DIV) R L = k V OT = ±V 7 G GAIN ERROR (%) Gain Error vs Temperature V OT = ±V R L = k *DOES NOT INCLDE TEMPERATRE EFFECTS OF R G G = G = * G = * G = * 7 TEMPERATRE ( C) CHANGE IN OFFSET VOLTAGE (µv) Warm-p Drift G = S N TIME AFTER POWER ON (MINTES) 9 G 9 G9 INPT BIAS CRRENT (pa) Input Bias Current vs Common Mode Input Voltage C C C 7 C C 9 9 COMMON MODE INPT VOLTAGE (V) 9 G COMMON MODE REJECTION RATIO (db) Common Mode Rejection Ratio vs Frequency. G = G = G = G = k SORCE IMBALANCE k k FREQENCY (Hz) k 9 G NEGATIVE POWER SPPLY REJECTION RATIO (db) Negative Power Supply Rejection Ratio vs Frequency. G = G = G = V = V G = k k FREQENCY (Hz) k 9 G

5 LT9 TYPICAL PERFOR A CE CHARACTERISTICS W POSITIVE POWER SPPLY REJECTION RATIO (db) VOLTAGE NOISE DENSITY (nv Hz) Positive Power Supply Rejection Ratio vs Frequency. G = G = G = V = V G = k k FREQENCY (Hz) Voltage Noise Density vs Frequency /f CORNER = Hz /f CORNER = 9Hz GAIN = GAIN = k 9 G /f CORNER = 7Hz GAIN =, BW LIMIT GAIN = k k k FREQENCY (Hz) 9 G9 GAIN (db) Gain vs Frequency G = G = G = G =.. FREQENCY (khz) NOISE VOLTAGE (µv/div).hz to Hz Noise Voltage, G = 9 G7 7 9 TIME (SEC) 9 G SPPLY CRRENT (ma) NOISE VOLTAGE (.µv/div) Supply Current vs Supply Voltage SPPLY VOLTAGE (±V) C C C 9 G.Hz to Hz Noise Voltage, RTI G = 7 9 TIME (SEC) 9 G CRRENT NOISE DENSITY (fa/ Hz) Current Noise Density vs Frequency R S FREQENCY (Hz) 9 G CRRENT NOISE (pa/div).hz to Hz Current Noise 7 9 TIME (SEC) 9 G OTPT CRRENT (ma) (SINK) (SORCE) Short-Circuit Current vs Time T A = C T A = C T A = C T A = C TIME FROM OTPT SHORT TO GROND (MINTES) 9 G

LT9 TYPICAL PERFOR A CE CHARACTERISTICS W V/DIV Large-Signal Transient Response G = R L = k C L = pf µs/div 7 G mv/div Small-Signal Transient Response G = R L = k C L = pf µs/div 7 G9

Frequency V/DIV mv/div OTPT IMPEDANCE (Ω) G = TO G = V S = ± V R L = k C L = pf µs/div 7 G G = R L = k C L = pf µs/div 7 G.

6 LT9 TYPICAL PERFOR A CE CHARACTERISTICS W V/DIV Large-Signal Transient Response G = R L = k C L = pf µs/div 7 G mv/div Small-Signal Transient Response G = R L = k C L = pf µs/div 7 G9 OVERSHOOT (%) 9 7 Overshoot vs Capacitive Load V OT = ±mv R L = A V = A V = A V CAPACITIVE LOAD (pf) 9 G Large-Signal Transient Response Small-Signal Transient Response Output Impedance vs Frequency V/DIV mv/div OTPT IMPEDANCE (Ω) G = TO G = V S = ± V R L = k C L = pf µs/div 7 G G = R L = k C L = pf µs/div 7 G. FREQENCY (khz) 9 G V/DIV Large-Signal Transient Response G = V S = ± V R L = k C L = pf µs/div 7 G mv/div Small-Signal Transient Response G = R L = k C L = pf µs/div 7 G PEAK-TO-PEAK OTPT SWING (V) ndistorted Output Swing vs Frequency G =,, G = FREQENCY (khz) 9 G7

= C L = pf R L = k TO.% V V TO.% 7 9 SETTLING TIME (µs) TO.% V OT V OT TO.% 9 G SLEW RATE (V/µs)..... Slew Rate vs Temperature V OT = ±V G =.

7 LT9 TYPICAL PERFOR A CE CHARACTERISTICS W Large-Signal Transient Response Small-Signal Transient Response Settling Time vs Gain V OT = V mv =.% V/DIV mv/div SETTLING TIME (µs) G = R L = k C L = pf µs/div 7 G7 G = R L = k C L = pf µs/div 7 G GAIN (db) 9 G OTPT STEP (V) Settling Time vs Step Size V S = ± G = C L = pf R L = k TO.% V V TO.% 7 9 SETTLING TIME (µs) TO.% V OT V OT TO.% 9 G SLEW RATE (V/µs)..... Slew Rate vs Temperature V OT = ±V G =. SLEW SLEW 7 TEMPERATRE ( C) 9 G Output Voltage Swing vs Load Current OTPT VOLTAGE SWING (V) (REFERRED TO SPPLY VOLTAGE) V S V S. V S. V S. V S. V S. V S. V S. V S. SORCE SINK C C C V S.. OTPT CRRENT (ma) 9 G9 7

8 LT9 BLOCK DIAGRAM W IN V R Ω V Q VB A R.7k C R k A R k OTPT R G R G IN V R Ω V VB A C R7 k R k Q V R.7k V REF 7 V V PREAMP STAGE Figure. Block Diagram DIFFERENCE AMPLIFIER STAGE 9 F THEORY OF OPERATIO The LT9 is a modified version of the three op amp instrumentation amplifier. Laser trimming and monolithic construction allow tight matching and tracking of circuit parameters over the specified temperature range. Refer to the block diagram (Figure ) to understand the following circuit description. The collector currents in Q and Q are trimmed to minimize offset voltage drift, thus assuring a high level of performance. R and R are trimmed to an absolute value of.7k to assure that the gain can be set accurately (.% at G = ) with only one external resistor R G. The value of R G in parallel with R (R) determines the transconductance of the preamp stage. As R G is reduced for larger programmed gains, the transconductance of the input preamp stage increases to that of the input transistors Q and Q. This increases the open-loop gain when the programmed gain is increased, reducing the input referred gain related errors and noise. The input voltage noise at gains greater than is determined only by Q and Q. At lower gains the noise of the difference amplifier and preamp gain setting resistors increase the noise. The gain bandwidth product is determined by C, C and the preamp transconductance which increases with programmed gain. Therefore, the bandwidth does not drop proportional to gain. The input transistors Q and Q offer excellent matching, which is inherent in NPN bipolar transistors, as well as picoampere input bias current due to superbeta processing. The collector currents in Q and Q are held constant due to the feedback through the Q-A-R loop and Q-A-R loop which in turn impresses the differential input voltage across the external gain set resistor R G. Since the current that flows through R G also flows through R and R, the ratios provide a gained-up differential voltage, G = (R R)/R G, to the unity-gain difference amplifier A. The common mode voltage is removed by A, resulting in a single-ended output voltage referenced to the voltage on the REF pin. The resulting gain equation is: V OT V REF = G(V IN V IN ) where: G = (9.kΩ /R G ) solving for the gain set resistor gives: R G = 9.kΩ/(G )

9 THEORY OF OPERATIO LT9 Input and Output Offset Voltage The offset voltage of the LT9 has two components: the output offset and the input offset. The total offset voltage referred to the input (RTI) is found by dividing the output offset by the programmed gain (G) and adding it to the input offset. At high gains the input offset voltage dominates, whereas at low gains the output offset voltage dominates. The total offset voltage is: Total input offset voltage (RTI) = input offset (output offset/g) Total output offset voltage (RTO) = (input offset G) output offset Reference Terminal The reference terminal is one end of one of the four k resistors around the difference amplifier. The output voltage of the LT9 (Pin ) is referenced to the voltage on the reference terminal (Pin ). Resistance in series with the REF pin must be minimized for best common mode rejection. For example, a Ω resistance from the REF pin to ground will not only increase the gain error by.% but will lower the CMRR to db. Output Offset Trimming The LT9 is laser trimmed for low offset voltage so that no external offset trimming is required for most applications. In the event that the offset needs to be adjusted, the circuit in Figure is an example of an optional offset adjust circuit. The op amp buffer provides a low impedance to the REF pin where resistance must be kept to minimum for best CMRR and lowest gain error. IN IN R G LT9 REF ±mv ADJSTMENT RANGE OTPT / LT k V V mv Ω Ω mv Figure. Optional Trimming of Output Offset Voltage 9 F Single Supply Operation For single supply operation, the REF pin can be at the same potential as the negative supply (Pin ) provided the output of the instrumentation amplifier remains inside the specified operating range and that one of the inputs is at least.v above ground. The barometer application on the front page of this data sheet is an example that satisfies these conditions. The resistance R SET from the bridge transducer to ground sets the operating current for the bridge and also has the effect of raising the input common mode voltage. The output of the LT9 is always inside the specified range since the barometric pressure rarely goes low enough to cause the output to rail (. inches of Hg corresponds to.v). For applications that require the output to swing at or below the REF potential, the voltage on the REF pin can be level shifted. An op amp is used to buffer the voltage on the REF pin since a parasitic series resistance will degrade the CMRR. The application in the back of this data sheet, Four Digit Pressure Sensor, is an example. Input Bias Current Return Path The low input bias current of the LT9 (na) and the high input impedance (GΩ) allow the use of high impedance sources without introducing additional offset voltage errors, even when the full common mode range is required. However, a path must be provided for the input bias currents of both inputs when a purely differential signal is being amplified. Without this path the inputs will float to either rail and exceed the input common mode range of the LT9, resulting in a saturated input stage. Figure shows three examples of an input bias current path. The first example is of a purely differential signal source with a kω input current path to ground. Since the impedance of the signal source is low, only one resistor is needed. Two matching resistors are needed for higher impedance signal sources as shown in the second example. Balancing the input impedance improves both common mode rejection and DC offset. The need for input resistors is eliminated if a center tap is present as shown in the third example. 9

10 THEORY OF OPERATIO LT9 THERMOCOPLE MICROPHONE, R G LT9 HYDROPHONE, R G LT9 R G ETC LT9 k k k CENTER-TAP PROVIDES BIAS CRRENT RETRN 9 F Figure. Providing an Input Common Mode Current Path APPLICATIONS INFORMATION W The LT9 is a low power precision instrumentation amplifier that requires only one external resistor to accurately set the gain anywhere from to. The output can handle capacitive loads up to pf in any gain configuration and the inputs are protected against ESD strikes up to kv (human body). Input Protection The LT9 can safely handle up to ±ma of input current in an overload condition. Adding an external k input resistor in series with each input allows DC input fault voltages up to ±V and improves the ESD immunity to kv (contact) and kv (air discharge), which is the IEC -- level specification. If lower value input resistors are needed, a clamp diode from the positive supply to each input will maintain the IEC -- specification to level for both air and contact discharge. A N9 drain/source to gate is a good low leakage diode for use with k resistors, see Figure. The input resistors should be carbon and not metal film or carbon film. RFI Reduction In many industrial and data acquisition applications, instrumentation amplifiers are used to accurately amplify small signals in the presence of large common mode voltages or high levels of noise. Typically, the sources of these very small signals (on the order of microvolts or millivolts) are sensors that can be a significant distance from the signal conditioning circuit. Although these sen- R IN V CC J N9 V CC J N9 R G R IN OPTIONAL FOR HIGHEST ESD PROTECTION V CC V EE Figure. Input Protection LT9 REF OT 9 F sors may be connected to signal conditioning circuitry, using shielded or unshielded twisted-pair cabling, the cabling may act as antennae, conveying very high frequency interference directly into the input stage of the LT9. The amplitude and frequency of the interference can have an adverse effect on an instrumentation amplifier s input stage by causing an unwanted DC shift in the amplifier s input offset voltage. This well known effect is called RFI rectification and is produced when out-of-band interference is coupled (inductively, capacitively or via radiation) and rectified by the instrumentation amplifier s input transistors. These transistors act as high frequency signal detectors, in the same way diodes were used as RF envelope detectors in early radio designs. Regardless of the type of interference or the method by which it is coupled into the circuit, an out-of-band error signal appears in series with the instrumentation amplifier s inputs.

11 LT9 APPLICATIONS INFORMATION W To significantly reduce the effect of these out-of-band signals on the input offset voltage of instrumentation amplifiers, simple lowpass filters can be used at the inputs. This filter should be located very close to the input pins of the circuit. An effective filter configuration is illustrated in Figure, where three capacitors have been added to the inputs of the LT9. Capacitors C XCM and C XCM form lowpass filters with the external series resistors R S, to any out-of-band signal appearing on each of the input traces. Capacitor C XD forms a filter to reduce any unwanted signal that would appear across the input traces. An added benefit to using C XD is that the circuit s AC common mode rejection is not degraded due to common mode capacitive imbalance. The differential mode and common mode time constants associated with the capacitors are: t DM(LPF) = ()(R S )(C XD ) t CM(LPF) = (R S, )(C XCM, ) Setting the time constants requires a knowledge of the frequency, or frequencies of the interference. Once this frequency is known, the common mode time constants can be set followed by the differential mode time constant. Set the common mode time constants such that they do not degrade the LT9 s inherent AC CMR. Then the differential mode time constant can be set for the bandwidth required for the application. Setting the differential mode time constant close to the sensor s BW also minimizes any noise pickup along the leads. To avoid any possibility of inadvertently affecting the signal to be processed, set the common mode time constant an order of magnitude (or more) larger than the differential mode time constant. To avoid any possibility of common mode to differential mode signal conversion, match the common mode time constants to % or better. If the sensor is an RTD or a resistive strain gauge, then the series resistors R S, can be omitted, if the sensor is in proximity to the instrumentation amplifier. IN IN R S.k R S.k C XCM.µF C XD.µF C XCM.µF EXTERNAL RFI FILTER f(db) Hz Figure. Adding a Simple RC Filter at the Inputs to an Instrumentation Amplifier is Effective in Reducing Rectification of High Frequency Out-of-Band Signals R G V LT9 V V OT 9 F PACKAGE DESCRIPTION.. (7..) Dimensions in inches (millimeters) unless otherwise noted... (..) N Package -Lead PDIP (Narrow.) (LTC DWG # --). ±. (. ±.7).* (.) MAX 7.9. (.9.) ( ). (.) TYP. ±. (. ±.). (.7) MIN. ±. (.7 ±.7). (.) MIN. ±.* (.77 ±.) *THESE DIMENSIONS DO NOT INCLDE MOLD FLASH OR PROTRSIONS. MOLD FLASH OR PROTRSIONS SHALL NOT EXCEED. INCH (.mm) Information furnished by Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. N 97

12 LT9 TYPICAL APPLICATION Nerve Impulse Amplifier IN PATIENT GROND IN PATIENT/CIRCIT PROTECTION/ISOLATION C.µF R M R k / LT R k R k R G k A V = POLE AT khz V V 7 LT9 G =.Hz HIGHPASS C V.7µF R M R Ω / LT V C nf 7 R7 k OTPT V/mV 9 TA PACKAGE DESCRIPTION.. (..).. (..) TYP Dimensions in inches (millimeters) unless otherwise noted. S Package -Lead Plastic Small Outline (Narrow.) (LTC DWG # --)..9 (..7).. (..).9.97* (..) RELATED PARTS..9 (..) *DIMENSION DOES NOT INCLDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED." (.mm) PER SIDE ** DIMENSION DOES NOT INCLDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED." (.mm) PER SIDE. (.7) TYP.. (.79.97)..7** (..9) SO 99 PART NMBER DESCRIPTION COMMENTS LTC Precision Chopper-Stabilized Instrumentation Amplifier Best DC Accuracy LT Precision, Micropower, Single Supply Instrumentation Amplifier Fixed Gain of or, I S < µa LT High Speed, JFET Instrumentation Amplifier Fixed Gain of or, V/µs Slew Rate LT7 Single Resistor Gain Programmable Precision pgraded Version of the LT9 Instrumentation Amplifier LTC -Bit, Low Power, ksps ADC with Serial and Parallel I/O Single Supply V or ±V Operation, ±.LSB INL and ±LSB DNL Max LT Precision Series Reference Micropower;.V, V, V Versions; High Precision LTC Active RC Filter Lowpass, Bandpass, Highpass Responses; Low Noise, Low Distortion, Four nd Order Filter Sections LTC -Bit, ksps, Sampling ADC Single V Supply, Bipolar Input Range: ±V, Power Dissipation: mw Typ Linear Technology Corporation McCarthy Blvd., Milpitas, CA 9-77 ()-9 FAX: () f LT/TP 99 K PRINTED IN SA LINEAR TECHNOLOGY CORPORATION 99

APPLICATIO S. LT1168 Low Power, Single Resistor Gain Programmable, Precision Instrumentation Amplifier DESCRIPTIO FEATURES TYPICAL APPLICATIO

APPLICATIO S. LT1168 Low Power, Single Resistor Gain Programmable, Precision Instrumentation Amplifier DESCRIPTIO FEATURES TYPICAL APPLICATIO FEATURES Supply Current: µa Max Meets IEC -- Level (±kv) ESD Tests with Two External k Resistors Single Gain Set Resistor: G = to, Gain Error: G =,.% Max Input Offset Voltage Drift:.µV/ C Max Gain Nonlinearity: