ECEN620: Network Theory Broadband Circuit Design Fall 2018

Transcription

1 EEN60: Network heory Broadband ircuit esign all 08 Lecture 3: ransimpedance mplifiers (Is) Sam Palermo nalog & Mixed-Signal enter exas &M University

2 nnouncements Project descriptions are posted on the website Preliminary report due /0 and will count as the final homework

3 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 3

4 Optical eceiver echnology Photodetectors convert optical power into current p-i-n photodiodes Waveguide Ge photodetectors Electrical amplifiers then convert the photocurrent into a voltage signal ransimpedance amplifiers Limiting amplifiers Integrating optical receiver I Output L Output 4

5 ransimpedance mplifier (I) ransimpedance Key design objectives v i lso expressed in units of db by 0log Z o i Z High transimpedance gain Low input resistance for high bandwidth and efficient gain or large input currents, the I gain can compress and pulse-width distortion/jitter can result 5

6 Maximum urrents Input Overload urrent he maximum peak-to-peak input current for which we can achieve the desired BE ssuming high extinction ratio Maximum Input urrent for Linear Operation Often quantified by the current level for a certain gain compression (db) pp lin i i pp ovl i P pp ovl ovl 6

7 esistive ront-end irect trade-offs between transimpedance, bandwidth, and noise performance 7 L in p db L in BW 3 L rms in n L L out n in n L n out n K I k V I k df f j k df Z I V,,,, 0 0, 4 in L out L n,out n, [azavi]

8 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 8

9 ommon-gate I [azavi] out i m i o mb i in in in r o gm gmb ro gm Input resistance (input bandwidth) and transimpedance are decoupled 9

10 ommon-gate I requency esponse [azavi] out B out in in B Neglecting transistor r o : v i out in s g m in g mb s out Often the input pole may dominate due to large photodiode capacitance (00 500f) 0

11 ommon-gate I Noise [azavi] V n, out I n, M I I n, in Neglecting transistor r n, 4k 3 4k g m 3 g m o : Hz V Hz Both the bias current source and contribute to the input noise current can be increased to reduce noise, but voltage headroom can limit this ommon-gate Is are generally not for low-noise applications However, they are relatively simple to design with high stability

12 egulated ascode (G) I Input transistor gm is boosted by commonsource amplifier gain, resulting in reduced input resistance [Park ESSI 000] equires additional voltage headroom Increased input-referred noise from the commonsource stage

13 MOS 0GHz I n additional commongate stage in the feedback provides further gm-boosting and even lower input resistance Shunt-peaking inductors provide bandwidth extension at zero power cost, but very large area cost [Kromer JSS 004] gm 3 gm33 3

14 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 4

15 eedback I w/ Ideal mplifier Input bandwidth is extended by the factor + ransimpedance is approximately an make large without worrying about voltage headroom considerations 5 I in p in p s s Z With Infinite Bandwidth mplifier:

16 eedback I w/ inite Bandwidth mplifier 6 Q s Q s s Z s s s in o o o : mplifier With inite Bandwidth inite bandwidth amplifier modifies the transimpedance transfer function to a secondorder low-pass function

17 eedback I w/ inite Bandwidth mplifier Non-zero amplifier time constant can actually increase I bandwidth!! [Sackinger] However, can result in peaking in frequency domain and overshoot/ringing in time domain i in I -(s) v out Often either a Butterworth (Q=/sqrt()) or Bessel response (Q=/sqrt(3)) is used Butterworth gives maximally flat frequency response Bessel gives maximally flat groupdelay Butterworth Bessel 7

18 eedback I ransimpedance Limit If we assume a Butterworth response for mazimally flat frequency response : Q or a Butterworth response : 3dB 0 times larger than 0 case of Plugging into above expression yields the maximum possible Maximum Maximum proportional to amp gain-bandwidth product If amp GBW is limited by technology f, then in order to increase bandwidth, must decrease quadratically! 3dB 3dB for a given bandwidth [Mohan JSS 000] 8

19 eedback I 9 m m out m in m m m g g g g g g gain of that the source follower has an ideal ssuming in out s power supply voltages drop, there is not much headroom left for and the amplifier gain degrades

20 MOS Inverter-Based eedback I [Ingels JSS 994] MOS inverter-based Is allow for reduced voltage headroom operation Multiple inverter stages in feedback provide higher gain at the cost of reduced stability iode-connected transistor loads allow for high-frequency internal poles 0

21 Input-eferred Noise urrent p Hz I noise is modeled with an input-referred noise current source that reproduces the output I output noise when passed through an ideal noiseless I his noise source will depend on the source impedance, which is determined mostly by the photodetector capacitance

22 Input-eferred Noise urrent Spectrum p Hz Input-referred noise current spectrum typically consists of uniform, high-frequency f, & lowfrequency /f components o compare Is, we need to see this noise graph out to ~X the I bandwidth ecall the noise bandwidth tables

23 Input-eferred MS Noise urrent he input-referred rms noise current can be calculated by dividing the rms output noise voltage by the I s midband transimpedance value i rms n, I BW 0 Z f I n, I If we integrate the output noise, the upper bound isn t too critical. Often this is infinity for derivations, or X the I bandwidth in simulation his rms current sets the I s electrical sensitivity pp rms i sens Qin, I o determine the total optical receiver sensitivity, we need to consider the detector noise and responsivity f df 3

24 veraged Input-eferred Noise urrent ensity I noise performance can also be quantified by the averaged input-referred noise current density i avg n, I i rms n, I BW 3dB his quantity has units of p Hz. Note, this is different than averaging the input - referred noise spectrum, I n, I f over the I bandwidth. 4

25 E eedback I Input-eferred Noise urrent Spectrum he feedback resistor and amplifier front-end noise components determine the input-referred noise current spectrum I f I f I f n, I n, res n, front he feedback resistor component is uniform with frequency 4k I, f n res 5

26 E eedback I Input-eferred Noise urrent Spectrum Gate current-induced shot noise his is I n, G qi G typically small for MOS designs E channel noise I n, 4kg m is the channel noise factor, typically depending on the process. 6

27 Input-eferring the E hannel Noise 7 n out n n I Z i v i i,,, we could calculate o do this,,,,,,,,,,, noise is - referred E channel the input transfer function, high - pass Using this the detector and amplifier input capacitance. the summation of, where to ground the output and calculate But it is easier (and equivalent) f g k g k g k g f f I g s i i i s g s i g v g i i i m m m m front n m n I n I n I m n I m n I m n n I n Uniform and f component!

28 otal Input-eferred E eedback I Noise I 4k n, If qig 4k 4k gm eedback esistor Gate Shot Noise E hannel Noise Note that the I input-referred noise current spectrum begins to rise at a frequency lower than the I bandwidth g m f 8

29 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 9

30 ommon-gate & eedback I [Mohan JSS 000] eedback I ommon-gate ecall that the feedback I stability depends on the ratio of the input pole (set by ) and the amplifier pole Large variation in can degrade amplifier stability ommon-gate input stage isolates from input amplifier capacitance, allowing for a stable response with a variety of different photodetectors ransimpedance is still approximately /(+) 30

31 BJ ommon-base & eedback I ransformer-based negative feedback boosts gm with low power and noise overhead Input series peaking inductor isolates the photodetector capacitance from the I input capacitance High frequency techniques allow for 6GHz bandwidth with group delay variation less than 9ps [Li JSS 03] 3

32 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 3

33 ifferential Is ifferential circuits have superior immunity to power supply/substrate noise differential I output allows easy use of common differential main/limiting amplifiers his comes at the cost of higher noise and power How to get a differential output with a single-ended photocurrent input? wo common approaches, based on the amount of capacitance applied at the negative input 33

34 Balanced I balanced I design attempts to match the capacitance of the two differential inputs X his provides the best power supply/substrate noise immunity, as the noise transfer functions are similar ue to double the circuitry, the input-referred rms noise current is increased by sqrt() ssuming an high BW amplifier Z s and v OP v i i ON I s Same transfer function as the single - ended design 34

35 Pseudo-ifferential I pseudo-differential I design uses a very large capacitor at the negative input, such that it can be approximated as an ground X While not good to reject power supply/substrate noise, it does provide significant filtering of the noise he differential transimpedance is approximately doubled relative to the single-ended case ssuming an high BW amplifier Z s and v OP v i i ON I s 35

36 Offset ontrol ue to the single-ended photodetector signal, the differential output signal swings from 0 to V ppd, which can limit the dynamic range dding offset control circuitry can allow for an output swing of ±V ppd / 36

37 ifferential Shunt eedback I 37

38 genda Optical eceiver Overview ransimpedance mplifiers ommon-gate Is eedback Is ommon-gate & eedback I ombinations ifferential Is Integrating Optical eceivers 38

39 Optical X Scaling Issues raditionally, I has high and low in 3 db IN Headrooom/Gain issues in V MOS 3 Power/rea osts 4 I I f L I f 3dB IN 3dB g V m V V GS GS 0.8* V V V 0.* V VO VO 39

40 Integrating eceiver Block iagram [Emami VLSI 00] 40

41 emultiplexing eceiver emultiplexing with multiple clock phases allows higher data rate ata ate = #lock Phases x lock requency Gives sense-amp time to resolve data llows continuous data resolution 4

42 V Modified Integrating eceiver ifferential Buffer ixes sense-amp common-mode input for improved speed and offset performance educes kickback charge ost of extra power and noise Input ange = 0.6.V 4

43 eceiver Sensitivity nalysis esidual S Offset =.5mV Max V in (I VG ) = 0.6mV lock Jitter Noise otal Input Noise clk k samp 0. 9 j b v samp b 0.65mV mv buffer. 03mV at 6Gb/s tot samp buffer S clk. 59 V b for BE = 0-0 = 6.4 tot + Offset =.9mV mv P avg S Gb/s V b pd mv b in P avg (dbm)

44 Integrating eceiver Sensitivity est onditions 8B/0B data patterns (variance of 6 bits) Long runlength data (variance of 0 bits) BE < 0-0 [Palermo JSS 008] 44

45 Integrating X with ynamic hreshold ynamic threshold adjustment allows for un-coded data [Nazari ISS 0] 45

46 Integrating X with ynamic hreshold [Nazari ISS 0] 46

47 Next ime Main/Limiting mplifiers 47