Electronic Circuit Casebook. Dr. Lynn Fuller

Transcription

1 ROCHESTER INSTITUTE OF TECHNOLOGY MICROELECTRONIC ENGINEERING Electronic Circuit Casebook Dr. Lynn Fuller Webpage: 82 Lomb Memorial Drive Rochester, NY Tel (585) Fax (585) Department webpage: Electronic_Circuit_Casebook.ppt Page 1

2 OUTLINE Power Conditioning, Voltage Regulators Analog Switches, Two Phase Clocks Voltage Inverters, Doublers Amplifiers, Log Amplifiers RC Oscillators Voltage Controlled Oscillators Page 2

3 INTRODUCTION This document contains information on some electronic circuits that have been used as subcomponents in various microsystems. In general these circuits are realized using a hybrid combination of packaged integrated circuits, passive and active components combined at the printed circuit board (PCB) level. Some circuits have been converted to custom CMOS integrated circuits to provide on chip electronics and signal conditioning for MEMS devices. Page 3

4 Unregulated 9 to 24 volts DC POWER CONDITIONING LM317M Vout 1.2 to mA Positive Voltage Regulator 5 Volts mic2950 Vout mA Positive Voltage Regulator 3.3 Volts MAX1044 Vin 1.5 to 10 Vout = ~ Vin Voltage Converter 5 Volts UCC384 Vout mA Negative Voltage Regulator 3.3 Volts Page 4

5 BASIC VOLTAGE REGULATOR Voltage Regulator BJT or MOS Transistor Pass Element Unregulated Power Supply Vx Vout R1 (e.g. transformer, rectifier, capacitor filter) Av Vref R2 Gnd R Load Vout = Vref(1R2/R1) Page 5

6 VOLTAGE REGULATOR Vin Positive Voltage Regulator LM317M Vout 1.2 to mA R1 Vout Vout = 1.25(1R2/R1) R2 Gnd See: data sheets for LM317MD.pdf Page 6

7 TWO PHASE NON OVERLAPPING CLOCK Synchronous circuits that use the two phase non overlapping clock can separate input quantities from output quantities used to calculate the results in feedback systems such as the finite state machine. F1 F2 Page 7

8 TWOPHASE CLOCK GENERATORS A CLOCK B C R CLOCK t 1 CLOCKBAR S t 2 t 3 Q F1 F2 R S Q 0 0 Qn INDETERMINATE CLOCK BAR F1 t 1 t 2 t 1 F2 t 3 = Page 8

9 TRANSISTOR LEVEL SCHEMATIC OF 2 PHASE CLOCK V F1 F2 Clock V Page 9

10 LAYOUT OF TWO PHASE CLOCK CLOCK R t 2 Q F1 t 1 CLOCKBAR S t 3 F2 Page 10

11 TwoPhaseClock1.txt 1 V1 5V 8 VOUT1 2 V2 CLOCK M12 M11 M13 M17 10 M15 11 M14 M16 M18 9 VOUT2 M1 M3 M5 M7 M M2 M4 M6 M8 Substrate of all NMOS go to ground, node 0 Substrate of all PMOS go to 5V, node 1 M10

12 WINSPICE SIMULATION Page 12

13 TWO PHASE NON OVERLAPPING CLOCK Clock Φ1 Φ2 Page 13

14 VERSION 2 OF TWO PHASE NON OVERLAPPING CLOCK CLOCK R t 2 F1 t 1 CLOCKBAR S t 3 F2 1 VOUT1 8 M11 M13 M17 10 M15 11 M3 M5 9 VOUT2 3 4 M12 M14 M4 M6 M16 M V2 CLOCK V1 5V 6 M9 M7 5 M10 M8 M1 M2 7 Page 14

15 WINSPICE SIMULATION Page 15

16 ANALOG SWITCHES I V1 PMOS Vt= 1 S zero D V2 D NMOS Vt=1 5 S For current flowing to the right (ie V1>V2) the PMOS transistor will be on if V1 is greater than the threshold voltage, the NMOS transistor will be on if V2 is <4 volts. If we are chargeing up a capacitor load at node 2 to 5 volts, initially current will flow through NMOS and PMOS but once V2 gets above 4 volts the NMOS will be off. If we are trying to charge up V2 to V1 = 1 volt the PMOS will never be on. A complementary situation occurs for current flow to the left. Single transistor switches can be used if we are sure the Vgs will be more than the threshold voltage for the specific circuit application. (or use larger voltages on the gates) Page 16

17 (V to V) ANALOG SWITCH WITH (0 to 5 V) CONTROL S D Vout Vin D S V 05V Logic Control 5 V Page 17

18 SWITCHED CAPACITOR VOLTAGE DOUBLER F2 F2 C1 F 1 Vdd F1 F1 F1 C1 C Load R Load F2 Page 18

19 BASIC TWO STAGE OPERATIONAL AMPLIFIER Page 19

20 SPICE ANALYSIS OF OP AMP VERSION 2.inclrit_sub_param.txt m cmosn w=9u l=5u nrd=1 nrs=1 ad=45p pd=28u as=45p ps=28u m cmosn w=9u l=5u nrd=1 nrs=1 ad=45p pd=28u as=45p ps=28u m cmosp w=21u l=5u nrd=1 nrs=1 ad=102p pd=50u as=102p ps=50u m cmosp w=21u l=5u nrd=1 nrs=1 ad=102p pd=50u as=102p ps=50u m cmosn w=40u l=5u nrd=1 nrs=1 ad=205p pd=90u as=205p ps=90u m cmosp w=190u l=5u nrd=1 nrs=1 ad=950p pd=400u as=950p ps=400u m cmosn w=190u l=5u nrd=1 nrs=1 ad=950p pd=400u as=950p ps=400u m cmosn w=40u l=5u nrd=1 nrs=1 ad=205p pd=90u as=205p ps=90u vdd vss cprobe p Rprobe 2 0 1meg cc p mr cmosp w=6u l=10u nrd=1 nrs=1 ad=200p pd=60u as=200p ps=60u mr cmosp w=6u l=10u nrd=1 nrs=1 ad=200p pd=60u as=200p ps=60u *************** ************* 13.5kV/V gain ***dc open loop gain********* vi vi *.dc vi u.dc vi m *****open loop frequency characteristics***** *vi *vi dc 0 ac 1u *.ac dec g.end Page 20

21 OPERATIONAL AMPLIFIER Page 21

22 SOME BASIC ANALOG ELECTRONIC CIRCUITS These circuits should be familiar: R1 R2 Vin Vo Vo= Vin R2/R1 Inverting Amplifier R1 R2 Vo Vin Vo= Vin (1 R2/R1) NonInverting Amplifier C Vin Unity Gain Buffer Vo Vo= Vin Vin R Vo Vo= 1/RC Vin dt Integrator Page 22

23 COMPARATOR Vin V Vo V Vo Theoretical Vref V V Vref V Vin V Measured Page 23

24 R1 BISTABLE CIRCUIT WITH HYSTERESIS R2 V V Vo Theoretical Vin V Vo V TL V TH Vin V Measured Sedra and Smith pg 1187 Page 24

25 RC INTEGRATOR Va Va Vin t1 Vin t R C Vout Vout Va Va Smaller RC t Vout = (Va) [2Va(1e t/rc )] for 0<t<t1 If R=1MEG and C=10pF find RC=10us so t1 might be ~20us Page 25

26 OSCILLATOR (MULTIVIBRATOR) R1 V T R2 V Vo Vo V V t1 t C V R Bistable Circuit with Hysteresis and RC Integrator Page 26

27 RIT 100X DIFFERENTIAL VOLTAGE AMPLIFIER Va Vb Va Vb Rin Rin Gnd Rf Rf Vo1 Rin Gnd Rf Vo2 Rf = 100K Rin = 10K Page 27 1 X 1.5

28 INSTRUMENTATION AMPLIFIER V1 R1 R2 Vo1 R3 R4 Vo V2 R2 Vo2 R3 Gnd R4 Vo = (V2V1) R4 R3 1 2R2 R1 Page 28

29 CIRCUIT FOR LOW FREQUENCY CV MEASUREMENTS Vin 1 V/sec Ramp C R1=18 Mohm LM081 R2=1 Kohm R3=33 Kohm LM081 Vout Vout = (C R1 R3 / R2) (dvin/dt) Page 29

30 CV MEASUREMENTS ON NTYPE SILICON Low Frequency Inversion Depletion C C FB Accumulation High Frequency V T V FB 0 C min V Page 30

31 CV MEASUREMENTS ON PTYPE SILICON Accumulation Depletion C C FB Low Frequency Inversion High Frequency C min V FB V T 0 V Page 31

32 PEAK DETECTOR Variable Vin C Vo Diode reverse leakage current ~100nA Page 32

33 DESIGN EXAMPLE CAPACITOR SENSOR R1 R2 V V R C C Vref V Vo C R Square Wave Generator RC Integrator & Capacitor Sensor Buffer Peak Detector Comparator Display Page 33

34 EXAMPLE LABORATORY RESULTS Smaller Capacitance Larger Capacitance Square Wave Generator Output Buffer Output Display Page 34

35 PHOTODIODE I TO V LINEAR AMPLIFIER 3.3V R1 10K I p n R2 20K 3.3V NJU R3 10K IR LED R4 100K 3.3V NJU Vout 0 to 1V Gnd Gnd Page 35

36 PHOTO DIODE I TO V LOG AMPLIFIER 3.3V R1 20K IR LED I n p 3.3V NJU703 1N Vout 0 to 1V Linear amplifier uses 100K ohm in place of the 1N4448 Vout vs. Diode Current Gnd Gnd Linear Amplifier Log Amplifier Output Voltage (V) Photodiode Diode Current (ua) Page 36

37 PHOTO DIODE I TO V INTEGRATING AMPLIFIER Reset C Internal 100 pf Ri Rf Analog Vout Integrator and amplifier allow for measurement at low light levels Page 37

38 SIGNAL CONDITIONING FOR TEMPERATURE SENSOR R1 20K 3.3V I p n 0.2 < Vout < 0.7V Gnd Page 38

39 SIGNAL CONDITIONING FOR TEMPERATURE SENSOR pmosfet 3.3V Constant Current Source I 0.2 < Vout < 0.7V Gnd Page 39

40 OP AMP CONSTANT CURRENT SOURCE Floating Load Grounded Load Vs Load Vo I = Vs/R Vs R2 R1 Rx R3 Rx/R1=R3/R2 Vo R Load I = Vs/R2 Page 40

41 OVER TEMPERATURE DETECTOR CIRCUIT R upper = 100k R lower = 47k 1N4448 V in (T) 3V MC33204 V ref 100k V out 220 LED Jirachai Getpreecharsawas, 2009 V in < V ref T > T ref : LED is ON V in V ref T T ref : LED is OFF Diode Temperature Sensor Based on the same principle of thresholdcrossing detection. Diode voltage (V diode ) and V in are the monotonically decreasing functions of temperature (T). V V ref To increase the sensitivity, use more diodes. in R upper ( T) = V ( T) R R upper R upper lower ( Tref ) = Vdiode ( Tref ) Vsupply upper R R lower lower diode R upper R Rlower R upper lower R R lower V supply Page 41

42 WATER CONDUCTIVITY LOG I TO V AMPLIFIER 1N4448 Conductivity (TDS) I Gnd 3.3V NJU703 Vg ~ 0.1 V V NJU Vout 0 to 1V K Gnd 10K Page 42

43 RC OSCILLATOR USING INVERTER WITH HYSTERESIS 1 R=10K V1 5V 3 M2 2u/16u Vout M3 2u/15u Inverter with Hysteresis R 2 7 V2=05 Or 50 M1 2u/16u C Vout RC Oscillator Page 43

44 RC OSCILLATOR, INVERTER WITH HYSTERESIS 1 M4 M5 All NMOS Realization V1 9V M7 C1 M8 2 7 M2 M1 3 M3 4 M6 3.0pF *TRANSISTORS M RITSUBN49 L=2U W=64U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=2U W=16U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=2U W=64U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=32U W=8U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=64U W=8U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=2U W=128U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=128U W=4U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 M RITSUBN49 L=64U W=4U ad=96e12 as=96e12 pd=44e6 ps=44e6 nrd=0.025 nrs=0.025 * Page 44

45 RC OSCILLATOR CMOS VERSION Page 45

46 VOLTAGE CONTROLLED OSCILLATOR Page 46

47 Page 47

48 PRESSURE SENSOR ZERO AND SPAN COMPENSATION Vo Rzt R1 R2 Rzb Vs Rst R3 R4 Rsb Gnd Vo Dr. Lynn Fuller 4/18/2007 Bridge_Balance.xls This spread sheet can be used to find resistor values used to compensate a wheatstone bridge resistor pressure sensor for output offset voltage and span. If we assume that the resistors are TaN thin film resistors that are adjusted by laser trimming then the trimmed value has to be higher than the nominal value. First adjust the value of Rzt and Rzb to set Vout trimmed to zero. Then set Rst and Rsb to make the trimmed stressed value equal to the specified output voltage at maximum applied pressure. Vout Vout Vout Vout Vsupply 10 volts no trim no trim trimmed trimmed nominal stressed nominal stressed nominal stressed R Vo volts R Vo volts R Vout mv R %change 0.5 when maximum pressure is applied (stressed) nominal Rst ohms Vo = Itotal * Rsb Iright * R4 Rsb ohms Vo = Itotal * Rsb Ileft * R2//Rzb Vout = Vo Vo Rzt ohms Rzb ohms no trim no trim trimmed trimmed nominal stressed nominal stressed Rleft (Rzt//R1) (Rzb//R2) Rright R3R4 Rtotal Rleft//Rritght Rst Rsb Itotal Vs/Rtotal Vbridge Vs Itotal (RstRsb) Ileft Vbridge/Rleft Iright Vbridge/Rright Page 48

49 POWER OUTPUT STAGE V V Vin V Vo Rload V Page 49

50 FLOW SENSOR ELECTRONICS 6 Volts Constant Power Circuit for the Heater R2 Upstream Resistor Downstream Resistor Vout R1 Gnd 6 Volts Vout near Zero so that it can be amplified Vref Analog Multiplier AD534 AD534_b.pdf Heater 10 Ω PLAY STOP Page 50

51 CONSTANT TEMPERATURE CIRCUIT Setpoint R=V/I Analog Divider Using AD534 MORE 1000 V I 10 Ω 9 Volts Gnd I Heater Page 51

52 WHEATSTONE BRIDGE CONSTANT TEMP CIRCUIT Page 52

53 DELTA CAPACITANCE TO AC VOLTAGE Vin Cx R Vo If Cx is fixed Vo is zero. If Cx changes there will be a change in current and a corresponding change in Vo Example: Let Vin = 3 volts, C = 10 pf, microphone action causes C to change by 0.1pF at 1000 Hz. Calculate the output voltage. Page 53

54 CAPACITIVE OIL LEVEL DETECTION CIRCUIT Jirachai Getpreecharsawas, M 100k 100k 3V MC M 3V MC N4448 3V MC p 0.1µ k 2p 3.5p 100k LED This circuit is designed to detect the presence of oil at a specified level. The fundamental operation is based on a thresholdcrossing detection of signal level (voltage) due to a capacitive change of the probe. Contacting with different dielectric materials, either in air or immerging in oil, results in this change. As a consequence, the RC time constant of the probe is also altered, causing the probe to charge up to a different maximum voltage level within a given period of time. A peak detector is used to measure this voltage level, and the thresholdcrossing detection is then carried out by the final stage opamp, as shown. Page 54

55 Jirachai Getpreecharsawas, 2008 WIRELESS CAPACITANCE TO DIGITAL RC Oscillator RC Oscillator 2.4 khz 5 Hz 5 Hz 3 V CTS Stop Bit TX SCLK RCLK RX 1 10bit (Left) Shift Register 1 RTS Bluetooth Serial RF Link Start Bit 0 0 CCKEN RCO CCLK 0 8bit Binary Counter 0 0 RCLK CCLR Internal Counter RC Oscillator Sensor Page 55

56 WIEN BRIDGE OSCILLATOR CIRCUIT PLAY STOP CMOS Analog Circuit Design, Phillip Allen, Douglas Holbert, Holt, Rinehard and Winston. 1987, pg Page 56

57 LOOP GAIN OF WIEN BRIDGE OSCILLATOR CMOS Analog Circuit Design, Phillip Allen, Douglas Holbert, Holt, Rinehard and Winston. 1987, pg Page 57

58 LOW PASS FILTER Vin R1 C2 R2 Vout Derive an expression for Vo/Vin Plot 20Log 10 (Vo/Vin) vs frequency Verify using SPICE Verify by building the circuit Vo/Vin = R2/R1 ω = 2 π f ω1 = 1/R2C2 1 1 j ω/ω1 1 SR2C2 1 f Page 58

59 HIGH PASS FILTER Vin C1 R1 R2 Vout Derive an expression for Vo/Vin Plot 20Log 10 (Vo/Vin) vs frequency Verify using SPICE Verify by building the circuit Vo/Vin = R2/R1 ω = 2 π f ω1 = 1/R1C1 j ω/ω1 1 j ω/ω1 SR1C1 SR1C1 1 f Page 59

60 Vin C1 R1 C2 R2 GENERAL FILTER Derive an expression for Vo/Vin Plot 20Log 10 (Vo/Vin) vs frequency Verify using SPICE Verify by building the circuit Vout Vo/Vin = R2/R1 SR1C1 1 SR2C2 1 ω = 2 π f ω1 = 1/R1C1, ω2 = 1/R2C2 = R2/R1 1 j ω/ω1 1 j ω/ω2 Page 60

61 COMBINATIONS OF FILTERS Vo/Vin = R2/R1 1 j ω/ω1 1 j ω/ω2 Two General Filters in series General ω1, ω2 General ω3, ω4 Vo/Vin = R2R4/R1R3 1 j ω/ω1 1 j ω/ω2 1 j ω/ω3 1 j ω/ω4 2 nd Order lowpass, highpass, bandpass, bandrejection and all pass filter Page 61

62 SKETCH OF VARIOUS FILTER FREQUENCY RESPONSE Vo/Vin = R2R4/R1R3 1 j ω/ω1 1 j ω/ω2 1 j ω/ω3 1 j ω/ω4 ω1 = ω3 < ω2 = ω4 ω2 = ω4 < ω1 = ω3 ω1 < ω2 < ω4 < ω3 ω2 < ω1 < ω3 < ω4 Page 62

63 REFERENCES 1. Dr. Fullers webpage Page 63