THE majority of current global positioning satellite (GPS)

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1 1618 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 45, NO. 11, NOVEMBER 1997 A Dual-Band Circularly Polarized Aperture-Coupled Stacked Microstrip Antenna for Global Positioning Satellite David M. Pozar, Fellow, IEEE, and Sean M. Duffy Abstract This paper describes the design and testing of an aperture-coupled circularly polarized antenna for global positioning satellite (GPS) applications. The antenna operates at both the L1 and L2 frequencies of 1575 and 1227 MHz, which is required for differential GPS systems in order to provide maximum positioning accuracy. Electrical performance, lowprofile, and cost were equally important requirements for this antenna. The design procedure is discussed, and measured results are presented. Results from a manufacturing sensitivity analysis are also included. Index Terms Microstrip antennas. I. INTRODUCTION THE majority of current global positioning satellite (GPS) receivers operate using only the L1 frequency of 1575 MHz and typically use a circularly polarized microstrip antenna on a relatively thick substrate of high-dielectric constant. Such an antenna has good low angle coverage, adequate bandwidth, and axial ratio and can be supplied at low cost in large volumes. For more demanding applications, such as surveying systems employing differential GPS, an antenna covering both L1 and L2 bands is required. The aperture coupled microstrip patch antenna [1] has a number of desirable features compared to other feeding techniques, such as doubled substrate area, a noncontacting feed transition, and shielding of the feed network from the radiating aperture. It has also been demonstrated that aperture coupling can be used to substantially increase bandwidth beyond the few percent that is typically obtained with traditional patch designs. Increased bandwidth can be achieved with a single patch using a relatively thick low-dielectric constant substrate [2], [3] or by using two (or more) stacked patches [4], [5]. Impedance bandwidths ranging from 20 to 35% have been obtained with these methods. Circular polarization can be generated from aperturecoupled elements by using off-center coupling apertures [6] or with crossed slots [3], [7]. In both cases, two orthogonal linearly polarized modes are independently excited with equal amplitudes and a 90 phase shift, leading to good axial ratio bandwidth. This is in contrast to the technique of using a single aperture feed with an almost square patch element, which typically results in a very narrow axial ratio bandwidth Manuscript received April 23, 1996; revised June 20, D. M. Pozar is with the Department of Electrical and Computer Engineering, University of Massachusetts at Amherst, Amherst, MA USA. S. M. Duffy is with Lincoln Laboratories, Lexington, MA USA. Publisher Item Identifier S X(97) [8]. Using two slots or a crossed slot has been shown to be capable of producing 2-dB axial ratio bandwidths of 25% or more, but requires feed networks with hybrid couplers. The present GPS application requires only a relatively narrow bandwidth of approximately 40 MHz (3%) at 1227 and 1575 MHz each, as opposed to complete coverage over this band. The bandwidth of the entire frequency range is 25%, which could be covered with a single aperture-coupled patch element [3], but it was felt that dual-band spot coverage of L1 and L2 was actually an advantage in terms of reducing outof-band interference. Such dual-band coverage is most easily obtained using two stacked microstrip elements as opposed to a single-patch element, so this is the configuration used for the antenna described here. Since the aperture coupled antenna radiates a small back lobe from the coupling slots [1], [3], it is often desirable to provide a ground-plane shield below the feed layer. If this ground plane is located at least 0.05 to 0.1 from the microstrip feed substrate, it will have little effect on the impedance or radiation properties of the antenna. It is also possible to reduce the back lobe by introducing lossy absorber material in the region below the feed layer. II. DESIGN PROCEDURE The design of the dual-band aperture-coupled stacked-patch antenna is complicated by the fact that there are at least eight interacting design parameters that must be determined in addition to details of the microstrip feed network. We summarize these parameters and their effects below in relation to the geometry shown in Fig. 1. Bottom Antenna Substrate Thickness and Dielectric Constant: A thick substrate with a low dielectric constant yields large bandwidth, but coupling to the bottom patch decreases with substrate thickness, requiring a compensating increase in aperture size and, thus, increased back radiation. Bottom Patch Length and Width: Since circular polarization is to be generated by exciting two orthogonal linear polarizations with a 90 phase shift, the patches must be square. It is tempting to think that the patches in a dualband stacked-patch design resonate independently at each frequency, but this is not true. Due to strong coupling between the patches, the resonant sizes of the patches must be determined simultaneously. Top Antenna Substrate Thickness and Dielectric Constant: The substrate thickness and dielectric constant control the X/97$ IEEE

2 POZAR AND DUFFY: DUAL-BAND CIRCULARLY POLARIZED APERTURE-COUPLED MICROSTRIP ANTENNA 1619 Fig. 2. Measured return loss of stacked-patch antenna with a single-aperture feed (linearly polarized prototype). Fig. 1. Geometry of the aperture-coupled stacked-patch microstrip GPS antenna. bandwidth of the resonance associated with the top patch as well as the coupling between the two patches. Top Patch Length and Width: The top patch must also be square for circular polarization. Since it is excited only by proximity coupling to the bottom patch, its coupling level is strongly dependent on the separation of the two resonant frequencies. The coupling is also affected by whether the top patch is larger or smaller than the bottom patch. Aperture Length and Width: Coupling from the feed line to the bottom patch is controlled primarily by the length of the aperture. A longer aperture increases coupling but also increases undesirable back radiation. The aperture also introduces a shift in the resonant frequency of the bottom patch. The aperture width also affects coupling strength, but to a lesser degree than the length. It should be obvious that a purely empirical design with such a large number of strongly interacting parameters is practically impossible. And while computer-aided design (CAD) tools are available for modeling such an antenna [9], the longrun time of such full-wave numerical solutions preclude their use unless many of the design parameters can be fixed using other means. For this project we relied on a combination of previously derived design data, empirical tuning, and computer modeling to arrive at a functional design that met the frequency and bandwidth requirements. This step was first carried out for a linearly polarized version of the antenna using a single slot, a convenient simplification that is possible because the two crossed slots for the final circularly polarized case are almost perfectly decoupled when fed with balanced microstrip lines [3]. CAD tools were then used to design a microstrip feed network for circular polarization and to study the effect of manufacturing tolerances on the performance of the antenna. These three steps are discussed in more detail below. A. Design of Stacked Elements with Single-Slot Feed To simplify the design process we began with a linearly polarized stacked-patch geometry using a single coupling aperture and a single centered microstrip feed line. The substrate and patch sizes determined from this starting point should be close to optimum for the final circularly polarized design, while the aperture size is expected to require a slight increase when balanced feed lines are used. Based on work done in [3], both antenna substrates were made from highdensity Rohacell foam material with thin Duroid skins for the patch metallization. This combination provides maximum bandwidth for a given substrate thickness, minimizes material cost, and results in a very lightweight structure. Scaling down in frequency from the aperture coupled antenna described in [3] and making a few empirical trials led to reasonable values for the antenna substrate thicknesses. The thickness for both antenna substrates was cm, with cm-thick Duroid 5870 layers for the patches. Patch sizes were initially estimated using simple approximations and then tuned empirically. This was a tedious process because the two patches interact with each other and with the slot, which must be adjusted in length for the proper impedance match. The main objective at this point is to obtain a good impedance match seen by the feed line at the frequencies of 1227 and 1575 MHz and preferably with a strong impedance mismatch outside these bands. The patch sizes were determined to be 8.35 cm square for the bottom patch, and 7.90 cm for the top patch. Having the top patch smaller than the bottom patch is in contrast to most stackedpatch designs (e.g., with probe feeds [8]), but this was found to give a slight capacitive shift in the impedance locus that compensated for the added inductance of the top patch relative to the bottom patch due to its larger substrate thickness. A slot length of 5.1 cm was found to give an input impedance close to 50. The length of the tuning stub on the feed line was chosen to be a quarter-wave long at the mid-band frequency. Fig. 2 shows the measured return loss of the linearly polarized prototype element with a single feed line. Note the good impedance match at frequencies near L1 and L2, and the mismatch away from those frequencies. There is a slight shift of a few percent in the optimum match at both frequencies and this was left to be adjusted during the next design step. As shown in Fig. 3, each of the crossed coupling slots in the circularly polarized antenna is fed with a balanced

3 1620 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 45, NO. 11, NOVEMBER 1997 Fig. 3. Microstrip feed network layout for circularly polarized stacked-patch antenna with crossed-slot feed. pair of microstrip lines. We thus require knowledge of the effective series impedance seen by a single feed line when both lines are driven in phase with equal amplitudes. For this purpose, we can consider the above linearly polarized prototype antenna with a single coupling aperture, but fed with a pair of balanced offset microstrip feed lines. A simple analysis shows that can be determined directly from the two-port parameters of the two feed line ports as where is the characteristic impedance of the feed lines and due to symmetry. The evaluation of (1) can be done on the HP-8510 network analyzer using the trace math option with the measured two-port parameters. The phase reference for these quantities should be taken at the center of the coupling aperture. The effective impedance level is reduced from that of the single-feed line case due to the offset of the balanced feed lines from the center of the aperture, thus requiring an increase in aperture length to 5.3 cm to obtain. B. Design of Microstrip Feed Network for Circular Polarization The purpose of the feed network is to drive the crossed slots with equal amplitudes and a 90 phase shift to obtain righthand circular polarization (RHCP) and to provide a matched input to the antenna. This was done using the microstrip feed network, shown in Fig. 3. Three Wilkinson power combiners are used to add the signals from the four feed lines at the slots with equal amplitudes and the proper phase shifts. As seen from Fig. 3, each pair of feed lines crosses each slot arm in opposite directions (the crossed slot is not shown in Fig. 3, but is oriented in the vertical and horizontal directions). This (1) introduces a 180 phase shift between lines at opposite ends of each slot arm. This phase shift, together with the 90 phase shift between arms required for circular polarization, results in a sequential rotation in phase of 0,90, 180, and 270 around the four feed points. The two Wilkinson combiners closest to the feed points combine power from adjacent feed ports after introducing a 90 phase shift in one of the input lines. The last Wilkinson combiner adds the outputs of the previous combiners after introducing a 180 phase shift in one of its input lines. The input and output impedances of the combiners is 50 for simplicity in impedance matching. Each Wilkinson combiner also requires a 100 isolation chip resistor located at the gap between the two input lines. This resistor does not absorb power unless there is an imbalance in the input signals. The feed network was designed for a nominal center frequency of 1.4 GHz. As discussed in [3], the parallel feed arrangement used here has the advantage of providing accurate phase and amplitude for signal combining. In contrast to the simpler series feeding technique, however, the parallel-feed method takes more space and incurs more loss due to the presence of three power combiners. C. Sensitivity Analysis of Design Since this antenna was intended for large-volume lowcost construction, the sensitivity of its design to variations in materials and fabrication from nominal values is important. While there are over 30 distinct dimensions and other parameters associated with this antenna, including several sets of substrate parameters, patch and slot dimensions, element alignments, loss effects, and deformations, most have only a slight effect on performance. This is fortunate, since it is difficult to give any sort of absolute tolerance for most of these parameters, although in virtually every case, performance gradually degrades with the variation of any single parameter. Thus, the goal of this sensitivity analysis was to first identify the most critical design parameters and then quantify an estimate of the required accuracy for these parameters. In addition, while it might be possible to define a priori how much performance degradation (e.g., loss in gain, rise in axial ratio, etc.), in practice it is often desirable to allow a tradeoff between performance and the cost of materials and manufacturing. Also, compared to other types of printed antennas, aperture-coupled microstrip antennas generally offer a more robust design because of the nature of the aperture feed and the lack of soldered feed points. Below, we discuss the most critical parameters associated with this type of antenna based on practical experience and computer modeling. Patch Dimensions: The patch lengths and widths directly control the resonant frequencies of the lower and upper bands and are therefore critical. The required accuracy of these dimensions can be estimated from the operating bandwidth of 40 MHz and the fact that resonant frequency is inversely proportional to patch size. If we assume the shift in operating frequency of the antenna should be less than 0.5%, then the corresponding tolerance on the 7.9-cm dimension of the smallest patch would be cm or about in. Most commercial etching can be routinely done to tolerances of

4 POZAR AND DUFFY: DUAL-BAND CIRCULARLY POLARIZED APERTURE-COUPLED MICROSTRIP ANTENNA 1621 Fig. 5. Measured return loss for the final dual-band circularly polarized aperture-coupled stacked-patch antenna. Fig. 4. Calculated effect on impedance locus (Z e ) caused by 65% errors in slot length in or less, so the accuracy of the patch size should not be a problem. Slot Size: The length of the coupling slots controls the strength of coupling from the feed lines to the patches and, to a lesser extent, the amount of reactive loading that can detune the patch elements. The primary effect of a change in the length of the slot will be to shift the locus of the impedance seen by a feed line from the center of the Smith chart. A typical example is shown in Fig. 4, which was computed using the full-wave moment-method model of [9]. The required tolerance for this parameter depends on the allowable impedance mismatch, but we have found from modeling that acceptable results can generally be obtained if the slot length is accurate to within 5%. For a slot length of 5.3 cm, this amounts to cm in. Again, this accuracy is well within the typical range of accuracies that can be done by commercial etching services. The slot width also controls the coupling level, but to a much lesser degree than the slot length. Substrate Thickness: A variation in the thickness of the antenna substrates can change the coupling level and detune the operating frequency. Both of these effects, however, are less sensitive to substrate thickness than either patch size or substrate permittivity. It is impossible to give a single tolerance for substrate thickness because there are many combinations of different layers having different thicknesses and dielectric constants; thus we take, as a worst case, a tolerance of 5%, which will give a change of a few ohms in the impedance locus. This is undoubtedly a very conservative value (high by perhaps a factor of two), giving thickness accuracies of about in for the in Duroid layers and in for the 0.25-in foam layers. The manufacturer s specifications for Duroid gives a thickness tolerance of in for in Duroid, well within our estimate of thickness accuracy. No data could be found for the thickness tolerance of Rohacell Fig. 6. Measured axial ratio versus frequency for the final dual-band circularly polarized aperture-coupled stacked-patch antenna. foam, but the estimated requirement of in should be easily attainable. A more significant concern is the need to avoid compressing the foam during the bonding procedure and the possibility of air gaps between layers due to nonuniform bonding. Substrate Permittivity: Substrate permittivity controls the resonant frequency of the antenna elements and the phasing of the feed network lines (the former being more critical). Since resonant frequency varies as the reciprocal of the square root of the dielectric constant, the above figure of an allowable 0.5% shift in frequency gives a tolerance of 1% on the dielectric constant of the dielectric materials. For the Duroid layers, this would be ; the manufacturer s specification for the dielectric constant of Duroid is 0.02, which is within this range. In fact, this tolerance is again very conservative because the Duroid layer occupies only a small fraction of the total foam/duroid antenna substrate thickness, allowing a significant upward weighting of this tolerance by approximately a factor of (0.25 in/0.03 in ) 8. The dielectric constant of the foam is specified as 1.08, but there is not an accurate tolerance given for this value. This is probably not critical because of the very low value of this permittivity. This discussion also suggests that it should be possible to use a cheaper PTFE-type substrate in place of the Duroid material. Alignment of Patches and Slots: A likely source of error during manufacturing of this antenna is misalignment of

5 1622 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 45, NO. 11, NOVEMBER 1997 Fig. 7. (a) (a) Measured spinning linear pattern of the circularly polarized aperture-coupled stacked-patch antenna at 1227 MHz. the patches with the feed slots. Fortunately, the center-fed aperture-coupled element couples to the magnetic field under the patch, which has a local maximum at the center of the patch. Thus, the first derivative of the variation in coupling with position is zero for a slot centered below the patch. Calculations have shown that there is virtually no change in impedance locus for a misalignment of the patches in either direction by 5%. For these patches this translates to about - in cm in, an alignment accuracy which should not be hard to achieve. One consideration, however, that we have not been able to estimate is the effect of misalignment on the axial ratio of the circularly polarized radiation. Good axial ratio depends critically on accurate phasing, which conceivably could be affected by misalignment. Deformation and Flatness of Substrates: Proper operation of this antenna assumes flatness of the substrates, with no voids between layers. It is practically impossible to model this type of error, as it depends on not only the amount of deviation from flatness, but also the location of the void relative to the patches. A void near the middle of a patch would probably have little effect, but one below the radiating edge (where the -field intensity is greatest) could have a significant effect. All we can say is that it would probably be adequate to keep any void smaller in thickness than the above dielectric thickness tolerances. It would seem, however, that flatness should be good if a quality bond is used. Feed Network Layout: The feed network fabrication is generally less critical than most of the above antenna parameters, with phase delay and matching sections probably being the most sensitive aspects of the feed network. Since the feed network is fairly broad band and designed for a nominal operating frequency at the center of the band, we do not expect the tolerances on the etching accuracy or the material parameters to be any more severe than those for the antenna elements. Standard commercial etching tolerances should be more than adequate. In summary, the circularly polarized aperture-coupled stacked-patch antenna described here is very robust in terms of its sensitivity to fabrication and material tolerances. This is primarily a result of the relatively wide-band design, the nature of the aperture coupling mechanism, and the relatively low frequency of operation. In comparison, a probe-fed patch would require four balanced probe feeds to yield equivalent symmetry and the resulting axial ratio and impedance match would be much more sensitive to positioning errors in the probe feeds. III. PERFORMANCE OF APERTURE-COUPLED STACKED-PATCH ANTENNA The final circularly polarized aperture-coupled stackedpatch antenna was tested for impedance match, axial ratio, radiation patterns, and group delay. Fig. 5 shows the measured

6 POZAR AND DUFFY: DUAL-BAND CIRCULARLY POLARIZED APERTURE-COUPLED MICROSTRIP ANTENNA 1623 (b) Fig. 7. (Continued.) (b) Measured spinning linear pattern of the circularly polarized aperture-coupled stacked-patch antenna at 1575 MHz. return loss over the frequency band of 1 2 GHz. Note that the distinct dual-band response seen in Fig. 2 for the linearly polarized prototype is washed out by the presence of the feed network and that the return loss is better than 15 db from approximately 1227 to 1700 MHz. Fig. 6 shows the measured axial ratio versus frequency where best results occur at approximately 1227 and 1475 MHz. This shift of the upper frequency band was also seen in the relative gain measurements of the antenna and was apparently due to a change in the adhesive used between layers. The prototype models were fabricated using vacuum bag techniques to allow easy modification to individual elements during the design stage, while the final version used cyanoacrylate glue to bond layers together. The resulting frequency shift could be corrected with a small change in patch size in future prototypes, but was not done at the time because of the likelihood that the substrate material would be changed. Spinning linear radiation patterns are shown in Fig. 7(a) and (b) for the two principal orientations of the antenna. Observe that good broadside axial ratio is obtained, with a back-lobe level of about 15 db or less, relative to the main lobe. This back radiation is due to the coupling slots and would generally be shielded by a ground plane located a short distance below the feed network. Also observe from the patterns that the 10- db beamwidths are about Due to the lack of a standard gain-horn antenna at this frequency, we were not able to measure absolute gain for the stacked-patch antenna. Relative gain measurements indicated equal gain at the L1 and L2 frequencies to within 0.5 db. Finally, we consider the group delay response of the aperture-coupled stacked-patch antenna. Group delay is used quite often to characterize microwave filters and other components [10], but rarely for antennas. Group delay is defined as the derivative of the transfer phase response versus frequency and is a measure of the time delay of a narrow-band signal packet. In microwave components, group delay versus frequency is of most interest, while in the case of an antenna it is useful to consider the variation of group delay versus elevation and azimuth angles. Thus, in GPS applications, maximum accuracy requires that the receiving antenna have a uniform group delay response for all angles of incidence in most of the upper hemisphere, otherwise signals from satellites at different elevation angles will experience timing errors. As discussed in [11], it is generally not possible to provide such ideal phase characteristics for practical antennas, but certain antenna designs have much better characteristics than others. (2)

7 1624 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 45, NO. 11, NOVEMBER 1997 reliably due to repeatability in the antenna mounting, but this level of variation is quite satisfactory for differential GPS systems. The data in Fig. 8(a) and (b) were measured with an off-board 180 coaxial power divider replacing the initial Wilkinson divider of the feed network (Fig. 3), as the unequal path lengths of the printed divider was found to degrade the group-delay characteristics significantly. (a) (b) Fig. 8. (a) Measured group-delay variation of the circularly polarized stacked-patch antenna near 1227 MHz for elevation angles of 0 (broadside) and 60. (b) Measured group-delay variation of the circularly polarized stacked-patch antenna near 1575 MHz for elevation angles of 0 (broadside) and 60. We have found that good-phase or group-delay response depends critically on the inherent symmetry of the antenna and its feed [12]. Thus, a GPS microstrip antenna with a single probe-feed point has an asymmetry that can result in up to 3-nS variation in group delay for elevation angles between 0 and 60 from the zenith. The inherent feed symmetry of an aperture-coupled microstrip antenna, however, can yield a group-delay variation of less than 500 ps. In the present design, care must also be taken to ensure that the feed network has equal path delays from the input connector to the slot-feed points, otherwise the feed network may degrade the group delay characteristics. Also, better phase response is generally obtained with a broad-band feed network, as opposed to one that is tuned to a narrow-band response. Fig. 8(a) and (b) shows the measured group-delay response of the aperture-coupled stacked-patch antenna at the L2 and L1 frequency bands. This data was measured using an HP8510B network analyzer with the test antenna oriented at broadside and at 60 (relative to the transmit antenna) inside an anechoic range. The data is taken versus frequency with a center frequency at 1227 and 1575 MHz in Fig. 8(a) and (b), respectively. An ideal phase response would be indicated by overlapping curves at the two elevation angles; here we see a maximum deviation near the center frequency of approximately 500 ps (one vertical division) at the lower frequency and somewhat less than this at the upper frequency. Variations in group delay less than about 500 ps are difficult to measure IV. CONCLUSION As a result of using low-dielectric constant antenna substrates, the patch elements of the circularly polarized aperturecoupled antenna were relatively large, which led to relatively narrow pattern beamwidths (10 db) on the order of While the broadside gain was good, the resulting gain at low-elevation angles was less than that obtained from GPS antennas on higher dielectric constant material. Improved lowelevation angle gain could be obtained by using antenna substrates with higher dielectric constants in order to reduce the size of the patch elements. This would simplify construction by reducing the number of layers from five to three, but the material cost of the antenna substrates would be higher than that of the foam material. Another improvement would be to design a more compact dual-band feed network, to replace the large broad-band feed of the present design. The feed should also have features of symmetry to maintain the intrinsically good group-delay response of the antenna itself. Alternatively, an off-board coaxial power divider assembly could be used if its price/performance qualities are advantageous. REFERENCES [1] D. M. Pozar, Microstrip antenna aperture coupled to a microstripline, Electron. Lett., vol. 21, pp , Jan [2] F. Croq and A. Papiernik, Wideband aperture coupled microstrip subarray, Electron. Lett., vol. 26, pp , Aug [3] S. D. Targonski and D. M. Pozar, Design of wideband circularly polarized aperture coupled microstrip antennas, IEEE Trans. Antennas Propagat., vol. 41, pp , Feb [4] J. Wang, R. Fralich, C. Wu, and J. Litva, Multifunctional aperture coupled stack antenna, Electron. Lett., vol. 26, pp , Dec [5] F. Croq and D. M. Pozar, Millimeter wave design of wide-band aperture coupled stacked microstrip antennas, IEEE Trans. Antennas Propagat., vol. 39, pp , Dec [6] A. Adrian and D. H. Schaubert, Dual aperture coupled microstrip antenna for dual or circular polarization, Electron. Lett., vol. 23, pp , Nov [7] C. H. Tsao, Y. M. Hwang, F. Killberg, and F. Dietrich, Aperture coupled patch antennas with wide-bandwidth and dual polarization capabilities, in IEEE Antennas Propagat. Symp. Dig., Syracuse, NY, June 1988, pp [8] D. M. Pozar and D. H. Schaubert, Eds., Microstrip Antennas: The Analysis and Design of Microstrip Antennas and Arrays. Piscataway, NJ: IEEE Press, [9] N. K. Das and D. M. Pozar, Multiport scattering analysis of general multilayered printed antennas fed by multiple feed ports: Part I Theory; Part II Applications, IEEE Trans. Antennas Propagat., vol. 40, pp , May [10] D. M. Pozar, Microwave Engineering. Reading, MA: Addison-Wesley, [11] J. M. Tranqilla and S. R. Best, A study of the quadrifilar helix antenna for global positioning system (GPS) applications, IEEE Trans. Antennas Propagat., vol. 38, pp , Oct [12] D. M. Pozar, Group delay characteristics of microstrip antennas, Trans. Antennas Propagat., to be published.

8 POZAR AND DUFFY: DUAL-BAND CIRCULARLY POLARIZED APERTURE-COUPLED MICROSTRIP ANTENNA 1625 David M. Pozar (S 74 M 80 SM 88 F 90) received the B.S.E.E. and the M.S.E.E. degrees from the University of Akron, Akron, OH, in 1975 and 1976, respectively, and the Ph.D. degree from Ohio State University, Columbus, in During the course of his undergraduate studies he spent one year as an Engineering Co-Op Assistant at the National Security Agency, Fort Meade, MD. He was a Graduate Research Assistant at the ElectroScience Laboratory, Ohio State University, Columbus, while pursuing the Ph.D. degree and became a Research Assistant upon completion of that degree. He joined the faculty at the University of Massachusetts at Amherst in In 1989 he became a Professor of electrical and computer engineering. In 1988 he spent a six-month sabbatical leave at École Polytechnique Federale de Lausanne, Lausanne, Switzerland. Since 1993 he has served as a Distinguished Lecturer for the IEEE Antennas and Propagation Society. He has published over 100 papers on printed antennas and phased arrays and is the author of Antenna Design Using Personal Computers (Norwood, MA: Artech House, 1985), Microwave Engineering (New York: Wiley, nd ed.), and Microstrip Antennas (Piscataway, NJ: IEEE Press, 1995). He is also the author of PCAAD, a software package for personal computer-aided antenna design. Dr. Pozar belongs to the Antennas and Propagation Society and the Microwave Theory and Techniques Society. He was a member of the IEEE AP-S AdCom ( ). In 1981 he received the Outstanding Professor for 1981 Award from Eta Kappa Nu, the Student Honor Society. In 1984 he received an NSF Presidential Young Investigator Award and the Keys to the Future Award from the IEEE Antennas and Propagation Society. In 1985 he received the University of Massachusetts Engineering Alumni Association Outstanding Junior Faculty Award. In 1986 he received the R. W. P. King Best Paper Award from the IEEE Antennas and Propagation Society. In 1987 he received the URSI Issac Koga Gold Medal for his work on printed antennas and phased arrays. He again received the R. W. P. King Best Paper Award in In 1989 he received the United Technologies Corporation Outstanding Teaching Award. In 1995 he received the College of Engineering Outstanding Senior Faculty Award. He received the College of Engineering College Outstanding Teacher Award in He served as an Associate Editor of the IEEE TRANSACTIONS ON ANTENNA AND PROPAGATION from 1983 to 1986 and again from 1989 to He was an Associate Editor of the IEEE AP-S Newsletter from 1982 to Sean M. Duffy received the B.S. and M.S. degrees in electrical engineering from the University of Massachusetts, Amherst, in 1993 and 1995, respectively. He is currently working toward the Ph.D. degree at the same university. Since 1995, he has been with the Lincoln Laboratory, Massachusetts Institute of Technology, Lexington, MA. His current research interests are in the design and analysis of printed antennas and in the development of multilayered millimeter-wave packages.

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