IP2 and IP3 Nonlinearity Specifications for 3G/WCDMA Receivers Chris W. Liu, Morten Damgaard, Broadcom Corporation

Transcription

1 IP and IP Nonlinearity Specifications for G/WCDMA Receivers Chris W. Liu, Morten Damgaard, Broadcom Corporation The complete analysis of the nonlinearity requirements of a WCDMA direct conversion (Zero-IF) receiver is presented. This paper focuses on IP and IP requirements and covers all test cases defined in GPP, such as TX leakage, Intermodulation, Adjacent channel blocking, Narrow band blocking, In-band and Out of band blocking etc. The two tone IP and IP concept is further expanded into modulated signals. The formulas for computing IP and IP with modulated signals for different test cases are presented. In a G/WCDMA mobile handset, the direct conversion receiver (DCR) architecture is commonly used due to its simplicity and low cost, where inter-stage filters are completely eliminated. However the linearity requirement for a direct conversion receiver is very critical. It is essential to define system linearity requirements properly for the receiver to meet the performance requirements. In FDD mode, the transmitter and receiver are continuously working at the same time. The transmitted signal leaks into the receiver due to limited TX to RX isolation of the duplexer and it is possibly the strongest interferer for the receiver in the handset and poses the most stringent linearity requirement for the receiver. I. GPP TEST CASES A number of test cases are specified in GPP for a WCDMA receiver, and each test case has different test conditions. Therefore nonlinearity performance requirements are needed of the receiver in each case. The related test cases are summarized below in Table A-F and are used through out the following discussion. A. Sensitivity requirement Test Conditions Band DPCH_Ec <REFÎor> I II III V VIII TX power: 4 dbm B. Adjacent channel blocking Test Conditions Parameter Case Case DPCH_Ec <REFSENS> + 4 db <REFSENS> + 4 db Îor <REFÎor> + 4 db <REFÎor> + 4 db I oac -5-5 F uw +5 or or -5 Tx power: 0 dbm C. Intermodulation (wideband and Narrowband) Wideband Narrow band Parameter Test Conditions Band II, V Band III, VIII DPCH_Ec <REFSENS> + db <REFSENS> + 0 db Îor <REFÎor> + db <REFÎor> + 0 db I ouw (CW) I ouw mean power (modulated) F uw (offse ± 0 ±.5 ±.6 F uw (offse ± 0 ± 5.9 ± 6.0 Tx power: 0 dbm D. Out of band blocker (OOB) Parameter Unit Freq range Freq range Freq range DPCH_Ec dbm/.84 <REFSENS>+ <REFSENS>+ <REFSENS>+ db db db Î or dbm/.84 <REFÎ or > + db <REFÎ or > + db <REFÎ or > + db I blocking (CW) dbm F uw 050<f < <f 050 < f 05 MHz (Band I) 85<f <0 0 f <55 55 f<750 Tx power: 0 dbm E. Narrow band blocking (NBB) Test Conditions Parameter Unit Band II, V Band III, VIII DPCH_Ec dbm/.84 <REFSENS> + 0 db Îor dbm/.84 <REFÎor> + 0 db I blocking dbm F uw offset MHz ±.7 ±.8 Tx power: 0 dbm F. In band blocking (IBB) Parameter Unit Test Conditions DPCH_Ec dbm/.84 <REFSENS> + db Îor dbm/.84 <REFÎor> + db I blocking dbm F uw offset MHz ± 0 ± 5 Tx power: 0 dbm The above test cases require the receiver to meet certain BER performance with these defined interferers. For RF system design, the requirements have to be interpreted into RF parameters such as Noise Figure (NF), Compression point (PdB), Second Order Intercept Point (IP), Third Order Intercept Point ( IP), etc. II. SECOND ORDER NONLINEARITY The second order nonlinearity of the receiver will square the modulated blocker signal, such as the TX leakage signal, producing DC and low frequency components which fall into the receive band of the direct conversion receiver. The AM (amplitude modulated) signal is demodulated into the RX channel with twice bandwidth of the original interferers. Moreover, a strong blocking signal will also intermodulate due to second order nonlinearity with the TX leakage signal to create a TX image which can fall into the band. A. Mathematic Formula of IIP In general, the output signal of a nonlinear system can be described as follows: V o = a V i + a V i + a V i + K where V o is the output and V i is the input voltage. If using a traditional two tone signal as input V i, we have: Vi = A cos( ω + A cos( ω The second order component of the output is: Vo = avi = aa (cos ( ω + cos ( ω + cos( ωcos( ω = aa ( + cos(ω + cos(ω + cos(( ω + ω ) + cos(( ω ω ) It shows that the second order IM products are created at three frequencies: DC, f+f and f-f. In terms of power level, IM products are distributed against total IM power

2 as: 50% (- db) at DC 5%(-6 db) at f+f 5% (-6 db) at f-f The IM of low frequency is considered only in this paper since it is the one falling into band. The power level of the IM product at f-f is 5% of the total IM power which is 6 db below the total IM power. So the power level of IM at low frequency (f-f) can be expressed as: P ( dbm) = P IIP db () im in 6 the modulated downlink blocking signal is injected into the receiver with offset frequency at +/- 5MHz. By using method similar to the one discussed above, with an adjacent blocker signal (Test Model ), the corrected formula for IIP with a WCDMA downlink signal is: P im _ adj T 87 = P IIP. db () 00 B. IIP with WCDMA TX uplink leakage If the signal is AM modulated, such as TX leakage signal, the tone formula above can not be fully applied. The difference between the formula and tone signal is called correction factor [4]. The following section derives the correction factor for a WCDMA signal using ADS simulation. The ADS simulation Model is shown in Figure. dbm(s0) m m freq=.000mhz dbm(s0)= freq, MHz N_Tones N DF DF Figure Low Frequency Output of Two tone simulation for IP GPP UL RF Source GPPFDD_RF_Uplink RF_Signal_Source N_Tones N_CWblocker SummerRF S FcChange F QAM_Demod Q dbm(s0) ResBW -00 SplitterRF S6 SplitterRF S5 MultiplierRF M GainRF G SummerRF S7 Figure ADS Bench for IP Simulation SpectrumAnalyzerResBW S0 The two tone test case and the case of single tone plus WCDMA uplink signal test case are simulated. The coefficients of the second order product in the model is set to 0. for simplicity since only the difference between using the two tone measurement and the TX uplink modulated signal is needed. The two tone simulation results are shown in Figure and the simulation results with TX uplink signal are shown in Figure. With the TX uplink signal, since it is modulated, the low frequency IM products are measured by integrating the power from KHz to.0mhz in the frequency domain. The difference of IM between the two tone and modulated signal is 9.7 db. The IP two-tone formula for TX leakage case with correction factor is as follows: Pim _ txleakage = Pin IIP 5. 7dB () C. IIP with In-band blocking One of GPP test cases is adjacent blocker test, in which freq, MHz IM_modulated_Power Figure Low frequency Output of WCDMA uplink simulation for IP D. IIP with Out of Band Blocker (OOB) In OOB test cases, depending on the blocker frequency, the IM products of the OOB blocking signal and TX leakage signal may fall into the RX band. The power of IM can be calculate using formula below. Pim _ OOB = PCW + Ptx IIP (4) III. THIRD ORDER NONLINEARITY For third order nonlinearity, the GPP intermodulation test case defines the IP requirements of the receiver. However the blockers such as adjacent channel blockers, narrow band blockers, out of band blockers, etc either leak into the RX channel, or cross-modulate, or intermodulate with the TX leakage signal, the distortion products falling

3 into wanted channel. The receiver needs to have good linearity performance under all blocker conditions. A. IIP with Two Tone test Assuming the input signal as V i with two tone signals, Vi = A cos( ω + A cos( ω The output signal y( can be expressed as: y( = a Vi ( + a Vi ( + a Vi ( The third order intermodulation products are: y( = a A A cos( ω ) t cos( ω ) t + rdorder a A A cos( ω ) t cos( ω ) t a A A a A A = cos(ω ω) t + cos(ω ω ) t +K 4 4 The third order IM products are created at: f+f, f- f, f-f and f+f. In a down conversion receiver, only the low frequency products are interesting. The power of IM at low frequency is expressed in following classical formula: ) IM at f-f: P IIM ( dbm) = P + P * IIP (5) ) IM at f-f P IIM ( dbm) = P + P * IIP (6) If two tones are equal, the formula becomes: PIIM ( dbm) = Pin * IIP (7) B. IIP with Cross modulation The TX signal leakage signal can be cross modulated by a strong blocker such as a narrow band blocker, adjacent channel blocker, with the cross modulated signal falling into the receive channel. An example is illustrated in the Figure 4 and Figure 5. Figure 4 shows the output of a nonlinear circuit with two tone signal. Replacing one tone by a TX uplink leakage signal in Figure 5, cross modulation products are observed around the single tone at.99 GHz. For mathematical analysis, we could assume that the signal at the LNA input comprises of two interferers besides the wanted signal, namely a CW blocker signal and the TX leakage signal. Note that the TX leakage signal is amplitude modulated, while the blocker is not modulated in this analysis. Assuming the input signal x(, with blocker and TX leakage signal then we get: x ( t ) = A cos( ω t ) + A [ + m ( t )] cos( ω tx t ) where m( is the amplitude modulation having a fundamental frequency at WCDMA chip rate. When the signal inject into the non-linear RF circuit, receiver in this case, the output signal can be expressed as follows, considering only up to third order nonlinearity: + a + a dbm(spectrum_out_overall) dbm(spectrum_out_overall) freq, GHz Figure 4 Two tone simulation freq, GHz Figure 5 One tone is replaced by TX leakage signal y( = a x( + a = a A cos( ω + a A ( + m( cos( ω ( + m( A A A x ( + a x ( + a A cos ( ω + a A ( + m( cos ( ω A cos ( ω cos( ω cos( ω cos( ω + a A cos ( ω + a A A ( + m( cos ( ωt ) cos( ωtx + a A A ( + m( cos( ωt ) cos ( ωtx The term a A A ( + m( cos( ωt ) cos ( ωtx can be further expanded to: / a A A ( + m( cos( ω + / a A A ( + m( cos( ωcos(ω tx The term, / a A A ( + m( cos( ωt ), shows the blocker signal being modulated by the square of the amplitude of the TX leakage. ) Cross Modulation with adjacent channel blocker The straight forward thinking is to use the IIP formula with little modification. The quantity of the crossmodulation product can be evaluated with approximated formula: tx tx tx P = C + PTX + P IIP (8) crossmod factor adj

4 where, C factor is the correction factor that takes into account the difference between using two CW tone measurement and modulated signal measurement. The correction factor has been determined by comparing the difference between two tone case and the TX leakage case in simulation. It is found that C factor is around 7.4 db. The corrected formula for the adjacent channel blocker test case is: P = PTX + P IIP 7.4 (9) crossmod_adj adj ) Cross modulation with Narrow Band Blocker The narrow band blocker appears at +/-.7(.8) MHz, the offsets are much closer to the carrier compared the adjacent channel blocker case. The approximate formula is found to be: P = PTX + P IIP.4 (0) crossmod_nb nb C. Adjacent channel leakage - ACLR The receiver front end nonlinearity can create spectrum re-growth for an adjacent channel blocker, partially falling into the wanted band. The approximate formula to calculate the leakage power is reported in [] as: Paclr = ( Pin IIP) +. 6 PAR () where PAR is the Peak to Average ratio of the downlink signal. IV. SYSTEM SPECIFICATION In order to determine the IP and IP requirements, it is essential to consider all impairments including IM, IM products, which must be low enough for the receiver sensitivity degradation to be acceptable. In the following sections, IP and IP are derived. The following analysis are applied to Band-I. A. IP at TX frequency requirement at Sensitivity level ) Maximum allowed noise First of all, let s look at the maximum allowed noise floor without any interferers. From the GPP specification, the specified sensitivity is: Total input power: I or = dbm/.84mhz, Dedicated physical channel power: DPCH_E C =P sens =-7 dbm/.84 MHz. The processing gain of.kps reference channel: G p =5 db It has been reported that the required E b /N t, the ratio of signal energy bit to noise spectra density, shall be better than 5.dB to guarantee the Base Band (BB) modem can demodulate the WCDMA signal properly. In this paper, we assume E b /N t as 5. db and add. db implementation margin. Therefore the allowed total noise power at antenna input is: Eb Pn = Psens + Gp = = 99. 4dBm Nt Typically some margin is needed to cover process variation for production yield. By applying db margin, the total noise allowed at Antenna connector is: N tmax_ant =P n -=-00.4 dbm/.84mhz At RFIC input, taking front end loss into account, the maximum allowed interfere level is: N tmax = N tmax_ant -L front =-04. dbm where L front =.8 db is assumed including switch loss and duplexer RX insertion loss etc. The distortion products due to component nonlinearity shall be kept below a certain level so that the receiver sensitivity will not be degraded too much. The interferers are not only from second and third order nonlinearity but also from other sources, which include: PA noise at RX band RFIC phase noise at RX band TX leakage IIP TX leakage reciprocal mixing Additionally, if blockers are present, other concerns also include: Blocker reciprocal mixing Cross modulation Adjacent blocker ACLR Front end switch IM/IM products ) TX noise at RX band First all, the typical performance of the PA, RFIC and duplexer is studied in this case. Though the components from different manufacturers could perform differently, the typical performances used in this paper are summarized in Table, and, based on major manufacturers datasheets. TABLE PA NOISE AND GAIN Band Gain Rx Noise (dbm/hz) PA 7-40 PA 7-40 PA TABLE DUPLEXER PERFORMANCE RX- TX-ANT(dB) RX-TX (db) ANT(dB) Band Insertion loss Insertion RX noise TX RX loss attenuation attenuation attenuation TABLE RFIC PHASE NOISE AT RX BAND Band Rx Noise (dbc/hz) RFIC -65 RFIC -65 RFIC 5-70 It should be noted that for the sensitivity test case, the transmitter output power is defined as maximum output power, 4 dbm, at the antenna connector. In all other test cases, the output power at the antenna connector is defined as 0 dbm according GPP specifications. The output power of the PA can be calculated by add loss

5 between the PA output and the antenna connector. PA + FE + Dup out = P max loss where FE loss is the front end loss, including switch, PCB traces, matching loss. Dup loss is the duplexer TX insertion loss. The RFIC output power can be calculated based on PA out and the PA gain. It is around 0.5 dbm, P rfic =0.5 dbm, in this case. The noise present in the RX band at the LNA input due to PA phase noise is P pa_noise =PA noise -ISO tx-rx +0*Log 0 (BW)=-8.5 dbm where ISO tx-rx is duplexer isolation from TX to RX at RX band, and channel bandwidth BW=.84 MHz. And the TX noise at RX band due to RFIC is: P rfic-noise =P n_rfic +PA gain +0*Log 0 (BW)-ISO tx-rx =-0.5dBm Since the total noise due to TX transmitter is the summation of both, it becomes: P loss PAtx _ noise Pr fic _ noise tx _ noise = 0log0( ) = ) TX leakage reciprocal mixing The TX leakage power at the RFIC input is: Pleakage = PAout ISOtx _ rx = 5. 6dBm dbm If assuming the phase noise of the RX LO at the TX frequency is -57 dbc/hz, then the reciprocal mixing at the RX band is: P tx_reciprocal =-6.75 dbm 4) TX Leakage IIP First, let s make an assumption that the allowed sensitivity degradation as X db, and then we will have: 0 Ptx _ noise Ptx _ im Nt max Ptx _ reciprocal Nt max + X ( ) = 0 Arranging the equation above and expressing the above equation in db, we get: 0 log0(0 = N t max Ptx _ noise log0(0 X 0 X Ptx _ im 0 ) + 0 Ptx _ reciprocal 0 The term 0 0log0(0 ) is the factor that determines how much the impairments shall be below N tmax. For example if X=0. db, meaning 0. db sensitivity ) degradation is allowed, the total impairment power level must be.4 db below the maximum allowed noise floor N tmax. In general, it is good to keep IM due to TX leakage between to 6 db below N tmax so that the IM product degrades the sensitivity by 0. to 0. db only. In this paper 6 db is chosen for better performance. If so, Pim shall be less than -0 dbm that is 6 db below N tmax. From equation (), IIP at the TX frequency can be derived as: IIP ( dbm) = P T ( dbm) Pim _ tx 5.7dB = 48dBm Therefore the IIP requirement due to TX leakage for Band I at the RFIC input should be better than 48 dbm at the TX frequency. With IIP=48 dbm, adding all other impairments, the overall sensitivity degradation is less than 0.6 db. Further improvements in the PA phase noise and duplexer attenuation will continue to minimize the degradation. B. Adjacent channel blocker case The adjacent blocker appears at 5 MHz offset with a power level of -5 dbm at antenna connector. Since the input wanted signal is 4 db higher than sensitivity, it is reasonable to assume that the degradation due to the total interferer power shall be less than 4 db. However in practice, margin is needed. Note that with an adjacent channel blocker, the major contributor of in channel distortion is from ACLR, rather than IP. First of all, we need to determine the power leakage into the channel due to adjacent channel blocker, P aclr. To minimize the impact of ACLR, in this paper, P aclr is kept 6 db below the wanted signal. The allowed total noise floor N t, is 4 db higher than N tmax in this test case, therefore N t = -90. dbm, The allowed P aclr will be Paclr= N t -6=-06. dbm. From the equation (0), IIP can be calculated, IIP = ( Paclr Padj +.6 PAR) = 7. dbm We also need to check the IP requirements due to Cross modulation. Based on the cross modulation formula derived in previous section, equation (8), the cross-modulation product can be calculated: P = PTX + P IIP 7.4 = 08. 7dBm crossmod adj The cross modulation product with IP=-7. db is 8 db below the N t, not a problem. Then P im_adj due to adjacent channel blocker can be assumed 6 db below the N t as well. Then IIP at 5MHz adjacent channel, IIP_adj is obtained: IIP ( dbm) = PIIM _ adj + Padj.87dB = 5dBm

6 C. Adjacent channel case The blocker is present at 5 MHz offset with a power level of -5 dbm and the input wanted signal is 4 db higher than sensitivity. In this case, N t = -6. dbm, the allowed P aclr =N t -6=-79. dbm. The IIP and cross modulation product are calculated below. IIP = ( Paclr Padj +.6 PAR) = 6. dbm P = PTX + P IIP 9.67 = 0. 7dBm crossmod adj Similar to the adjacent channel blocker case, IIP can be derived for the adjacent channel blocking case : IIP ( dbm) = PIIM _ adj + Padj.87dB = dbm D. In band blocker 0MHz and 5 MHz The input wanted signal is db higher than sensitivity and the blocker at 0 MHz offset has a power level at -56 dbm, and -44 dbm at 5 MHz offset. Since the offset frequency is far from the carrier, there is no concern for ACLR in this case. Using equation (), keeping the IM product 6 db below the N t, IIP_adj at 0 MHz and 5 MHz are computed: IIP IIP 0 0 MHz = PIIM _0mhz + P mhz.87db = dbm 5 5 = MHz = PIIM _5mhz + P mhz.87db dbm E. Intermodulation For band I, the narrow band blocking test case does not apply, but for other bands. It does in the wideband intermodulation test case, one of the blocking signals is a CW tone at -46 dbm, while another blocker is a WCDMA modulated signal with power level of -46 dbm. The wanted signal is db higher than sensitivity power level. Keeping the IM product 6 db below the N t which is db above N tmax we can compute the IIP using equation (7). IIP = 0.5*( P P ) = dbm in IIM 5 F. Out of band blocker For Out of band OOB blocking cases, we need to consider both the second order and third order distortion from the switches that is between the duplexer and the antenna. The switches in the front end module generate second and third order nonlinearity components falling in RX band as well. Bases on manufacturer s datasheet, it is assumed in this paper that IM and IM of the front end switch module are: P sw_im =-0 dbm P sw_im =-0 dbm With the OOB test case, not all blocker frequencies produce distortion that falls into wanted band. For the second order nonlinearity, we will consider: F cw =F rx -F tx F cw =F rx +F tx For the third order nonlinearity, we will consider: F cw = F rx - F/ F cw = F rx -* F F cw =* F tx + F rx F cw = F / where F cw is the blocker frequency, F rx is the RX channel frequency, F tx is the TX frequency related to F rx, and F is the frequency separation between F rx and F tx. ) F rx -F tx The CW blocking signal is at very low frequency, which is the separation between RX and TX frequencies. The second order IM results in mixing of F tx + (F rx -F tx ) = F rx. As the CW blocker level is -5 dbm, and the duplexer attenuation at the CW blocker frequency is around 0 db for Band I, we get P cw =-45 dbm at the input of the RFIC and the TX leakage power level P tx =-9.6 dbm. Using equation (4), with P im 6 db below N t, which is db above noise floor of -04. dbm, IIP is calculated as: IIP = PCW + PTX PIM = 8dBm ) F tx +F rx In this case, the OOB signal is at high frequency. The mixing of (F tx +F rx )-F tx =F rx, falling in the RX band. The blocker level is -5 dbm, and the duplexer attenuation at the blocker frequency is around 0 db for Band I so that P cw =-45 dbm, and P tx =-9.6 dbm. Using equation (), with Pim 6 db below N t, we have IIP = PCW + PTX PIM = 8dBm ) F rx - F / The blocker power level actually varies depending on frequency offset according the GPP requirements. When F cw is between 050MHz and 075 MHz, the blocker level is -44dBm. As the duplexer attenuation at blocker frequency is around 5 db, Pcw is -49 dbm. IIP is calculated as: IIP = 0.5( PCW + PTX Pim) = 7dBm When Fcw is located between 05MHz to 050 MHz, the blocker power level is -0dBm. Assuming duplexer attenuation at blocker frequency is 5 db, P cw is -45 dbm. IIP is calculated as: IIP = 0.5( PCW + PTX Pim) = dbm When Fcw is between 05MHz and 05 MHz, the

7 blocker power level is -5dBm. With the duplexer attenuation at the blocker frequency of 0 db, P cw is -45 dbm. IIP is calculated as: 0.00 IP Wide band intermodulation IIP = 0.5(* PCW + PTX Pim) = dbm 4) F rx -* F The CW blocker frequency is located between 70MHz and 780 MHz. To allow P im =-7 dbm, which is 6 db below the max allowed noise floor N t, the blocker power level is -5dBm. Assuming duplexer attenuation at the blocker frequency is 8 db, P cw is -54 dbm. IIP is calculated as: IIP = 0.5( PCW + PTX Pim ) = dbm 5) Fw=*Ftx+Frx When the blocker frequency is between 5950MHz and 60 MHz, the CW blocker has a level of -5 dbm. As the duplexer attenuation at the blocker frequency is around 0 db, P cw is -45 dbm. With P im =-7 dbm, which is 6 db below max allowed noise floor N t, IIP is calculated as: IIP = 0.5( P + P P ) = dbm CW TX im 5 6) Fw= F / In this case the blocker is defined as -5 dbm CW signal, and the duplexer loss at this frequency is 0 dbm. Then P cw is -45 dbm. Similar to the previous case, to have P im 6 db below max allowed noise floor N t, IIP is calculated as: IIP = 0.5( PCW + PTX Pim ) = 5dBm V. MEASRURED RESULTS The IIP and IIP have been fully verified on a Broadcom WCDMA transceiver IC. The results of IIP at TX frequency and IIP wideband intermodulation are shown in Figure 6 and Figure 7, respectively. Both results show that the receiver has excellent linearity. IIP (dbm) IIP@TX frequency Frequency Figure 6 Measured Results of IIP at TX frequency IIP@TX IIP (dbm) Frequency (MHz) IP_IMD, positive Figure 7 Measured Results of IIP Wideband Intermodulation VI. CONCLUSION The system analysis of nonlinearity of a WCDMA receiver is discussed. The formulas for IP and IP for all test cases are presented and applied. The formulas are useful for system analysis. The assumed margins can be different in each test case. The formulas may be slightly different from other published results due to difference in simulation setup though. ACKNOWLEDGMENT The authors would like to thank colleagues at Broadcom Corporation for support, discussion and review. REFERENCES [] [] [] Harald Pretl, Linus Maurer, Werner Schelmbauer, Robert Weigel, LINEARITY CONSIDERATIONS OF W-CDMA FRONT-ENDS FOR UMTS, Microwave Symposium Digest., 000 IEEE MTT-SVolume, -6 June 000 Page(s):4-46 [4] Walid Y. Ali-Ahmad, Effective IM estimation for two-tone and. WCDMA modulated blockers in zero-if, in RFDesign, Apr, 004 [5] Ranta, T.; Ella, J.; Pohjonen, H., Antenna switch linearity requirements for GSM/WCDMA mobile phone front-ends, Wireless Technology, 005. The European Conference on -4 Oct. 005 Page(s): - 6 [6] Qizheng Gu, RF System Design of Transceivers for Wireless Communications, Springer. Chris W. Liu received his M.Sc from University of Montreal, Canada and B.Sc from Tianjin University, China. Chris has more than 7 years experience in the wireless communications. He is currently with Broadcom as Principal Systems Engineer. Before joining Broadcom, he holds various positions in RFMD, Intel, Focus microwaves etc. Chris can be reached by at: chrisl@broadcom.com. Morten Damgaard received his M.Sc. degree from University of Aalborg, Denmark in 989. Morten has 0 years of experience in the wireless and cellular industry doing cell phone RF design, semiconductor RF system design, and radio architecture design for GSM, EDGE, WCDMA, and multi-mode transceivers and radios. He currently works with Broadcom Corporation in Irvine, California as a Senior Manager of Cellular RF System Engineering. Before joining Broadcom Morten has worked for Axiom Microdevices, Skyworks Solutions, Conexant, Rockwell Semiconductors, and Dancall Telecom. Morten holds U.S. patents, mainly in the field of cellular radio design.

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