Design of buck-type current source inverter fed brushless DC motor drive and its application to position sensorless control with square-wave current

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1 Publihed in IET Electric Power Application Received on 4th January 2013 Revied on 17th February 2013 Accepted on 4th March 2013 ISSN Deign of buck-type current ource inverter fed bruhle DC motor drive and it application to poition enorle control with quare-wave current Hung-Chi Chen, Hung-He Huang Department of Electrical and Computer Engineering, National Chiao Tung Univerity, HinChu, Taiwan Abtract: Owing to the widely ued bruhle DC motor (BDCM) in high-efficiency application, many poition enorle control method baed on voltage ource inverter had been developed in the literature. Recently, current ource inverter (CSI) are receiving more and more attention becaue of their inherent hort-circuit protection characteritic. But no poition enorle control for buck-type CSI with quare-wave current had been found in the literature. In thi tudy, the buck-type CSI-fed BDCM drive i deigned and it application to the quare-current poition enorle control i firt propoed. The provided imulation and experimental reult verify the effectivene of the propoed CSI-baed poition enorle control. 1 Introduction Owing to the permanent-magnetic rotor field, bruhle DC motor (BDCM) poe higher efficiency than the popular induction motor. Therefore more and more BDCM are ued in variou high-efficiency variable-peed application, uch a fan motor [1, 2], compreor motor [3, 4], vehicle motor [5 7] and home application [8]. In the normal quare-wave current operation of BDCM, the dicrete rotor poition hould be monitored by poition enor to yield adequate current commutation [9]. The ix-witch voltage ource inverter (VSI) fed BDCM i plotted in Fig. 1 where the feedback poition ignal are ued to yntheie the witching ignal. The DC voltage amplitude can be equivalently varied via the pule width modulation (PWM) ratio by the peed controller. In order to avoid hort-circuit condition, both witche in the ame leg could not conduct in the ame time. A 120 conduction (ix-tep) method [3 9] and 150 conduction (12-tep) method [1, 2] are two commutation cheme for ix-witch VSI. Additionally, variou VSI topologie can alo be found in [10, 11]. Although VSI are widely ued in motor drive, reliability concern have been raied on the motor becaue of the high dv/dt that come from the PWM output voltage. Voltage urge reulting from thee rapid voltage tranition can caue motor inulation degradation, bearing failure becaue of eroion caued by the reulting haft leakage current, and unacceptable electromagnetic interference effect on the control circuit, a well a acoutic noie in the motor. In addition, the concern about the reliability of the electrolytic capacitor ha forced uer to ue cotly and bulky film capacitor [12]. Moreover, the poible hoot-through problem in VSI ha alway been a concern aociated with ytem reliability. The other topology i the current ource inverter (CSI). CSI ue an inductor a the energy torage component, and thu avoid many drawback of VSI. It alo ha an inherent advantage of the hoot-through hort-circuit protection capability, and no PWM voltage in the motor terminal. The bulky inductor ha longer lifetime than the capacitor. In fact, CSI topology had been widely ued in high-power application. The common thyritor-baed load commutated inverter (LCI) topology ha been reported for BDCM drive [10]. Four quadrant operation and current enorle control over mot of the operating peed range are good feature of thi topology. However, thyritor-baed CSI i uitable for the high-power utility-connected indutry application, but it i not uitable for kilowatt (kw)-level reidential application, uch a electric vehicle and variable-peed air conditioner. Recently, more and more reearche are focued on inulated gate bipolar tranitor (IGBT)-baed CSI drive for the automotive application [5 7]. The common IGBT-baed CSI for BDCM i plotted in Fig. 2 where the current ource i controlled by the peed controller. In order to provide current conducting path, at leat one of the three upper witche and at leat one of the three lower witche need to turn on at the ame time. The erie-connected diode need to withtand the negative voltage. Three AC capacitor are connected between the CSI and BDCM, and they provide the current flowing path during the current commutation [5 7]. For quare-wave current commutation in BDCM drive, the hall poition enor are required to provide the poition information. However, the hall enor may be faulted and may not be ued in high-temperature environment, uch a refrigerant compreor. 416 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

2 Fig. 1 Conventional VSI-fed BDCM Fig. 2 Ideal CSI-fed BDCM It i difficult to make clear comparion between CSI and VSI. For potential application, uch a electric vehicle, ytem reliability i an important iue. It i clear that the life time of the bulky electrolytic capacitor i horter than the bulky inductor. To further improve the reliability of the CSI-baed drive, poition enorle control i required to be able to ride through enor-fault condition. Owing to the concern of ytem reliability for BDCM operation, many poition enorle control method for VSI had been developed in the literature and mot of them are baed on the back-electromotive force (EMF) detecting method [1 5, 11, 13 16]. The method in [11] wa developed for a four-witch VSI topology and the other are developed for the common ix-witch VSI topology. The current-mode IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

3 enorle control in [13] hould be een a a VSI-baed method becaue that the terminal voltage had ignificant PWM voltage tranition. In VSI-baed enorle control method, only zero-croing point of the phae back-emf can be detected from the terminal voltage. Therefore delaying thee zero-croing point by 30 i required to obtain the correct commutation intant [11, 14 17], but it may introduce ome commutation error. In CSI, the line-to-line back-emf can be eaily ened from the terminal voltage, and thu the commutation intant can be obtained directly without the proceing of the additional 30 delay. However, no poition enorle control method for IGBT-baed CSI with quare-wave current commutation had been found in literature. In thi paper, the current ource in Fig. 2 i implemented by the buck converter and the buck-type CSI-fed BDCM drive i deigned. It i noted that the poition enorle control for buck-type CSI-fed BDCM i firt propoed. The provided imulation and experimental reult upport the validity of the propoed CSI-baed poition enorle control. 2 Current ource inverter fed BDCM For ervo application, the torque performance i mot important. However, in the promotion of reidential variable-peed product, uch a air conditioner, the cot iue i more critical to the performance iue. Thu, the manufacture take a imple trategy producing near-inuoid-emf BDCM to reduce the cot of motor. For the high-performance erie of the air conditioner, the high-cot inuoid-current drive i applied with the other high-efficiency ytem technique. However for the popular erie, the low-cot quare-current drive i utilied even when the yielded torque i not contant. Thu, the inuoidal rotor magnet are initially aumed. 2.1 Ideal current ource inverter The variable-peed drive BDCM fed by ideal current ource i plotted in Fig. 2, where the current ource i variable according to the peed controller output I d. According to the winding ditribution in Fig. 1, the flux linkage of three phae winding can be aumed a φ a = Φ MAX in θ e, φ b = Φ MAX in(θ e 2π/3) and φ c = Φ MAX in(θ e +2π/3), where Φ MAX i the maximum flux linkage. Thu, the induced voltage for each phae winding can be expreed a e a = dw a dt e b = dw b dt e c = dw c dt = F MAX v e co u e = P 2 F MAXv r co P 2 u r = P 2 F MAXv r co P 2 u r 2p 3 = P ( 2 F MAXv r co P 2 u r + 2p ) 3 (1) where P i the pole number, ω e i the ynchronou frequency in rad/, and θ r =2θ e /P i the rotor poition. When the poition θ e i between 0 and π/3 (0 θ e < π/3), both witche T 1 and T 2 are conducting and the current I d flow through the a-phae and c-phae winding I d = i a = i c. The winding current i a, i b and i c are in phae with the back-emf voltage e a, e b and e c, repectively. The voltage e i denoted by (ee (2)) where k i an integer and K E i the voltage gain K E = PΦ MAX /2. It i clear that the voltage e i a periodic waveform and it average value i e = 3 3KE v r /p. Since i a + i b + i c =0 in Y-connected winding, the neutral voltage v n can be repreented in term of the terminal voltage v n = v a + v b + v c 3 When the witch T 3 i conducting, the terminal voltage v b i equal to the bu voltage v b = V bu, and when the witch T 6 i conducting, the terminal voltage v b mut be zero. When both witche T 3 and T 6 are not conducting, b-phae winding i floating and the terminal voltage v b i equal to the um of the neutral voltage plu the induced b-phae voltage v b = v n + e b. Therefore when the poition θ e i located between 0 and π/3 (0 θ e < π/3), both witche T 1 and T 2 are conducting. The terminal voltage v a can be expreed a v a = V bu = e a e c + 2I d R e and the terminal voltage v c i zero. From (3), the neutral voltage terminal i v n =(V bu + v b )/3 and thu, the floating b-phae voltage v b i cloed to v b = V bu e b e a e c e b e b e c 3KE v r in u e when 0 u e, p/3 The illutrated waveform are plotted in Fig. 3. Three poition ignal H a, H b and H c are ued a the commutation ignal to obtain the BDCM peed ω r and generate ix witching ignal by the common ix-tep conduction cheme. 2.2 Buck-type current ource inverter In addition, there i alway one upper witch and one lower witch at the ame time in common ix-tep commutation and thu, the BDCM can be modelled a a erie-connected circuit with an inductance 2L, a reitance 2R and the voltage e a hown in Fig. 4, where L and R are the winding inductance and the winding reitance, repectively. L B i the output inductance of the buck converter. In practice, the current ource can be implemented by a buck converter a plotted in Fig. 4 becaue of the output inductor L B in the buck converter. The bu current I d i (3) (4) e a e c = 3KE v r co (u e p/6), when 2kp u e, 2kp + p/3 e b e c = 3KE v r co (u e p/2), when 2kp + p/3 u e, 2kp + 2p/3 e e = b e a = 3KE v r co (u e 5p/6), when 2kp + 2p/3 u e, (2k + 1)p e c e a = 3KE v r co (u e 7p/6), when (2k + 1)p u e, 2kp + 4p/3 e c e b = 3KE v r co (u e 3p/2), when 2kp + 4p/3 u e, 2kp + 5p/3 e a e b = 3KE v r co (u e 11p/6), when 2kp + 5p/3 u e, (2k + 2)p (2) 418 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

4 From Fig. 4, the voltage drop V drop acro the inductance (L B +2L ) and the reitance 2R i the difference between the diode voltage V oi and the equivalent voltage e. Thu, the tranfer function between the voltage drop V drop and the yielded bu current I d can be expreed a I d () V drop () = 1 (6) (L B + 2L ) + 2R From (6), the equivalent block diagram of Fig. 4 can be plotted in Fig. 5. The current controller G c () i the proportional and integral controller and it can be expreed a G c () = k cp + k ci (7) To deign the controller parameter, the ratio of the proportional gain k cp and the integral gain k ci i et to be k cp k ci = ˆV tri V in L B + 2L 2R (8) Fig. 3 Illutrated waveform for ideal CSI-fed BDCM Then, the tranfer function between the yielded bu current I d and the bu current command I d can be obtained by I d () I d () = ( k ) ci/2r + k ci /2R (9) Equation (9) behave like a low-pa filter with the cut-off frequency f cc = k ci /(4πR ) in hertz (Hz). Since the frequency of the equivalent voltage e i ix time the electrical frequency in Hz correponding to the BDCM peed ω r in revolution per minute (rpm). The cut-off frequency f cc in Hz hould be carefully elected between one tenth the PWM frequency ( f tri /10) and ix time the electrical frequency of the maximum BDCM peed 6 P 2 v r, MAX 60 = Pv r, MAX 20 f cc f tri 10 (10) where ω r,max i the maximum operating peed of BDCM in rpm. 2.3 Terminal voltage of CSI-fed BDCM Fig. 4 Buck-type CSI with equivalent circuit of BDCM ened and a current loop i deigned for regulating the bu current I d to the current reference Id. The gate ignal G TB of the controllable witch T B i obtained from the comparion of the controller output v cont and the fixed-frequency awtooth ignal v tri with a fixed amplitude ˆV tri. The duty ratio d B of the controllable witch T B can be expreed a d B = v cont / ˆV tri and thu, the average diode voltage kv oi l T within the witching period T become When both T 1 and T 2 are conducting (i.e. 0 θ e < π/3), the bu current I d flow through T 1 and T 2. It how that both a- and c-phae winding can be een a the excited phae and the b-phae winding i een a the floating phae. The equivalent circuit with turning on and turning off the witch T B are plotted in Fig. 6a and b, repectively. With kv oi l T = v cont ˆV tri V in (5) Fig. 5 Equivalent block diagram for the circuit in Fig. 4 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

5 Fig. 6 Circuit and the current flowing path a With turning on T B b With turning off T B conideration of the conducting tate of the witch T B, the terminal voltage v a and v b may be expreed a the following two equation, repectively buck-type CSI-fed BDCM are plotted in Fig. 7. The ripple in the bu voltage V bu i fixed and equal to 2L V in /(L B +2L ). 2L v a = G TB V in e a + e c 2I d R + e L B + 2L a e c + 2I d R 0 u e p 3 (11) L v b = G TB V in e a + e c 2I d R + e L B + 2L b e c + I d R 0 u e p 3 (12) where the witching ignal G TB i { G TB = 1, 0, when v cont. v tri and the witch T B turn on when v cont, v tri and the witch T B turn off (13) By neglecting the voltage drop acro the equivalent reitance R, the terminal voltage in (11) and (12) can be implified to be 2L v a G TB V L in + e B L L B + 2L a e B L B + 2L c L B + 2L 0 u e p 3 (14) L L v b G TB V in e L B + 2L a + e L B + 2L b e B + L c L B + 2L L 0 u e p 3 (15) From (14) and (15), it i clear that the witching ignal G TB contribute to the voltage ripple in the terminal voltage. From (14), the voltage ripple 2L V in /(L B +2L ) in the terminal voltage v a i double the ripple L V in /(L B +2L ) in the terminal voltage v b. The illutrated waveform for Fig. 7 Illutrated waveform for buck-type CSI-fed BDCM 420 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

6 3 Poition enorle control for buck-type CSI-fed BDCM The propoed poition enorle control for buck-type CSI-fed BDCM i plotted in Fig. 8, where a buck converter i connected in front of the common CSI inverter. There are even witche T 1 T 6 and T B in the buck-type CSI. The gate ignal of witch T B i generated from the comparion of the current controller output v cont and the awtooth ignal v tri. The other gate ignal are obtained from three commutation ignal S ab, S bc and S ca. Their generating rule are a follow G T1 = S ab S ca (16) G T2 = S bc S ca (17) G T3 = S bc S ab (18) G T4 = S ca S ab (19) G T5 = S ca S bc (20) G T6 = S ab S bc (21) The low-pa filter circuit compoed of the capacitance C 2 and the reitance R 1 are ued to ene the three terminal voltage and to generate three filtered ignal v a, v b and v c. Then, the three ignal S ab, S bc and S ca are obtained by comparing two of the three filtered ignal v a, v b and v c. In order to attenuate the PWM voltage ripple acro the terminal voltage, the cut-frequency f LP =(R 1 + R 2 )/ (CR 1 R 2 ) of the low-pa filter i elected between the witching frequency f tri in Hz and ten time the electrical frequency correponding to the maximum BDCM peed ω r, MAX. f tri. f LP 10 P 2 v r, MAX 60 = Pv r, MAX 12 (22) Aume that the PWM voltage ripple are filtered out without introducing phae delay by the low-pa filter. From (14) and (15), the filtered ignal v a and v b can be implified to be v a = R 2 L B e R 1 + R 2 L B + 2L a e c 0 u e p 3 v b = R 2 L B R e R 1 + R 2 L B + 2L b e c + 2 L R 1 + R 2 L B + 2L 2e b e a e c 0 u e p 3 (23) (24) By electing the buck inductance much larger than the winding inductance L B L, the econd term in (24) can be neglected and the filtered ignal v b of the floating phae can be expreed a v b R 2 L B e R 1 + R 2 L B + 2L b e c = Av e b e c 0 u e p (25) 3 where A v = R 2 /(R 1 + R 2 ) i the gain factor. According to the variou conducting tate, the expreion of the filtered ignal are tabulated in Table 1, and they are alo illutrated in Fig. 7. It i clear that the commutating intant occur at the croing point of the three filtered Fig. 8 Propoed poition enorle control for buck-type CSI-fed BDCM IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

7 Table 1 Filtered ignal in variou conducting tate v a v b v c T 1, T 2 A v (e a e c ) A v (e b e c ) 0 exciting phae floating phae exciting phae T 2, T 3 A v (e a e c ) A v (e b e c ) 0 floating phae exciting phae exciting phae T 3, T 4 0 A v (e b e a ) A v (e c e a ) exciting phae exciting phae floating phae T 4, T 5 0 A v (e b e a ) A v (e c e a ) exciting phae floating phae exciting phae T 5, T 6 A v (e a e b ) 0 A v (e c e b ) floating phae exciting phae exciting phae T 6, T 1 A v (e a e b ) 0 A v (e c e b ) exciting phae exciting phae floating phae ignal. Therefore the ignal S ab, S bc and S ca can be ued to generate the witching ignal in (16) (21). By uing the excluive (XOR) operator, the ignal S i generated from three commutation ignal S ab, S bc and S ca by S = S ab S bc S ca (26) where i the common XOR operator. Then, the period t S (in ) between the riing/falling edge of the combined ignal S i counted and the BDCM peed ω r in rpm i calculated by v r = 1 t = (27) P P t Three AC capacitor C 1 are connected acro the BDCM terminal to provide flowing path for the commutation current. However, the commutation current flow through the capacitor and may make the terminal voltage either maller than zero or larger than the bu voltage V bu. So, each witch i connected with a diode in erie to withtand the negative voltage becaue of the commutation current. When both T 4 and T 5 are conducting (i.e. π θ e <4π/3), the bu current I d flowing through T 4 and T 5, and the terminal voltage i zero v a = 0 a hown in Fig. 9a. Since the bulky inductance i elected L B L, the bu current can be een a a current ource in Fig. 9. After turning on the witch T 6 and turning off the witch T 4, the commutation current i CC may flow through the capacitor and lift the terminal voltage v a until the commutation current i CC decay to zero. The teady-tate equivalent circuit during turning on both T 5 and T 6 (i.e. 4π/3 θ e <5π/3) i plotted in Fig. 9b. From Fig. 3, the induced voltage of phae a and b i cloed to each other e a e b at the commutation intant θ e =4π/3, and thu, the Kirchhoff voltage law (KVL) equation for the commutation current path can be expreed a di 2L CC (t) + 2R dt i CC (t) + 1 i C CC (t)dt 0 (28) 1 The commutation current i CC ha initial value I d at the beginning of the current commutation, and the initial energy E tored in the inductance L can be expreed a E 1 2 L I 2 d (29) Aume quarter energy E i tranferred to the capacitance C 1, and then diipated by the reitance. The peak terminal voltage during the commutation intant θ e =4π/3 would be v pk I d L 4C 1 Therefore the capacitance C 1 hould be elected a (30) C 1. L Id 2 2 (31) 4 v limit where v limit i the maximum voltage with conideration of the diode blocking voltage. When permanent magnet ynchronou motor (PMSM) i tandtill, the back-emf are zero and the propoed enorle control need tarting trategy [18]. The propoed tarting trategy plotted in Fig. 10 i divided into contant current mode (CCM) and contant peed mode (CSM). In CCM, the bu current command i given by I d = I 1 and the current commutate with an increaing frequency until the rotor run at the contant peed ω 1 in rpm. After a given time t 1, the operating mode change from CCM to CSM. Fig. 9 Steady-tate current flowing path a When both T 4 and T 5 are conducting b When both T 5 and T 6 are conducting 422 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

8 Table 2 Simulated parameter of buck-type CSI and BDCM BDCM tator reitance/phae R = 0.3 Ω BDCM tator inductance/phae L = 1.7 mh BDCM pole number P =8 BDCM maximum peed ω r,max = 2000 rpm BDCM voltage gain K E = 75 V/krpm buck input voltage V in = 300 V buck inductor L B =20mH buck witching frequency f tri = 10 khz CSI output capacitor C 1 = μf Fig. 10 Operating mode during tart-up In CSM, the bu current command I d decreae linearly, but the current commutate with fixed frequency. At the ame time, the rotor peed ω r i calculated and checked if the difference between the peed ω 1 and ω r i near zero. Once the peed difference ω 1 ω r i maller than 30 rpm, the operation mode will change from CSM to the enorle run mode (SRM). In SRM, the current commutate according to the commutation ignal S ab, S bc and S ca, and the bu current command I d i obtained from the peed controller. 4 Simulation reult In thi ection, ome imulation reult are provided and the imulated parameter are hown in Table 2. A eight-pole BDCM with the reitance R = 0.3 Ω and the inductance L = 1.7 mh i ued, and it maximum operating peed i ω r,max = 2000 rpm (i.e Hz in electrical frequency). According to (22) and the parameter in Table 2, the cut-off Fig. 11 Simulated waveform during the tarting proce IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

9 frequency f LP mut be within the range khz f LP = R 1 + R 2 R 1 R 2 C 2 10 khz (32) After chooing the gain factor A v = 0.07 in (25) and the cut-off frequency f LP = 3 khz from (32), the parameter for the terminal voltage ening circuit are obtained by R 1 = 130 kω, R 2 =7.5kΩ and C 2 =0.047μF. Since the blocking voltage of the diode i 600 V and the maximum amplitude of the quare-wave current i 5 A, the CSI AC capacitor i elected with C 1 = μf from (31). A hown in Fig. 8, two identical BDCM are ued and their haft are coupled together to become a motor-generator et. One BDCM i connected to CSI, and the other i connected to the Y-connected reitor R g. Therefore both BDCM have the ame motor peed ω r = 2000 rpm and the ame BDCM induced voltage e a, e b and e c. The waveform during the tarting proce are plotted in Fig. 11 and the ued parameter are I 1 = 1 A and ω 1 = 400 rpm. BDCM i uccefully changed from CCM and CSM to SRM. The imulated waveform of the peed command v r = 2000 rpm with load reitor R g = 100 Ω and R g = 33.3 Ω are plotted in Fig. 12a and b, repectively. The buck current I d i well regulated and the yielded motor current i a i in phae with the induced voltage e a. Owing to CSI, the riing time and the falling time of the quare-wave current are relatively mall. But the peak of the bu voltage V bu increae with the yielded bu current I d. Fig. 13 Waveform during the tarting proce From Fig. 12, the terminal voltage v a may change rapidly becaue of the commutating current i CC illutrated in Fig. 9. In Fig. 12b, the peak of the terminal voltage v a may be 400 Fig. 12 Simulated reult for the propoed poition enorle control a ω r = 2000 rpm and R g = 100 Ω b ω r = 2000 rpm and R g = 33.3 Ω. Fig. 14 Experimental reult for the propoed poition enorle control a ω r = 500 rpm and R g = 100 Ω b ω r = 500 rpm and R g = 33.3 Ω 424 IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

10 The waveform during the tarting proce are plotted in Fig. 13. After 1.5, BDCM uccefully enter into SRM. The experimental waveform of the peed command v r = 500 rpm with load reitor R g = 100 Ω and 33.3 Ω are plotted in Fig. 14a and b, repectively. BDCM run tably at 500 rpm with variou load. From (14) and (15), the ripple of the terminal voltage during the exciting and the floating phae can be calculated to be 43.6 and 21.8 V, repectively, which meet the obervation from the experimental waveform. From Fig. 14b, the peak voltage appearing in the terminal voltage v a are near 200 and 100 V, which are maller than the diode blocking voltage 600 V. Thu, the voltage pule would not damage to the witch. The experimental waveform of the peed command v r = 2000 rpm with load reitor R g = 100 Ω and 33.3 Ω are plotted in Fig. 15a and b, repectively, which how that the propoed poition enorle control for BDCM run tably at 2000 rpm with variou load. From Fig. 15b, the peak voltage appear in the terminal voltage v a are near 600 and 400 V, which are maller than the diode blocking voltage 600 V. Thu, the voltage pule would not damage to the CSI circuit. From Fig. 14 and 15, the voltage ripple in the terminal voltage are fixed, which alo demontrate the derived (14) and (15). In (14) and (15), the profile of the voltage ripple becaue of the witching ignal G TB i fixed regardle of the BDCM peed ω r. 6 Concluion In thi paper, the behaviour of the buck-type CSI ha been tudied and the deign rule for the buck inductor and the output AC capacitor have been provided. Additionally, the poition enorle control method with quare-wave current for buck-type CSI-fed BDCM ha been propoed. The provided imulation and experimental reult how that the propoed poition enorle control work tably. 7 Reference Fig. 15 Experimental reult for the propoed poition enorle control a ω r = 2000 rpm and R g = 100 Ω b ω r = 2000 rpm and R g = 33.3 Ω and 100 V. It mean that the witch voltage may be negative. Since the controllable emiconductor device are not able to withtand the negative voltage, the diode are connected to the witche in erie to provide the bipolar withtanding ability. 5 Experimental reult The propoed poition enorle control i implemented in a FPGA-baed ytem. The nominal parameter are the ame a thoe in Table 2. Owing to no A/D and no D/A function in commercial FPGA XC3S200 chip, an external A/D converter i ued to ene the current input and ome D/A converter are ued to how the control variable in the cope. 1 Lelke, A., Krotch, J., De Doncker, R.W.: Low-noie external rotor BLDC motor for fan application. Proc. IAS, 2002, pp Wang, C.M., Wang, S.J., Lin, S.K., Lin, H.Y.: A novel twelve-tep enorle drive cheme for a bruhle DC motor, IEEE Tran. Magn., 2007, 43, (6), pp Chen, H.C., Chang, Y.C., Huang, C.K.: Practical enorle control for inverter-fed BDCM compreor, IET Electr. Power Appl., 2007, 1, (1), pp Lee, K.W., Kim, D.K., Kim, B.T., Kwon, B.I.: A novel tarting method of the urface permanent-magnet BLDC motor without poition enor for reciprocating compreor, IEEE Tran. Ind. Appl., 2008, 44, (1), pp Shao, J.: An improved microcontroller-baed enorle bruhle DC (BLDC) motor drive for automotive application, IEEE Tran. Ind. Appl., 2006, 42, (5), pp Wu, Z., Su, G.-J.: High-performance permanent magnet machine drive for electric vehicle application uing a current ource inverter. Annual Conf. IEEE Indutrial Electronic Society (IECON), November 2008, p Tang, L., Su, G.J.: Boot mode tet of a current-ource-inverter-fed permanent magnet ynchronou motor drive for automotive application, Control Model. Power Electron. (COMPEL), 2010, pp Park, J.W., Hwang, S.H., Kim, J.M.: Senorle control of bruhle DC motor with torque contant etimation for home application, IEEE Tran. Ind. Appl., 2012, 48, (2), pp Han, Q., Samoylenko, N., Jatkevich, J.: Average-value modeling of bruhle DC motor with 120 voltage ource inverter, IEEE Tran. Energy Conver., 2008, 23, (2), pp IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp

11 10 Singh, B., Singh, S.: Single-phae power factor controller topologie for permanent magnet bruhle DC motor drive, IET Power Electron., 2010, 3, (2), pp Lin, C.T., Huang, C.W., Liu, C.W.: Poition enorle control for four-witch three-phae bruhle DC motor drive, IEEE Tran. Power Electron., 2008, 23, (1), pp Gu, B., Nam, K.: A DC-link capacitor minimization method through direct capacitor current control, IEEE Tran. Ind. Appl., 2006, 42, (2), pp Chen, H.C., Liaw, C.M.: Current-mode control of enorle BDCM drive with intelligent commutation tuning, IEEE Tran. Power Electron., 2002, 17, (5), pp Su, G.J., Mckeever, W.: Low-cot enorle control of bruhle DC motor with improved peed range, IEEE Tran. Power Electron., 2004, 19, (2), pp Kim, D.K., Lee, K.W., Kwon, B.I.: Commutation torque ripple reduction in a poition enorle bruhle DC motor drive, IEEE Tran. Power Electron., 2006, 21, (6), pp Lai, Y.S., Lin, Y.K.: Novel back-emf detection technique of bruhle DC motor drive for wide range control without uing current and poition enor, IEEE Tran. Power Electron., 2008, 23, (2), pp Damodharan, P., Vaudevan, K.: Senorle bruhle DC motor drive baed on the zero-croing detection of back electromotive force (EMF) from the line voltage difference, IEEE Tran. Energy Conver., 2010, 25, (3), pp Wang, Z., Lu, K., Blaabjerg, F.: A imple tart-up trategy baed on current regulation for back-emf-baed enorle control of PMSM, IEEE Tran. Power Electron., 2012, 27, (8), pp IET Electr. Power Appl., 2013, Vol. 7, I. 5, pp