ADVANCE MITSUBISHI ELECTRIC CONTENTS. High Frequency and Optical Devices. Technical Reports. Sep / Vol Overview...

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2 Sep. 29 / Vol. 127 MITSUBISHI ELECTRIC ADVANCE High Frequeny and Optial Devies Editorial-Chief Kiyoshi Takakuwa Editorial Advisors Chisato Kobayashi Kanae Ishida Makoto Egashira Koji Yasui Hiroaki Kawahi Masayuki Masuda Akio Toda Kiyoji Kawai Tetsuji Ishikawa Taizo Kittaka Keiji Hatanaka Itsuo Seki Kazufumi Tanegashima Kazumasa Mitsunaga Vol. 127 Feature Artiles Editor Junihiro Yamashita Editorial Inquiries Makoto Egashira Corporate Total Produtivity Management & Environmental Programs Fax Produt Inquiries Hiroaki Seki (p2-4) Mobile Phone Devies Marketing Set. Tel: Seki.Hiroaki@j.MitsubishiEletri.o.jp Kazuhiko Sato (p5-16) High Frequeny Devie Marketing Set. Tel: Sato.Kazuhiko@aj.MitsubishiEletri.o.jp CONTENTS Tehnial Reports Overview...1 by Kazuo Hayashi A 2.4-V Low-Referene-Voltage Operation HBT-MMIC Power Amplifier Module for CDMA Appliations...2 by Takao Moriwaki and Kazuya Yamamoto HBT High Power Amplifier Modules for WiMAX CPE Appliations...5 by Hitoshi Kurusu and Toshio Okuda Mirowave Triple Tuned Wideband VCO...8 by Masaomi Tsuru and Ryoji Hayashi 6 W Output Power C-Band High-Effiieny Broadband GaN-HEMT...11 by Yoshitsugu Yamamoto and Koji Yamanaka Breakdown Voltage Enhanement in AlGaN Channel Transistors...14 by Takuma Nanjo and Muneyoshi Suita High Sensitivity 2.5/1 Gbps InAlAs Avalanhe Photodiodes...17 by Eitaro Ishimura and Eiji Yagyu 43-Gbps EAM-LD Module / PD Module...2 by Norio Okada Jun Morita (p17-22) Optial Communiation Devies Marketing Set. Tel: Morita.Jun@ea.MitsubishiEletri.o.jp Mitsubishi Eletri Advane is published on line quarterly (in Marh, June, September, and Deember) by Mitsubishi Eletri Corporation. Copyright 29 by Mitsubishi Eletri Corporation; all rights reserved. Printed in Japan.

3 Overview Author: Kazuo Hayashi* The Future of High Frequeny and Optial Devies Compound semiondutor-based high frequeny and optial devies play an important role in improving the apability of information and ommuniation systems, for whih market demand is growing. Mobile ommuniation devies require not only a high bit rate but also multi-funtionality, better performane and low power onsumption. Thus, high frequeny power amplifiers for transmitters are required to have low distortion, broad- or multi-band properties, multi-funtionality, and low drive voltage. High-output power amplifiers urrently used for satellite ommuniations are one of vauum tube alled a traveling wave tube amplifier (TWTA), and therefore ompat, lightweight, and long servie life equipment based on ompound semiondutor is desired. In the field of fixed line teleommuniations, optial ommuniation systems, whih were mainly for business use, are now widely used in the home. In response to this trend, in addition to ost-effetive devies, high-speed and high-sensitivity devies are being introdued for the metro and trunk line systems, whih are handling inreasing amounts of data ommuniation traffi. Mitsubishi Eletri has developed and ommerialized high frequeny and optial devies utilizing design and manufaturing tehnologies aquired over the years. This issue presents our reent ativities geared to these markets and tehnial trends. *High Frequeny & Optial Devie Works Mitsubishi Eletri ADVANCE September 29 1

4 A 2.4-V Low-Referene-Voltage Operation HBT-MMIC Power Amplifier Module for CDMA Appliations Authors: Takao Moriwaki* and Kazuya Yamamoto* 1. Introdution This paper desribes the iruit design and measurement results of the prototype HBT MMIC power amplifier module whih operates with a low referene voltage of 2.4 V. The module has been developed for 9-MHz band CDMA handset appliations in response to CDMA s needs for low-voltage operation. 2. Bakground Reently, gallium arsenide based heterojuntion bipolar transistor (GaAs-based HBT) amplifiers have been widely used for ode division multiple aess (CDMA) mobile handsets. To extend the battery life of suh handsets, the HBT amplifier and peripheral iruits should onsume minimal power. We aimed to develop a low-voltage bias iruit, whih an lower the battery s end-of-disharge voltage in the handset and thus effetively prolong the battery life. However, this is not easy to ahieve beause the referene voltage for an emitter follower iruit generally needs to be at least twie the base-emitter voltage (Vbe: +1.3 V). This paper presents our new power amplifier for CDMA mobile handset appliations that operates with a low referene voltage of 2.4 V. It has been made possible by: (i) dividing the RF signal power stage into two AC-oupled bloks and providing one blok with a voltage and urrent drive bias, and the other a urrent Vref IN Vb Input mathing Current drive path Tr1 Tr2 Fig. 1 Ciruit onfiguration used for the low referene voltage operation V Output mathing drive bias; and (ii) adding a diode linearizer to the power stage with the voltage and urrent drive bias to prevent deterioration of adjaent hannel leakage power at low temperatures. 3. Ciruit Design Figure 1 shows a iruit blok diagram of the power stage with a bias iruit for low referene voltage operation. The design target for the referene voltage was set to 2.5 V or lower so that the Si-LDO (low voltage drop out regulator) an operate even at the battery s end-of-disharge voltage of 2.7 V. If the referene voltage is set to 2.6 V or lower, it is not suffiiently greater than twie the base-emitter voltage, and thus satisfatory operation is not expeted with the emitter follower iruit used as a bias iruit for the amplifier. Therefore, a urrent-drive bias iruit is added to the power stage so that the desired idle urrent an be supplied. In addition, in order to redue the amount of hanges in the power gain and the phase with respet to the output power and to obtain smooth and monotoni gain and phase harateristis, the power stage for amplifying RF signals is divided into two AC-oupled bloks, Tr1 and Tr2, as shown in Fig. 1. Smooth gain and phase harateristis of the power stage are important for minimizing the distortion of the amplifier, Gain [db], phase [deg] 2 1 Gain -1 phase Divided power stage -2 Undivided power stage -3 f=9mhz V=3.5V, -4 Vref=Vb=2.4V Pout [dbm] Fig. 2 Calulation results (Input-output harateristis): Comparison of divided and undivided power stages *High Frequeny & Optial Devie Works 2

5 whih is a key parameter for CDMA appliations. As depited in Fig. 1, one of the divided power stage bloks, Tr1, is supplied with a base bias urrent through the emitter follower iruit as well as the resistor in the added urrent-drive supply iruit, resulting in high output operation. The other power stage blok, Tr2, is supplied with a base bias urrent through the urrent-drive iruit only, to ensure the desired amount of idle urrent. It was onfirmed by simulation that the power stage with two kinds of bias supply iruit an ahieve output harateristis having only small hanges in the power gain or phase by adjusting the individual amount of supply from eah urrent-drive base-bias iruit. Figure 2 ompares the output harateristis alulated for the iruit onfigurations with divided power stage (Fig. 1) and undivided power stage (Fig. 3). In these simulations, the frequeny was set to 9 MHz; and as the bias onditions, the referene voltage and power supply voltage for the bias iruit were set to 2.4 V, and the olletor voltage of Tr1 and Tr2 was set to 3.5 V. Figure 2 learly shows that the divided power stage provides smooth gain and phase harateristis with respet to the output power. However, omputer simulation of the temperature dependeny indiated that at low temperatures below ºC, the iruit in Fig. 1 exhibited onave-shaped gain harateristis and onvex-shaped phase harateristis similar to those of the iruit without a divided power stage. These harateristis at low temperatures were aused by the HBT s built-in base-emitter voltage that beame higher than that at normal temperature, resulting in a ondition similar to that when lowering the referene voltage at normal temperature. To prevent these hanges in the power gain and phase at low temperatures, a linearizer onsisting of a diode and a resistor is added to the power stage as shown in Fig. 4. Figure 5 ompares the simulated output harateristis of the power stage at 1ºC with and without a diode linearizer. It was onfirmed that the hanges in the gain and phase were effetively suppressed by adding a diode linearizer even at low temperatures. As desribed above, it was onfirmed by simulation that the HBT amplifier fabriated by the onventional HBT proess an operate with a low referene voltage over a wide temperature range by means of the divided power stage with different bias supply methods and a diode linearizer. 4. Evaluation Results Based on the simulation results desribed above, a prototype HBT monolithi mirowave integrated iruit (MMIC) power amplifier module was fabriated and evaluated for the 9-MHz band J/WCDMA. A blok diagram of the power amplifier module is shown in Fig. 6. The amplifier module was evaluated using 9-MHz band JCDMA (1S-95B) and WCDMA (3GPP-R99) modulation signals, and with the bias onditions of a power stage olletor voltage of 3.5 V, and a power supply voltage for the bias iruit inluding referene voltage of 2.4 V. The input-output harateristis with the JCDMA modulation signal are shown in Fig. 7. Results with the JCDMA modulation obtained at an output power (Pout) of 27.5 dbm were: power gain (Gp) = 26.5 db, power added effiieny (PAE) = 4%, and adjaent hannel power ratio (ACPR) = 5 db. These results adequately satisfy the key output harateristis required for JCDMA transmitter power amplifiers. Measurement results with the WCDMA modulation signal were Pout = 28. dbm, Gp = 26.7 db, PAE = 42%, and adjaent hannel leakage ratio (ACLR) = 42 db. Figure 8 ompares the input-output harateristis measured with and without the diode linearizer at the ase temperature (T) = 1ºC. Vb Vb Current drive path Vref V Vref V Output mathing Output mathing IN Input mathing IN Input mathing Fig. 3 Ciruit onfiguration with an undivided power stage Fig. 4 Ciruit onfiguration with a diode linearizer Mitsubishi Eletri ADVANCE September 29 3

6 2 Gain V1 V2 Gain [db], phase [deg] 1 phase -1 Without linearizer -2 With linearizer f=9mhz -3 V=3.5V, Vref=Vb=2.4V Pout [dbm] Fig. 5 Calulation results (Input-output harateristis): Comparison of harateristis with and without diode linearizer (Temperature: 1ºC) IN Input math Vref(=Vb) GaAs die Inter Output -stage math math Bias Bias For 1st st For 2nd st Fig. 6 Blok diagram of amplifier OUT Gp [db], PAE [%] f=912mhz (IS-95B) V=3.5V, Vref=Vb=2.4V Gp PAE Pout [dbm] ACPR Fig. 7 Evaluation results (Input-output harateristis): JCDMA modulation (T = 3ºC) ACPR [db] Gp [db] Gp Without linearizer With linearizer ACLR -2-4 f=912mhz(is-95b) -6 V=3.5V, Vref=Vb=2.4V Pout [dbm] Fig. 8 Evaluation results (Input-output harateristis): Comparison of harateristis with and without diode linearizer (Temperature: 1ºC) ACPR [db] Gp [db], PAE [%] V=3.5V,Vref=Vb=2.4V PAE Gp T=-1 C T=3 C T=9 C ACPR Frequeny [MHz] Fig. 9 Frequeny harateristis with JCDMA modulation (Pout = 27.5 dbm [T = 1ºC, 3ºC, 9ºC]) ACPR [db] As predited by the previously desribed simulation results, if the diode linearizer is not built on the hip, the power gain signifiantly hanges, whih is assoiated with a signifiant deterioration of ACPR. In ontrast, when a diode linearizer is added, flat gain harateristis are obtained with respet to the output power and ACPR is improved. The above evaluation results learly show, as predited by the simulation, that our bias iruit and power stage onfiguration enable a lower referene voltage to be used over a wide temperature range (1). We will improve the iruit design and strive to develop new amplifiers, in response to the needs for low power supply voltage. Referene (1) K. Yamamoto et al., A CDMA InGaP/GaAs-HBT MMIC Power Amplifier Module Operating with a Low Referene Voltage of 2.4 V, IEEE J. SSC, Vol. 42, No. 6, pp , June 27. 4

7 HBT High Power Amplifier Modules for WiMAX CPE Appliations Authors: Hitoshi Kurusu* and Toshio Okuda* 1. Introdution For WiMAX power amplifier appliations, we have developed three models of heterojuntion bipolar transistor (HBT) power amplifier modules, MGFS36EXXXX, whih all have high output power and low distortion with a 5-Ω input/output interfae. These modules have an average output power of 27 dbm and a gain of 33 db in both the 2.3- and 2.5-GHz bands with an error vetor magnitude (EVM) of 2.5%, and an average output power of 25 dbm and a gain of 3 db in the 3.5-GHz band with an EVM of 2.5%. 2. Bakground Worldwide interoperability for mirowave aess (WiMAX) overs mid- and long-distane areas and enables high-speed ommuniations, and thus is a promising tehnology for the next generation of high-speed wireless ommuniation systems. Although different ountries have alloated various frequeny bands, various systems are being developed onurrently for the orresponding frequeny bands. In South Korea, for example, ommerial servie is already available. This ommuniation system uses an orthogonal frequeny division multiplexing (OFDM) modulation signal, whih has an extremely high peak output power relative to the average output power, and thus needs a power amplifier having high saturation power and low distortion harateristis. In addition, WiMAX systems for ustomer premises equipment (CPE) appliations will be installed in PC ards and mobile handsets, and thus need to be small, low ost and operable from a single power supply. In response to these needs, Mitsubishi Eletri has developed three models of high-output-power and low-distortion HBT power amplifier modules, MGFS36EXXXX, for the 2.3-GHz, 2.5-GHz and 3.5-GHz bands, using an indium gallium phosphide/gallium arsenide (InGaP/GaAs) HBT proess that is well proven for fabriating mobile phone amplifiers. On a pakage measuring only 4.5-mm square, the amplifier module integrates a bias iruit inluding amplifiers and olletor power supply line, as well as funtions unique to the WiMAX power amplifier suh as a step attenuator and output power detetor iruit using Mitsubishi s proprietary AC-oupled stak type base-olletor diode swith. In addition, the 5-Ω input/output interfaes eliminate the need for an external mathing iruit and help make the overall system smaller and heaper. The developed modules offer high output power and low distortion harateristis: the modules have an average output power of 27 dbm and a gain of 33 db with an EVM of 2.5% in the 2.3-GHz band (MGFS36E2325) and 2.5-GHz band (MGFS36E2527), and an average output power of 25 dbm and a gain of 3 db with an EVM of 2.5% in the 3.5-GHz band (MGFS36E3436). 3. Configuration of Power Amplifier Module Figure 1 shows the shemati onfiguration of the newly developed power amplifier modules for the 2.3-, 2.5- and 3.5-GHz bands (1). To ahieve a gain of 3 db or greater, a three-stage amplifier is employed; and a /2-dB step attenuator with its ontrol iruit and an output power detetor iruit are also integrated. The bias urrent to eah iruit is shut down by turning off the referene voltage (Vref). 3.1 Step attenuator The step attenuator onsists of Mitsubishi s proprietary AC-oupled stak type base-olletor diode swith (ACCS-DSW), whih has a high permissible transmission power even when operating with a low bias urrent (2). Figure 2 shows its iruit diagram. Under the same bias urrent onditions, the ACCS-DSW an improve the permissible transmission power harateristis by at least 6 db ompared to onventional diode swithes. The distortion harateristis have been further improved by plaing a diode linearizer at the input terminal, whih ompensates for the gain deviation at a high input power level. This type of step attenuator makes it possible to insert an attenuator between the first and seond stages of the amplifier, and to prevent a hange in the input return loss and deterioration in the noise figure (NF) harateristis while the attenuator is being turned on and off. The ontrol iruit for the step attenuator is onfigured to allow a omplementary signal to be output aording to the ontrol signal (/3 V) input to the external ontrol terminal (V ont ). A power supply swith transistor is added to the ontrol iruit so that no urrent is onsumed by the iruit while the Vref voltage is turned off. *High Frequeny & Optial Devie Works Mitsubishi Eletri ADVANCE September 29 5

8 Step attenuator Linearizer RF input RF output Attenuator ontrol voltage (Vont = /3 V) IN OUT Colletor voltage (V = 6 V) Output power detetor voltage (Vdet) Bias iruit Output power detetor iruit Referene voltage (Vref = 2.85 V) Fig. 1 Configuration of power amplifier module 3.2 Output power detetor iruit A diode detetor iruit is used as the output power detetion iruit. The iruit is designed to provide a hange in the detetor output voltage (Vdet) of at least 1 V when the output power level hanges from 7 to 27 dbm. This detetor iruit is biased diretly by the Vref terminal via a resistor, and thus an also be shut down by turning off the Vref voltage. 4. Basi Charateristis of Power Amplifier Module Figure 3 shows a photograph of the WiMAX power amplifier module. A small module size of mm 3 ommon to all the 2.3-, 2.5- and 3.5-GHz bands has been Fig. 3 Photograph of module Fig. 2 Ciruit diagram of step attenuator ahieved. The InGaP/GaAs HBT proess was used to fabriate the monolithi mirowave integrated iruit (MMIC) power amplifier, whih is mounted on the module. Figure 4 shows the measured frequeny harateristis of the 2.5- and 3.5-GHz band modules with the attenuation being swithed on or off and a power supply voltage of 6 V. The 2.5-GHz band power amplifier module exhibited a linear gain of at least 28 db and an attenuation of 19 db at frequenies between 2.5 and 2.7 GHz, and maintained an input return loss of 1 db or greater regardless of whether the attenuation was turned on or off. The 2.3-GHz band module exhibited nearly the same harateristis. Meanwhile, the 3.5-GHz band power amplifier module also exhibited a linear gain of at least 27 db and an attenuation of 21 db at frequenies between 3.4 and 3.6 GHz, and showed similar input return loss harateristis as the 2.5-GHz band module, maintaining an input return loss of 1 db or greater regardless of whether the attenuation was turned on or off. Figure 5 shows the measured output-power dependenies of the gain, effiieny and EVM of the 2.5- and 3.5-GHz band modules with the attenuation being turned on or off and using the IEEE ompliant 64 QAM-OFDM modulation signal. The power 4 3 S21 Vont=V 4 3 S21 Vont=V S21, S11 (db) Vont=3V Vont=3V S21, S11 (db) Vont=V Vont=3V -2-3 V=6V Vref=2.85V S11 Vont=V -2-3 S11 Vont=3V V=6V Vref=2.85V Frequeny (GHz) Frequeny (GHz) (a) 2.5-GHz band module (b) 3.5-GHz band module Fig. 4 Frequeny harateristis when attenuation is turned on or off Attenuation off (Vont = V), Attenuation on (Vont = 3V) 6

9 supply voltage was 6 V and the referene voltage (Vref) was 2.85 V. When the attenuation was turned off, the 2.5-GHz band module exhibited a power gain of 33 db, an EVM of 2.5%, and an effiieny of 12% with an output power of 27 dbm at frequenies between 2.5 and 2.7 GHz. The 2.3-GHz band module has nearly the same harateristis. The 3.5-GHz band module exhibited a power gain of 3 db, an EVM of 2.5%, and an effiieny of 11% with an output power of 25 dbm at frequenies between 3.4 and 3.6 GHz. The 2.3-, 2.5- and 3.5-GHz band modules have all ahieved high power gain and low distortion harateristis. When the attenuation was turned on, the output power that satisfied EVM = 2.5% was a suffiiently high value of 12 dbm or greater with all three power module models. Figure 6 shows the measured output voltage from the detetor iruit. Both the 2.5- and 3.5-GHz band modules have ahieved a suffiient hange in the detetor output voltage (Vdet) of 1 V or greater when the output power level is hanged from 7 to 27 dbm. The basi harateristis of these three models of power amplifier modules (MGFS36EXXXX) are summarized in Table 1. These modules will be useful for developing small, low-ost handsets. We will ontinue to develop produts with even higher output and effiieny. Referenes (1) Miyo Miyashita et al., Fully Integrated HBT MMIC Power Amplifier Modules for Use in 2.5/3.5-GHz-Band WiMAX Appliations, IEICE Tehnial Report, ED (2) K. Yamamoto et al., A /2 db Step Linearized Attenuator with GaAs-HBT Compatible, AC-oupled, Stak Type Base-olletor Diode Swithes, IEEE International Mirowave Symposium Digest, pp , 26. Power Gain, Gp (db) V=6V Vref=2.85V fo=2.6ghz Gp Vont=3V Vont=V Gp PAE EVM, PAE (%) Power Gain, Gp (db) V=6V Vref=2.85V fo=3.5ghz Gp Vont=3V Vont=V Gp PAE EVM, PAE (%) EVM PAE EVM EVM PAE EVM Pout (dbm) (a) 2.5-GHz band module Pout (dbm) (b) 3.5-GHz band module Fig. 5 Large signal harateristis when attenuation is turned on or off Attenuation off (Vont = V), Attenuation on (Vont = 3 V) -5 Deteted voltage Vdet (V) V=6V Vref=2.85V fo=2.6ghz fo=3.5ghz Pout (dbm) Fig. 6 Measurement result of detetor output voltage Table 1 Basi harateristis of power amplifier modules for WiMAX CPE appliations Charateristis Measurement onditions MGFS36E2325 MGFS36E2527 MGFS36E3436A Operating frequeny V=6V GHz Power gain Vref=2.85V IEEE db Effiieny signal input % Output power (EVM = 2.5%) Unit dbm Input return loss db Detetor output voltage V Attenuation db Total olletor urrent ma Module size 4.5mm 4.5mm 1.mm Mitsubishi Eletri ADVANCE September 29 7

10 Mirowave Triple Tuned Wideband VCO Authors: Masaomi Tsuru* and Ryoji Hayashi* 1. Introdution In this paper a triple tuned voltage ontrolled osillator (VCO) is proposed for small, low-ost transeiver appliations. Analytial alulations showed that the proposed VCO enables a wider osillation bandwidth than the onventional double tuned type (1). A fabriated prototype exhibited a wide osillation bandwidth of 5.6 to 16.8 GHz (relative bandwidth: 1%) and onfirmed the effetiveness of the proposed onfiguration. 2. Configuration Figure 1 shows the basi equivalent iruit of the triple tuned VCO. The triple tuned VCO is a series feedbak osillator onsisting of one ative devie and three tuned iruits. In our onfiguration, a hetero-juntion bipolar transistor (HBT) having a low 1/f noise is used as the ative devie. The three tuned iruits onsist of a series-onneted variable apaitor and indutor. 3. Analysis The following setions desribe the onditions required to ahieve a wide osillation bandwidth and the relationship between the osillation bandwidth and iruit parameters. 3.1 Conditions for wideband osillation If the tuned iruits are assumed to be lossless, the real and imaginary parts of Z a are respetively expressed as the following equations: C 1 be g ω m X + X e X Re( ) = Cb Cb Z (1) a 2 2 α + β 1 ωc + be X e α g ωc Im 2 be ( Z ) = + a X 2 α β ωcb where 1 1 α = ωcbe + e ωcbe ωcb m ( X X ) e m X e β ( X X ) 1 (2) β = g (4) (3) The osillation onditions are expressed as the following equations: ( ) Re Z < (5) a ( ) X Im Z = (6) a + b Fig. 1 Small signal equivalent iruit of the triple tuned VCO In the basi onfiguration of series feedbak osillator, the tuned iruit onneted either to the base or to the olletor is indutive, whereas the one onneted to the emitter is apaitive. A iruit onsisting of an HBT and tuned iruits onneted to either emitter or olletor is onsidered to be an ative iruit. Z a is the input impedane of this ative iruit when looking toward the HBT s base terminal. The onditions for negative resistane are derived from equations (1) and (5) and expressed as follows: 1 X < (7) ωc b In the ase of the double tuned onfiguration, X is indutive and unontrollable, and thus equation (7) is not satisfied at high frequenies. In the ase of the triple tuned onfiguration, X an be ontrolled to be smaller than 1/ωC b. Consequently, the osillation bandwidth of the proposed triple tuned VCO is wider than that of the double tuned VCO. 3.2 Osillation bandwidth Sine the tuned iruit onneted to the base is assumed to be lossless, Re(Z a ) beomes zero in a stable osillation state. Therefore, in a stable osillation state, the value of g m beomes zero and the osillation angular frequeny ω is expressed by the following *Information Tehnology R&D Center 8

11 equation derived from equation (6): 1 X b 1 = 1 + X ω C be e X ω C b be k (9) b k j Now, the following parameters are defined: C >> 1 C C je min C C n C (1) be je max je min (8) (11) where C jemin is the minimum value of variable apaitane and C jemax is the maximum value of variable apaitane. In order to ontrol three varator diodes using a single voltage supply, the following equations are assumed to hold: C = C = C (12) j jb je In addition, sine X and X b are indutive and X e is apaitive, the following equation is also assumed to hold: L L L = L >1 bb k l (13) ee ee By solving equation (8), the osillation frequeny hange ratio is obtained as follows: ω ω max min χ n k j k k l k k l k l nk j k kl k k l k l where the following approximation is used. 2 ( k 1) l + 2 ( 2k k ) C k ( k + 2) l C k C 2 l l l + C be je + l l C 2 be k 2 C 2 je (14) je ( k 1) + k ( k 2) (15) be Figure 2 shows the alulation results of the osillation bandwidth versus apaitane ratio. Equation (14) was used for the alulations. It is understood from Fig. 2 that an osillation bandwidth of 1% is obtained when k j is.5 and n is approx. 14. Figure 2 also indiates that a smaller k j provides a wider osillation bandwidth. Therefore, from equation (1), a transistor that has a large C b, and hene a larger emitter size, is better suited for a wideband VCO. 4. Prototype Fabriation Results Figure 3 shows a onfiguration shemati of the fabriated triple tuned VCO. The ative devie is an InGaP/GaAs HBT with an emitter size of 12 μm 2 and ut-off frequeny of 31.6 GHz. The apaitane ratio of the varator diode is approx with a reverse voltage between and +16 V. All varator diodes are ontrolled using a single voltage soure. Figure 4 shows a photograph of the fabriated triple tuned VCO. Beause of high frequeny operation and to prevent errors in fabriating the VCO, the HBT and varator diodes are mounted on an alumina substrate using flip-hip tehnology. The size of the VCO is 8.6 mm 6.8 mm. Figure 5 shows the measured osillation frequeny of the fabriated VCO. Bias voltages of the HBT are V = 3 V, V b =.3 V, and V e = 1 V. Osillation Bandwidth (%) K j = n Fig. 2 Calulation results of the osillation bandwidth versus apaitane ratio of the tuned apaitane: k = 28, kl = 6 Ve Vb Vnt InGaP/ GaAs HBT V Load Fig. 3 Configuration of the fabriated triple tuned VCO Mitsubishi Eletri ADVANCE September 29 9

12 Tuned Ciruit Tuned Ciruit Tuned Ciruit 8.6mm HBT Fig. 4 Photograph of the fabriated triple tuned VCO Osillation frequeny (GHz) Measurement Simulation 6.8mm Corresponding to the tuning voltage, V nt, of.35 V to +16 V, the osillation frequeny varied from 5.6 GHz to 16.8 GHz (relative bandwidth: 1%), the phase noise was 112. db/hz or lower at 1 MHz offset from the arrier, and the output power was 3.4 dbm +/ 2. db. The urrent onsumption was 76.1 ma or lower. The measurement results are in lose agreement with the alulation result, onfirming the effetiveness of this onfiguration. VCOs inorporating our tehnology are expeted to be used in a wide variety of appliations. Referene (1) K. Tajima, Y. Imai, Y. Kanagawa, and K. Itoh, A 5 to 1 GHz Low Spurious Triple Tuned Type PLL Synthesizer Driven by Frequeny Converted DDS Unit, IEEE MTT-S International Mirowave Symposium Digest, vol. 3, pp , Jun Tuning voltage (V) Fig. 5 Measurement results of the osillation frequeny of the fabriated VCO 1

13 6 W Output Power C-Band High-Effiieny Broadband GaN-HEMT Authors: Yoshitsugu Yamamoto* and Koji Yamanaka** The AlGaN/GaN high eletron mobility transistor (HEMT) has superior harateristis of high-voltage, high-power density, and high-frequeny operation, and is expeted to be used for next-generation high-power devies. To date, traveling wave tube amplifiers (TWTAs) have been widely used as high-power devies for C-band or higher frequeny appliations, but are now expeted to be replaed by GaN-HEMT thanks to its improved performane. We have reently demonstrated superior harateristis of GaN-HEMT with an output power of over 6 W and power added effiieny of over 5% with relative bandwidth of 1%, onfirming that GaN-HEMT is a promising high-power devie at C-band and higher frequeny. 1. Introdution High-frequeny devies are now desired to have, in addition to high power and high performane, improved harateristis for pratial appliations suh as small size, low power onsumption, high reliability and low ost. Reently, silion laterally diffused metal-oxide semiondutors (Si-LDMOSs) are widely used as high-output-power devies for mobile phone base stations and other appliations, establishing their presene in the L/S band field. Meanwhile, for C-band or higher frequeny appliations, gallium arsenide (GaAs) based field effet transistors (FETs) and TWTAs are widely used as high-power devies. However, the former has an upper limit of output power density beause of its low breakdown voltage, making it diffiult to fabriate high-power or broadband devies; and the latter has problems suh as large equipment size and short life time. Therefore, new devies that solve these problems need to be developed, suh as GaN-HEMT. Beause of its material properties, GaN-HEMT is apable of high-voltage and high-power-density operation, and thus small, highly effiient and wide-band devies are easily ahievable. Consequently, if TWTA-omparable effiieny is to be ahieved in addition to the good inherent reliability of solid-state devies, GaN-HEMT would even replae TWTA. 2. Challenges for High-frequeny Appliations Currently, TWTAs are widely used as C-band or higher frequeny and over 1-W lass high-power devies, and ahieve operating effiienies exeeding 7%. To verify the advantage of GaN-HEMT for high-frequeny operation, TWTA-omparable performane needs to be demonstrated. Although a C-band GaN-HEMT with an output power of over 1 W has already been reported (1), the devie had a low effiieny and needs to be improved. In addition, when the devie is used for a high-power ommuniations amplifier, it is also important to have a wide operation bandwidth and provide a stable high performane. With these bakgrounds, we have worked on improving the harateristis of GaN-HEMT devies and have developed a broadband internal mathing iruit, and demonstrated its high effiieny at C-band with a relative bandwidth of about 1%. 3. Improvement of Transistor Struture To enhane transistor performane, it is important to improve the drain effiieny and operational gain. We have already suessfully prevented the urrent ollapse phenomenon by forming a surfae passivation film using atalyti hemial vapor deposition (Cat-CVD) (2), improved the pulse I-V harateristis, and redued the on-resistane by reduing the ohmi ontat resistane by means of Si ion implantation (3). With these efforts, the drain effiieny of the transistor has been improved, but parasiti effets beome dominant at higher frequeny. Consequently, to improve the performane, we have developed a transistor having via holes to redue soure indutane. Considering heat management, the newly developed transistor is fabriated on a highly heat ondutive SiC substrate. However, SiC is a very hard material and diffiult to proess, and thus via holes an not be easily formed through thik SiC substrate. To overome this problem, we have developed new tehnologies to redue the thikness of the SiC substrate with high preision and to perform high-speed ething of via holes. Figure 1 shows an image of a via hole (8 μm di- Fig. 1 SEM image of via hole *High Frequeny & Optial Devie Works Mitsubishi Eletri ADVANCE September 29 11

14 ameter) observed by a sanning eletron mirosope (SEM). Before we fabriated the transistor with via holes, we onfirmed that good shape via holes as shown in the photograph ould be stably formed without residue. Then, we formed via holes on an atual transistor and onfirmed the improvement in the gain. Figure 2 shows the devie harateristis with and without via holes. The evaluated devie has a gate width of.6 mm, and operates at a C-band frequeny with a drain voltage of 5 V and a drain urrent of 3 ma/mm. Even without via holes, the transistor operates at high effiieny with a drain effiieny of 73%, whih is alulated from the power added effiieny (PAE) and the gain at the saturated output power level. By forming via holes, the parasiti indutane is redued, and both operational gain and maximum PAE are respetively improved by about 2 db and 4% at the saturated output power level. These effets are likely to be even greater above C-band frequenies, and to be extremely useful for further high frequeny appliations in the future. 4. Development of Broadband High-Effiieny Mathing Ciruit With regard to the improvement of transistor effiieny, we verified that the drain effiieny of the transistor itself was improved, and that the effiieny was boosted by improving the gain due to the redued parasiti effet by forming via holes. However, for use as a high-power amplifier, the effiieny needs to be improved further by optimizing the internal mathing iruit. In partiular, if higher harmonis are generated and output from the transistor, they will redue the effiieny. The loss in the mathing iruit itself also needs to be minimized. Consequently, we have worked on improving the effiieny by effetively refleting higher harmonis other than the fundamental wave by means of a higher harmoni refletion iruit (4) (5). Figure 3 shows a shemati diagram of the output mathing iruit that ahieves high effiieny over a wide bandwidth. A seond harmoni refletion iruit is diretly onneted to the transistor and reflets higher harmonis generated in the transistor. The refletion iruit uses parallel-onneted short and Output power [dbm], Gain [db] Output power With via hole Without via Gain Power added effiieny Input power [dbm] Fig. 2 Comparison of devie harateristis with and without via holes Power added effiieny [%] Fig. 3 Broadband high-effiieny output mathing iruit open-iruited stubs for effetive refletion within the iruit. With this onfiguration, seond harmoni waves generated in the transistor are refleted bak to the transistor with very little attenuation. In addition, by appropriately setting the design parameters of the shortand open-iruited stubs, the phase ondition for seond harmoni refletion is satisfied over a wide bandwidth. Figure 4 shows the evaluation results of the frequeny dependenies of the output power, drain effiieny, and power added effiieny of an internal mathing type amplifier using the above-desribed transistor and mathing iruit. The horizontal axis indiates the frequeny ratio with respet to the enter measurement frequeny in C-band, F. For this prototype evaluation, a transistor without via holes having a gate width of 16 mm was used, and the drain voltage was 4 V during the measurement. These results onfirm that the iruit shown in Fig. 3 effetively funtions as a broadband high-effiieny output mathing iruit, ahieving a drain effiieny of over 53% and an output power of over 6 W aross a relative bandwidth of 1% at C-band. These results are harateristis of the transistor without via holes, so we then fabriated a devie using a transistor with via holes. For this prototype, the internal mathing iruit was redesigned to be optimized for the transistor with via holes. Figure 5 shows the evaluation results of the frequeny dependenies of the output power and power added effiieny of a broadband high-effiieny amplifier using a transistor with via holes. The horizontal axis indiates the normalized frequeny with respet to the enter frequeny in C-band. During Output Power (dbm) Output power Drain effiieny Power added effiieny Fo Fo 1.75Fo 3 Frequeny (GHz) Fig. 4 Frequeny harateristis of broadband high-effiieny amplifier Drain effiieny, Power added effiieny (%) 12

15 these measurements the drain voltage was 4 V. This hart onfirms that the prototype has an output power of over 6 W and a power added effiieny of over 5% aross a relative bandwidth of 1% at C-band. Figure 6 ompares these results with the reported values of C-band high-effiieny high power amplifiers (HPAs) (6) - (12). The horizontal axis shows the output power and the vertial axis the drain effiieny. Our newly developed broadband HPAs exhibit high effiieny over a wide bandwidth at C-band and have ahieved the highest drain effiieny among broadband devies. These results indiate that GaN-HEMT devies offer exellent potential even at frequenies higher than C-band. Drain Effiieny (%) Output power (W), PAE (%) Output power PAE Normalized frequeny Fig. 5 Frequeny harateristis of broadband high-effiieny amplifier using a transistor with via holes GaAs (broadband) TWTA (broadband) GaN (broadband) with Via Hole GaN (broadband) without Via Hole GaN (broadband) 3 GaN (narrow band) Output Power (W) Fig. 6 Performane omparison of C-band high-effiieny amplifiers 5. Conlusion We have developed C-band high-power broadband amplifiers using AlGaN/GaN HEMT. We developed via hole forming tehnology to redue the parasiti indutane of the transistor, and suessfully demonstrated an improved operational gain. We ombined this improved transistor with our newly developed broadband high-effiieny mathing iruit to realize a high-frequeny, broadband power amplifier devie, whih has the world s highest power added effiieny of over 5% with an output power of over 6 W aross a relative bandwidth of 1% at C-band. These results indiate that GaN-HEMT devies are promising for operation at frequenies higher than C-band, with power and effiieny omparable with those of TWTA devies. Referenes (1) Yamanaka, K., et al.: S and C-Band Over 1 W GaN HEMT 1 Chip High Power Amplifiers with Cell Division Configuration, 25 European Gallium Arsenide and Other Semiondutor Appliation Symposium, pp (25) (2) Kamo, Y., et al.: A C-Band AlGaN/GaN HEMT with Cat-CVD SiN Passivation Developed for an Over 1W Operation, Mitsubishi Tehnial Report, 8, No. 5, (26) (3) Oishi, T., et al.: High Performane GaN Transistors with Ion Implantation Doping, Mitsubishi Tehnial Report, 79, No. 8, (25) (4) Yamanaka, K., et al.: C-band GaN HEMT Power Amplifier with 22 W Output Power, 27 IEEE MTT-S Int. Mirowave Symp. Dig. TH1A-2 (27) (5) Iyomasa, K., et al.: GaN HEMT 6 W Output Power Amplifier with Over 5% Effiieny at C-Band 15% Relative Bandwidth Using Combined Short and Open Ciruited Stubs, 27 IEEE MTT-S Int. Mirowave Symp. Dig. TH1A-3 (27) (6) Ui, N., et al.: A 1 W Class-E GaN HEMT with 75% Drain Effiieny at 2 GHz, Pro. 36th European Mirowave Conf., Manhester (26) (7) Colantonio, P., et al.: A C-band High-effiieny Seond-harmoni-tuned Hybrid Power Amplifier in GaN Tehnology, IEEE Trans. MTT-S., Vol. 54, No. 6, pp (26) (8) Otsuka, H., et al.: Over 65% Effiieny 3 MHz Bandwidth C-band Internally-mathed GaAs FET Designed with a Large-signal FET Model, 24 IEEE MTT-S Int. Mirowave Symp. Dig., pp (24) (9) Menninger, W. L., et al.: 7% Effiient Ku-Band and C-Band TWTs for Satellite Downlinks, IEEE Trans. Eletron Devies, pp (25) (1) Wu, Y.-F., et al.: 14-W GaN-based Mirowave Power Amplifiers, 2 IEEE MTT-S Int. Mirowave Symp. Dig., pp (2) (11) Chini, A., et al.: Power and Linearity Charateristis of Field-plate Proessed-gate AlGaN-GaN HEMTs, IEEE Eletron Devie Letters, pp (24) (12) Okamoto, Y., et al.: C-band Single-hip GaN-FET Power Amplifiers with 6-W Output Power, 25 IEEE MTT-S Int. Mirowave Symp., pp (25) Mitsubishi Eletri ADVANCE September 29 13

16 Breakdown Voltage Enhanement in AlGaN Channel Transistors Authors: Takuma Nanjo* and Muneyoshi Suita* 1. Introdution Gallium nitride (GaN) has a high eletri breakdown field, saturated eletron veloity and other superior harateristis. The high eletron mobility transistor (HEMT) based on aluminum gallium nitride (AlGaN)/GaN, whih uses GaN as a hannel layer, may have a high-density two-dimensional eletron gas formed at the heterojuntion interfae, and thus many exellent harateristis have been reported (1) - (5). Consequently, Al- GaN/GaN HEMT is expeted to be used as a high-power high-frequeny devie for next-generation information ommuniation system suh as satellite ommuniations and mobile ommuniation stations. The inreasing amount of information ommuniation is likely to ontinue growing, with data traffi aelerating suh as for high-speed wireless video ommuniations. Suh appliations will require even higher power devies. One effetive way of improving the output power of Al- GaN/GaN HEMT is to enhane the breakdown voltage of the hannel layer by using a material that has a higher eletri breakdown field than GaN. AlN has a band gap about twie that of GaN, hene about four times greater breakdown field. In addition, the saturated eletron veloity of AlN, whih affets the drain urrent, is nearly the same as that of GaN. Consequently, the use of Al-rih AlGaN for the hannel layer is expeted to enhane the breakdown voltage without reduing the drain urrent. In this onfiguration, however, the Al-rih AlGaN layer may signifiantly inrease the ohmi ontat resistane. Instead, for proper operation of HEMT that uses AlGaN as the hannel layer (AlGaN hannel HEMT), a new tehnology that redues the ontat resistane is needed. We have already demonstrated with a onventional GaN hannel HEMT that Si ion implantation doping effetively redues the ohmi ontat resistane (6). This time, we have fabriated ohmi eletrodes on an AlGaN hannel HEMT using an Si ion implantation doping tehnique, and demonstrated suessful transistor operation and a signifiantly higher breakdown voltage. This paper desribes the results. 2. Experimental Method Figure 1 shows a ross-setional shemati view of the newly fabriated AlGaN hannel HEMT. On a sapphire substrate, an AlN buffer layer was grown at high temperature, and then an unintentionally doped Al x Ga 1-x N hannel layer and an Al y Ga 1-y N barrier layer were onseutively grown by metal organi hemial vapor deposition (MOCVD) (7). In our experiments, two kinds of Al- GaN hannel strutures having different Al ompositions were used, namely Al.39 Ga.61 N / Al.16 Ga.84 N and Al.53 Ga.47 N / Al.38 Ga.62 N. For omparison, a onventional epitaxial substrate with GaN hannel struture (Al.18 Ga.82 N / GaN) was also fabriated. Fig. 1 A ross-setional shemati view of the fabriated AlGaN hannel HEMT The HEMT was fabriated by forming eah element in the following order: implanted ontat area, soure and drain eletrodes, devie isolation area, and gate eletrode. The implanted ontat area was formed by implanting 28 Si ions at the dose onentration of m 2 with the implantation energy of 5 kev, followed by ativation heat treatment using the rapid thermal annealing (RTA) method in a nitrogen atmosphere at 12ºC for 5 minutes. It should be noted that these ion implantation and ativation heat treatment proesses were performed with the semiondutor surfae overed by a SiN layer. In the next stage, the soure and drain eletrodes were formed by depositing Ti/Al using the eletron beam evaporation tehnique, followed by RTA treatment in a nitrogen atmosphere at 6ºC for 2 minutes. Then, the devie isolation area was formed by the multi-stage implantation of Zn ions (8). Finally, the gate eletrode was formed using Ni/Au that was deposited by the eletron beam evaporation tehnique. Note that no surfae passivation films onsisting of SiN x, for example, are formed. 3. Experimental Results 3.1 Epitaxial harateristis Figure 2 shows the depth profile of arrier onentration in the Al.38 Ga.62 N hannel struture determined by measuring the apaitane-voltage (C-V) harateristis using Shottky diodes formed on the epitaxial substrate having the Al.38 Ga.62 N hannel layer. At a depth of about 2 nm, whih nearly orresponds to the thikness of the barrier *Advaned Tehnology R&D Center 14

17 layer, the arrier onentration abruptly inreases, indiating that a two-dimensional eletron gas is formed in this area. Similar results were also obtained with the other two kinds of epitaxial substrates. By integrating these arrier onentrations in the depth diretion, the sheet arrier onentrations in the GaN hannel, Al.16 Ga.84 N hannel and Al.38 Ga.62 N hannel strutures were determined as 3.46, 3.29 and m 2, respetively. The arrier onentration at a suffiiently deep position of around 1 μm was suffiiently low at less than m 3. Carrier onentration (m -3 ) Al.53 Ga.47 N Al.38 Ga.62 N Depth (nm) Fig. 2 Carrier onentration-depth relationship in the epitaxial substrate having Al.38Ga.62N hannel struture 3.2 Ohmi harateristis Figure 3 shows the urrent-voltage (I-V) harateristis between the ohmi eletrodes, whih were formed on the two kinds of epitaxial substrates having different AlGaN hannel strutures with and without Si ion implantation doping. In either epitaxial substrate, if Si ions are not implanted, the resistane is extremely high and almost no urrent flows. In ontrast, if Si ions are implanted, the resistane is signifiantly redued and the urrent drastially inreases. Using the irular transmission line method (CTLM), the ontat resistane values were determined as and Ω m 2 on the Al.16 Ga.84 N hannel and Al.38 Ga.62 N hannel strutures, respetively. As desribed, Si ion implantation doping may be a very effetive method of forming ohmi ontats on an Al-rih Al y Ga 1-y N / Al x Ga 1-x N heterojuntion struture that has a wide band gap. 3.3 HEMT harateristis We used this Si ion implantation doping tehnique to fabriate the HEMTs and evaluated their harateristis. Figure 4 shows the drain urrent - drain voltage (I d -V d ) harateristis of Al.38 Ga.62 N hannel HEMT in the on-state. The measured HEMT has a gate length (L g ) of 1 μm, a gate width (W g ) of 1 μm, a soure-gate distane (L sg ) of 1 μm, and gate-drain distane (L gd ) of 2 μm. As shown in Fig. 4, this HEMT exhibited good pinh-off harateristis with the maximum drain urrent of 114 ma/mm. Also, the HEMTs fabriated on the other two kinds of epitaxial substrates similarly exhibited good pinh-off harateristis. Figures 5 (a) and (b) show the I d -V d harateristis of three kinds of HEMTs having different L gd values in the off-state, measured at a gate voltage (V g ) of 5 V. The L g, W g, and L sg values of these samples are the same as those shown in Fig. 4. Figure 5 (a) shows the harateristis of HEMTs with an L gd of 3 μm, whih are mainly used as high-frequeny devies suh as low-noise amplifiers. In omparison with the onventional type GaN hannel struture, the present devie struture very effetively enhanes the breakdown voltage with a relatively small inrease in the Al ontent suh as.16. The resulting breakdown voltage was 381 V with the Al.16 Ga.84 N hannel HEMT, and 463 V with the Al.38 Ga.62 N hannel HEMT. In ontrast, Fig. 5 (b) shows the harateristis of HEMTs with an L gd of 1 μm, whih are mainly used as high-power devies suh as high-power swithing devies. With this devie struture, a higher Al ontent effetively enhanes the breakdown voltage, resulting in an extremely high breakdown voltage of 1,65 V with the Al.38 Ga.62 N hannel HEMT. Figures 6 (a) and (b) respetively show the L gd dependenies of the breakdown voltage and maximum drain urrent of the fabriated three kinds of HEMTs. As shown in Fig. 6 (a), the maximum drain urrent of the AlGaN hannel HEMTs is almost independent of L gd, exept for a small derease when L gd is over 1 μm. In ontrast, the breakdown voltage is signifiantly enhaned as the Al ontent is inreased, and a longer L gd provides Current density (ma/mm) d = 4 μm With Si implantation Without Si implantation Al.38 Ga.62 N hannel Al.16 Ga.84 N hannel Drain urrent density (A/mm) L g = 1μm L gd = 2μm Al.38 Ga.62 N hannel Voltage (V) Fig. 3 I-V harateristis between the ohmi eletrodes formed on various AlGaN hannel epitaxial substrates Drain voltage (V) Fig. 4 Id-Vd harateristis of the on-state Al.38Ga.62N hannel HEMT Mitsubishi Eletri ADVANCE September 29 15

18 Drain urrent density (A/mm) (a) GaN hannel Al.16 Ga.84 N hannel Al.38 Ga.62 N hannel L gd = 3 μm L g = -5 V TECHNICAL REPORTS Drain urrent density (A/mm) (b) GaN hannel Al.16 Ga.84 N hannel Al.38 Ga.62 N hannel L gd = 1 μm L g = -5 V Drain voltage (V) Drain voltage (V) Fig. 5 Id-Vd harateristis of the off-state HEMT Drain urrent density (A/mm) (a) (b) GaN hannel Al.38 Ga.62 N hannel Al.16 Ga.84 N hannel Al.16 Ga.84 N hannel GaN hannel Al.38 Ga.62 N hannel Gate-drain distane (μm) Gate-drain distane (μm) Fig. 6 Lgd dependenies of maximum drain urrent and breakdown voltage of fabriated HEMTs Breakdown voltage (V) better results (Fig. 6 (b)). To the best of our knowledge, the ahieved breakdown voltage is the highest among HEMTs having omparable L gd lengths (3) - (5). It is striking that these results were obtained without using any eletri-field relaxation means suh as a field plate struture. In other words, it may be possible to enhane the breakdown voltage even further by using suh means. 4. Conlusion To enhane the breakdown voltage of HEMT, we hanged the hannel layer struture from the onventional GaN to AlGaN, and examined the harateristis of the resulting HEMTs. We ahieved a redution in the ohmi ontat resistane of the soure and drain eletrodes by using Si ion implantation doping, and suessfully demonstrated the operation of AlGaN hannel HEMT. We obtained very high breakdown voltages with the Al.38 Ga.62 N hannel HEMTs: 463 V with a gate-drain distane of 3 μm, and 165 V with 1 μm. To the best of our knowledge, these values are the highest ahieved by HEMTs having similar strutures. From these results, it is expeted that the AlGaN hannel HEMT proposed in this paper will inrease the output power of next-generation high-frequeny devies as well as high-power devies. 5. Aknowledgements The authors would like to thank Prof. Y. Aoyagi and Prof. M. Takeuhi of Ritsumeikan University. Referenes (1) Keller, S., et al.: Gallium Nitride Based High Power Heterojuntion Field Effet Transistors: Proess Development and Present Status at UCSB, IEEE Trans. Eletron Devies, 48, (21) (2) Kamo, Y., et al.: A C-Band AlGaN/GaN HEMT with Cat-CVD SiN Passivation Developed for an Over 1W Operation, Mitsubishi Tehnial Report, 8, No. 5, (26) (3) Kikkawa, T.: Highly Reliable 25 W GaN High Eletron Mobility Transistor Power Amplifier, Jpn. J. Appl. Phys. 44, (25) (4) C. S. Suh, et al.: High-Breakdown Enhanement-Mode AlGaN/GaN HEMTs with Integrated Slant Field-Plate, in IEDM Teh. Dig. (26) (5) Uemoto, Y., et al.: 83 V Bloking Voltage Al- GaN/GaN Power HFET with Thik Poly-AlN Passivation, in IEDM Teh. Dig. (27) (6) Oishi, T., et al.: High Performane GaN Transistors with Ion Implantation Doping, Mitsubishi Tehnial Report, 79, No. 8, (25) (7) Takeuhi, M., et al.: Al- and N-Polar AlN Layers Grown on C-plane Sapphire Substrates by Modified Flow-modulation MOCVD, J. Cryst. Growth, 35, (27) (8) Oishi, T., et al.: Highly Resistive GaN Layers Formed by Ion Implantation of Zn Along the C Axis, J. Appl. Phys., 94, (23) 16

19 High Sensitivity 2.5/1 Gbps InAlAs Avalanhe Photodiodes Authors: Eitaro Ishimura* and Eiji Yagyu** We have developed an InAlAs avalanhe photodiode (InAlAs-APD) using a low-noise InAlAs multipliation layer that multiplies the signal. Compared to onventional InP-APD, the noise level was almost halved, and a good minimum reeiver sensitivity of 29.9 dbm was ahieved at 9.95 Gbps. The newly developed high-sensitivity InAlAs-APD provides suffiient harateristis for 2.5-Gbps and 1-Gbps high-sensitivity transmitter appliations. 1. Struture of High-sensitivity Avalanhe Photodiodes (APD) In line with rising volumes of traffi on the Internet and other means of information ommuniation, the speed of trunk and metro systems in Japan and overseas is being inreased (from 2.5 Gbps to 1 Gbps) on the publi optial-fiber ommuniation networks. Efforts are also underway to inrease the distane between repeaters, and a transmission distane of 8 km at 1 Gbps is defined in the international standard ITU-T G.691 L64.2. Meanwhile, as fiber-to-the-home (FTTH) onnetions have spread, Gigabit/Ethernet Passive Optial Networks (G/E-PON) are being onstruted at a rapid pae both in Japan and overseas. For the reeivers used in these systems, a high-sensitivity avalanhe photodiode (APD) is indispensable. In response to the needs of these systems for a high-sensitivity light-reeiving element, we have reently developed new APDs for 1-Gbps (light-reeiving diameter of 2 μm) and 2.5-Gbps (5 μm) appliations. To ahieve a 2 db inrease in sensitivity (i.e., higher S/N ratio) ompared to previous devies, the multipliation layer that multiplies the signal was fabriated using InAlAs that generates only low exess noise during multipliation [1-5], while a new type of planar struture was adopted to improve the reliability [5]. Figure 1 shows the struture of the guardring-free planar type InAlAs-APD, whih has been developed for 2.5/1-Gbps high-sensitivity reeiver appliations. Eah layer was grown by moleular beam epitaxy. First, an n-type distributed Bragg refletor layer (DBR layer) was grown on the n-type InP substrate. Then, an unintentionally doped InAlAs multipliation layer, a p-inp field ontrol layer, an InGaAs light-absorbing layer with low arrier density, a arrier pile-up prevention layer onsisting of InAlGaAs/InAlAs, and an i-inp window layer were staked in this order. A p-type light-reeiving area was formed by the seletive diffusion of zin into the window layer. The outermost surfae was proteted by a SiN passivation film, whih also funtions as a non-refletive film. An anode eletrode was formed enirling the light-reeiving aperture (2 μm in diameter for 1 Gbps, 5 μm for 2.5 Gbps), and then a athode eletrode was formed on the bak side. This APD is a high-sensitivity type with top surfae illuminated by using the light refletion from the DBR layer. In this struture, as shown in Fig. 1, a high eletri field area is onfined beneath the p-type area formed by the seletive diffusion and between the field ontrol layer and the n-type layer, i.e., within the InAlAs multipliation layer, and thus no edge breakdown ours. Although 1-nm level ontrol needs to be performed, suffiient ontrollability is possible with this struture beause the thiknesses of the multipliation layer and field ontrol layer are determined by the rystal growth proess. To ahieve a high speed of operation, the absorbing layer needs to be made thin, and assoiated low sensitivity is ompensated by returning the transmitted light to the absorbing layer using a DBR layer inserted below the absorbing layer. The peak refletivity wavelength of the DBR layer is tuned in the viinity of 1,55 nm. Most of the 1-Gbps ompatible APDs are of the bak surfae illuminated type, and so reeive light from the substrate side and transmit light through the absorbing layer, whih is refleted bak by the top surfae eletrode. In ontrast, the top surfae illuminated type, InP window layer Light-absorbing layer InAlAs multipliation layer DBR layer Cathode eletrode Low eletri field area High eletri field area Light-reeiving area P-type dopant diffusion area N-type InP substrate p+ Light n-inp substrate Fig. 1 Shemati diagram Anode eletrode Anode eletrode i-inp window layer SiN Graded l InGaAs light-absorbing layer p-inp field ontrol layer i-inalas multipliation layer N-type DBR layer Cathode eletrode *High Frequeny & Optial Devie Works **Advaned Tehnology R&D Center Mitsubishi Eletri ADVANCE September 29 17

20 whih is used for the present struture, does not need any speial mounting or implementation tehnique, and so is easy to use in ombination with a preamplifier. 2. Devie Charateristis The following setions desribe the devie harateristis. Figure 2 shows the typial urrent-voltage harateristis of a 1-Gbps devie with a diameter of 2 μm. The breakdown voltage V br is 28 to 32 V at 1 μa, and the dark urrent I d at 9% of the breakdown voltage is as low as 1 to 2 na. The sensitivity is as high as.95 A/W at the wavelength of 1.55 μm, and the hip apaitane is.17 pf at 9% of the breakdown voltage, whih is suffiiently low for 1-Gbps appliations. The 3 db bandwidths are 1 GHz and 8.5 GHz with multipliation fators of 3 and 1, respetively, and the gain-bandwidth produt is 12 GHz with multipliation fator exeeding 2. By installing the devie with a preamplifier, the minimum reeiver sensitivity was evaluated, providing a result of 29.9 dbm at 1 Gbps (bit error rate (BER) = 1 12 ) [6] as shown in Fig. 3. Charateristis of the 2.5-Gbps devie with a diameter of 5 μm are now desribed. The breakdown voltage V br is 34 to 4 V at 1 μa, and the dark urrent I d at 9% of the breakdown voltage is as low as 2 to 3 na. The sensitivity is as high as 1. A/W at the wavelength of 1.55 μm, and the hip apaitane is.27 pf at 9% of the breakdown voltage, whih is suffiiently low for 2.5-Gbps appliations. The 3 db bandwidths are GHz and 5. GHz with multipliation fators of 4 and 1, respetively, and the gain-bandwidth produt is 7 GHz with multipliation fator exeeding 2. By installing the devie with a preamplifier, the minimum reeiver sensitivity was evaluated, providing a result of 36.8 dbm at 2.48 Gbps (BER = 1 1 ) as shown in Fig Realization of High Reliability Reliability tests were performed in a nitrogen atmosphere at four temperature levels of 175ºC, 2ºC, 225ºC and 25ºC, using five samples at eah temperature. Eah APD was biased so that a urrent of 1 μa flowed orresponding to the breakdown voltage. The dark urrent was measured at room temperature when eah devie was periodially taken out of an aging test hamber. The degradation riterion was defined suh that the dark urrent reahes 2 na, i.e. twie the initial value. Figures 5 and 6 show the hanges of dark urrent with time. At 225ºC and 25ºC, the dark urrent starts to inrease gradually at a ertain time and the devies are eventually short iruited. At 175ºC and 2ºC, there is no sign of degradation until 1, and 8, hours, respetively. Figure 7 shows Weibull plots for the aging tests at 225ºC and 25ºC. The mean time to failure (MTTF) at 225ºC and 25ºC was 3,94 and 1,354 hours, respetively. The distributed parameter, m-value, was about 7 at both distributions, whih indiated that the degradation mode was due to wear-out failure. Figure 8 shows an Arrhenius plot of MTTF values at eah temperature. The ativation 9.95 Gbps 1.7 Gbps 11.1 Gbps Current (A) M=1 Photo urrent Bit error rate Bit error rate Dark urrent Voltage (V) Fig. 2 Current-voltage harateristis Average optial powers (dbm) Fig. 3 1-Gbps reeiving harateristis Fig Gbps reeiving harateristis Dark urrent (na) at R.T..9 Vbr Aging bias urrent: 1 μa, Aging temperature: 175ºC, Nitrogen atmosphere, N = 5 Aging time (hours) Fig. 5 Reliability test results at 175ºC Dark urrent (na) at R.T..9 Vbr ºC 25º C Criterion of degradation 2 2ºC 1 Aging bias urrent = 1μA Aging time (h) Fig. 6 Reliability test results at 2ºC, 225ºC and 25ºC 18

21 F(t) (%) ºC 225ºC MTTF=1354h MTTF=394h m=7 m= Time (hrs) Fig. 7 Weibull plots of servie life 1 MTTF (h) Ea=.96eV 2ºC (no degradation) 225ºC 25ºC Inverse of temperature (1/K) Fig. 8 Arrhenius plot of MTTF energy determined from the MTTF values at 225ºC and 25ºC was.96 ev. Using this ativation energy, the servie life at 85ºC was determined to be a very long time of 25 million hours. The degradation mode differs from the surfae degradation reported on InP-APD [7], where the degradation ours in the passivation film and rystal interfae with high eletri field [8, 9]. The inrease in the dark urrent ourred in the InGaAs absorbing layer, and the degradation was observed only at a high temperature of 225ºC or higher, and thus this degradation has a high ativation energy and is unlikely to our at pratial operating temperatures. It is inferred that this planar struture suppresses the surfae eletri field, resulting in a high reliability. 4. Conlusion As desribed above, we have developed InAlAs-APDs for 1-Gbps appliations (light-reeiving diameter of 2 μm) as well as those for 2.5-Gbps appliations with a larger reeiving diameter of 5 μm to failitate module installation. To realize a high sensitivity (= high S/N ratio), we fabriated a multipliation layer that multiplies the signal using InAlAs that generates only low exess noise during multipliation. The 1-Gbps devie ahieved a good minimum reeiver sensitivity of 29.9 dbm at 9.95 Gbps, and the 2.5-Gbps devie ahieved 36.8 dbm at 2.48 Gbps. In addition, we adopted a unique planar struture that is easy to fabriate and highly reliable to ahieve a long-term reliability of 25 million hours (85ºC), whih is equivalent or better than that of InP-APD having proven reliability. The newly developed high-sensitivity 2.5-Gbps and 1-Gbps InAlAs-APDs have suffiient harateristis for high-sensitivity reeiver appliations. Beause of their simple struture, these devies have a high prodution yield and thus a low ost. They are expeted to be widely used for optial aess networks, where low ost is a prerequisite. Referenes (1) J. C. Campbell, Reent Advanes in Teleommuniations Avalanhe Photodiodes, IEEE J. Lightwave Tehnol., Vol. 25, pp , Jan. 27. (2) I. Watanabe, T. Nakata, M. Tsuji, K. Makita, T. Torikai, and K. Taguhi, High-Speed, High-Reliability Planar-Struture Superlattie Avalanhe Photodiodes for 1-Gb/s Optial Reeivers, IEEE J. Lightwave Tehnol., Vol. 18, pp , De. 2. (3) S. Tanaka, S. Fujisaki, Y. Matsuoka, T. Tsuhiya, and S. Tsuji, 1 Gbit/s Avalanhe Photodiodes Appliable to Non-Hermeti Reeiver Modules, in Tehnial Digest of Optial Fiber Communiation Conf. 23, Vol. 1, MF55, pp (4) B. F. Levine, R. N. Saks, J. Ko, M. Jazwieki, J. A. Valdmanis, D. Gunther, and J. H. Meier, A New Planar InGaAs-InAlAs Avalanhe Photodiode, IEEE Photon. Tehnol. Lett., Vol. 18, pp , Sept. 26. (5) E. Yagyu, E. Ishimura, M. Nakaji, T. Aoyagi, and Y. Tokuda, Highly Produtive and Reliable 1 Gb/s AlInAs Avalanhe Photodiodes, in Pro. 31st European Conf. on Optial Communiations, Vol. 3, We , pp , Sept. 25. (6) E. Yagyu, E. Ishimura, M. Nakaji, H. Itamoto, Y. Mikami, T. Aoyagi, K. Yoshiara, and Y. Tokuda, High Speed and Low Noise Guardring-Free AlInAs Avalanhe Photodiode, Proeedings 1 of the 27 IEICE Soiety Conferene, C-3-42, pp. 165, Sept. 27. (7) H. Sudo and M. Suzuki, Surfae Degradation Mehanism of InP/InGaAs APD s, IEEE J. Lightwave Tehnol., Vol. 6, pp , Ot (8) E. Ishimura, E. Yagyu, M. Nakaji, Y. Tokuda, T. Aoyagi, and T. Ishikawa, High Reliability of Planar Type AlInAs-APD, 27 Spring Conferene, Japan Soiety of Applied Physis, 29p-SG-2, Mar. 27. (9) E. Ishimura, E. Yagyu, M. Nakaji, S. Ihara, K. Yoshiara, T. Aoyagi, Y. Tokuda, and T. Ishikawa, Degradation Mode Analysis on Highly Reliable Guardring-Free Planar InAlAs Avalanhe Photodiodes, IEEE, J. Lightwave Tehnol., Vol. 25, pp , De. 27. Mitsubishi Eletri ADVANCE September 29 19

22 43-Gbps EAM-LD Module / PD Module Author: Norio Okada* We have developed an eletroabsorption modulator laser diode (EAM-LD) module with a built-in driver IC and a photodiode (PD) preamp module, both for the next-generation 4-Gbps optial ommuniation system appliations. We applied a new offset iruit design to the EAM-LD module with a built-in driver IC to redue the load on the driver IC and also to redue heat generation from the termination resistor, and ahieved a good optial output waveform and low power onsumption. In the PD preamp module, we applied an asymmetri waveguide struture to the PD, ahieving a high light-reeiving sensitivity and wide bandwidth, as well as a 15.3-dB dynami range of the reeiving sensitivity harateristis. 1. Module Struture Figures 1 (a) and (b) show shemati drawings of the 43-Gbps EAM-LD module and PD module, respetively. In the 43-Gbps EAM-LD module, a built-in driver IC is integrated in one pakage, and GPPO TM onnetors are used as the high-frequeny input interfae. These devies redue the number of expensive high-frequeny pakages and ables, making optial transeivers smaller and heaper. A differential eletri signal input to the module is amplified by the driver IC. The operating onditions of the driver IC are adjusted to provide an optimum optial output waveform. Sine the harateristis of the EAM and LD are sensitive to the temperature, a thermo-eletri ooler (TEC) is used to maintain a onstant temperature. The PD module has a built-in trans-impedane amplifier (TIA) to onvert and amplify an optial signal to a voltage signal. The onverted high-frequeny signal is then AC oupled in the module and onneted to the GPPO TM onnetors for output. Figure 2 shows photos of the pakages. The pakage size of the EAM-LD module is 18 mm (length) 22 mm (width) 8.5 mm (height) exluding the fiber and pins, and the PD module is 15 mm (length) 22 mm (width) GPPO TM Connetor GPPO TM Connetor 8.5 mm (height), both ompliant with the 4 Gbit/s Miniature Devie Multi Soure Agreement (XLMD-MSA), whih is the standard for 4-Gbps optial modules. 2. EAM-LD Module Built-in Driver IC 2.1 Design As the operating speed of a transistor inreases, olletor-emitter breakdown voltage generally dereases. The driver IC for the modulator needs to be arefully designed in terms of the breakdown voltage in partiular, beause the IC needs to operate at high speed with a high output voltage swing. Figure 3 shows shemati drawings of (a) the present drive iruit, (b) the DC oupled drive iruit, and () the bias-t drive iruit. In the onventional DC-oupled drive iruit, the EAM, termination resistor, and driver IC are DC oupled, and the voltage swing, ross point, and offset voltage (a high level voltage applied to the EAM) are adjusted by the driver IC (1). Considering that the EAM needs a voltage swing of 2.5 V pp, and the offset voltage of the EAM is lowered to 1 V for optimizing the transmission harateristis, the olletor-emitter breakdown voltage of the transistor needs to be at least 3.5 V. In the present drive iruit, an adjustable offset voltage is diretly applied to the athode of the EAM. Sine no offset DC voltage is required at the anode eletrode, the breakdown voltage required for the transistor an be redued to 2.5 V. In addition, heat generation from the termination resistor is suppressed, and the power onsumption of the TEC an also be redued. Compared to the bias-t iruit, although the same olletor-emitter breakdown voltage of over 2.5 V is required, the present drive iruit redues high-frequeny losses beause it does not need any apaitor or oil for AC oupling to the signal line. The operating temperature of the EAM is set to 4ºC. At a high temperature, this setting redues the differene between ambient temperature and the EAM setting temperature, thus reduing the power on- Driver IC PD EAM LD Thermistor Lens TEC Fiber TIA Lens PD Fiber (a) EAM-LD module (b) PD module Fig. 1 Shemati drawing of modules Fig. 2 Photo of modules *Information Tehnology R&D Center 2

23 sumption of the TEC. 2.2 Results The module was evaluated by onneting it to a 16:1 multiplexer (MUX). Figure 4 shows 43-Gbps optial output waveforms with the PN31 pattern when the pakage temperature was varied from 5ºC to 8ºC. The LD urrent was 7 ma, and the setting temperature of the TEC was 4ºC. We ahieved an extintion ratio of 13.8 db to 14.1 db (speifiation: 8.2 db or greater), and a mask margin of 13% to 16% (speifiation: % or greater) using the ITU-T standard mask. Good transmission harateristis have also been ahieved. As shown in Fig. 5, the evaluation results indiate the power penalty after 4 ps/nm dispersion to be 1.2 db or lower at the full temperature range (speifiation: 2. db or lower). Figure 6 shows the TEC power onsumption. By setting the operating temperature of the EAM to 4ºC and suppressing the heat generation from the termination resistor, the TEC power onsumption has been suppressed to.7 W at a ase temperature of 8ºC. 3. PD Preamp Module 3.1 Design Figure 7 shows the struture of the PD. The PD has an edge illuminated type waveguide struture to ahieve wideband frequeny response and high sensitivity harateristis, with the anode and athode eletrodes arranged on the same plane. Generally, in an optial waveguide, the differene in refrative index between the InP lad layer and InGaAsP optial onfinement layer varies, and thus the amount of onfinement varies with the wavelength. Therefore, we applied our proprietary asymmetri waveguide struture, resulting in high sensitivity at both the 1.55 μm and 1.31 μm wave bands. Figure 8 shows the frequeny response harateristis of the PD. A 3-dB bandwidth of over 5 GHz has been ahieved at either wavelength. The light-reeiving sensitivity was.82 A/W and.91 A/W at 1.31 μm and 1.55 μm, respetively. High sensitivity and wide bandwidth have been ahieved at both wavelengths (2, 3). Driver Output (~ -2.5V) EAM DC offset (~+1.V) Mathing Resistor Driver Output Mathing (~ -3.5V) Resistor EAM Fig. 3 Bias iruits Driver pull up Driver Output (~ -2.5V) DC offset (-1.25V~-3.75V) Mathing Resistor EAM Extintion ratio 13.8dB Extintion ratio 14.1dB Extintion ratio 14.1dB Mask Margin 16% Mask Margin 15% Mask Margin 13% (a) degrees C (b) 25 degrees C () 8 degrees C Fig Gbps optial output waveforms Fig. 5 Measured 43Gbps BER versus reeived optial power Fig. 6 TEC power onsumption Mitsubishi Eletri ADVANCE September 29 21

24 Anode eletrode Optial input Contat hole for Anode eletrode Cathode eletrode SiN Semi-insulated Fe-InP p-inp lad p-ingaasp SCH InGaAs absorption n-ingaasp SCH n-inp lad d1 d2 was 12.7 to 11.7 dbm for the minimum sensitivity and +3.6 to +3.7 dbm for the maximum sensitivity, both orresponding to a temperature range of 5ºC to +8ºC, resulting in a dynami range of 15.3 db. The International Teleommuniations Union - Teleommuniation (ITU-T) G.693 speifiation is 6 dbm to +3 dbm, and thus our devie omplies with the standards with suffiient margin. The power onsumption was.19 W. Fig. 7 Shemati drawing of PD hip 25mV/div Fig. 1 43Gbps eletrial output waveforms 5mV/div Fig. 8 Frequeny response of PD hip 3.2 Results All evaluations were performed using a light soure with a wavelength of 1.55 μm. Figure 9 shows the frequeny response harateristis of the PD preamp module. A good frequeny response without any large ripples was ahieved with a 3-dB bandwidth of 3 GHz. Figure 1 shows 43-Gbps eletrial output waveforms at input levels of (a) 1 dbm and (b) +3 dbm. During either small signal or large signal operation, lear eye opening with a low jitter waveform was ahieved. The small signal differential gain was 62 dbω. Figure 11 shows the bit error ratio with the PN31 pattern. The bit error ratio was measured with the module onneted to a 1:16 demultiplexer (Demux). The error-free range, i.e. the range where the bit error ratio is or lower, Fig. 9 Frequeny response of PD preamp module Fig. 11 BER 4. Conlusions We have developed a 43-Gbps EAM-LD module and a PD module. Both modules exhibited good optial, high frequeny and bit error ratio harateristis, in addition to muh smaller size, lower power onsumption, and lower ost. The newly developed produts are expeted to beome established as ore elements of next-generation optial ommuniation tehnologies, and rapidly introdued for ommerial appliations. Referenes (1) N. Okada, et al.,.5 Vpp-drive Small-hirp 4 Gbit/s Eletroabsorption Modulator Module with Hybrid-integrated Driver IC, OFC23, paper FO6 (23). (2) S. Zaizen, et al., 4 Gbit/s PD Preamp Module for 1.3/1.55 μm Wavelength, 23 IEICE Eletronis Soiety Conferene, C (23). (3) M. Nakaji, et al., Asymmetri Waveguide Photodiode Over 5 GHz with High Sensitivity for Both 1.3 um and 1.55 um Wavelength, ECOC23, paper Th3.4.2 (23). 22