An Innovative Bidirectional Isolated Multi-Port Converter with Multi-Phase AC Ports and DC Ports

Transcription

1 An Innoatie Bidirectional Isolated Multi-Port Conerter with Multi-Phase Ports and DC Ports F. Jauch, J. Biela Laboratory for High Power Electronic Systems, ETH Zurich Physikstrasse 3, CH-892 Zurich, Switzerland This material is posted here with permission of the IEEE. Such permission of the IEEE does not in any way imply IEEE endorsement of any of ETH Zürich s products or serices. Internal or personal use of this material is permitted. Howeer, permission to reprint/republish this material for adertising or promotional purposes or for creating new collectie works for resale or redistribution must be obtained from the IEEE by writing to pubs-permission@ieee.org. By choosing to iew this document you agree to all proisions of the copyright laws protecting it.

2 An Innoatie Bidirectional Isolated Multi-Port Conerter with Multi-Phase Ports and DC Ports F. Jauch, J. Biela Laboratory for High Power Electronic Systems, ETH Zurich Physikstrasse 3, CH-892 Zurich, Switzerland Abstract This paper presents an innoatie bidirectional isolated multi-port conerter with multi-phase ports and DC ports, which is a key element of Solid-State Transformers (SST) utilized for example in a wind energy generation system. The multi-port conerter allows the direct coupling of the threephase system of the power generator with the three-phase utility grid and an additional DC storage unit applying a single high-frequency transformer structure. The topology is stackable and hence, single conerter modules can be connected in series at the input/output ports for medium or high oltage or in parallel for low oltage applications. The conerter is operated utilizing a time-arying phase-shift control to draw or inject sinusoidal currents with a corresponding amplitude and phase at the ports. The topology, its operating principle including the theoretical analysis and simulation results of a prototype system are proided. Keywords Multi-Port Conerter, -DC Conerter, Multi- Phase Port, Bidirectional, Isolated LV Generator a b N abc c LF S a LF S b LF S c : 1 n abc T a : 1 n abc T b : 1 n abc T c 1 : n ABC T A 1 : n ABC T B 1 : n ABC T C LF S A LF S B LF S C A B N ABC C MV Grid I. INTRODUCTION The use of renewable energy sources is constantly increasing in order to replace limited energy sources like coal, oil or uranium to reduce greenhouse gas emissions and account for the exhaustible primary energy sources. Wind energy represents an important part of today s renewable energy generation which demands suitable high power electronic equipment interfacing the generator, energy storage systems and the utility grid. For wind energy generation systems, different wind turbine concepts exist [1], [2]: For instance fixed speed wind turbines employing asynchronous squirrel cage induction generators or partial ariable speed wind turbines with ariable aerodynamic rotor resistance where both concepts show a direct connection of the generator and the grid ia a transformer. Concepts with ariable speed wind turbines show either a partial-scale (also known as the doubly-fed induction generator concept) or a full-scale - power conerter with a subsequent low-frequency transformer to connect to the utility grid. The partial- or full-scale - power conerter used in the latter two concepts usually consists of two stages, a threephase -DC rectifier and a subsequent three-phase DC- inerter [3]. The common DC-link additionally allows the connection of energy storage systems [4] to compensate for energy shortages during low wind conditions. Besides unidirectional power conerter solutions employing a costefficient diode rectifier on the generator side, seeral bidirectional single-cell and multi-cell power conerters hae been proposed [5]. Single-cell power conerters include two-leel and multi-leel conerters in back-to-back configuration like diode-clamped or flying-capacitor multi-leel conerter [6]. DC S dc DC Storage Fig. 1. Bidirectional isolated multi-port conerter with two three-phase ports (connected to the power generator and the utility grid) and one DC port (connected to an energy storage unit) in a wind energy generation system. For medium or high oltage applications, single conerter modules can be connected in series or parallel at the input/output ports and properly scaled according to oltage and power needs at the ports. Multi-cell power conerters comprise the modular multileel conerter [7], [8] in back-to-back configuration [9] and the cascaded H-bridge conerter [8], where additional dual-actie- (full-)bridge or dual-half-bridge modules form an additional isolation stage applying a high-frequency transformer [5], [1]. In this paper, an innoatie bidirectional isolated multi-port conerter with multi-phase ports and DC ports for use as a full-scale power conerter in a wind energy generation system + - C dc

3 as shown in Fig. 1 is proposed. The multi-port conerter allows the direct coupling of a three-phase system of a power generator with the three-phase utility grid and an additional DC storage unit applying a single high-frequency transformer structure. The single-stage power conersion leads to a low number of switching deices and gate dries. Due to the integrated transformers proiding isolation and oltage adaptation, the low-frequency transformer on the grid side is fully eliminated which in turn saes olume and weight. Furthermore, compared to three-phase conerter solutions, the proposed multi-port conerter is stackable and hence, single conerter modules can be connected in series or parallel at the input/output ports for medium or high oltage applications and properly scaled according to oltage and power needs at the ports. The operating principle allows to control the apparent power at the ports independently from each other, the power at the DC port is then gien inherently. Additionally, the proposed conerter concept offers the possibility to couple the power generator directly to a Medium or High Voltage DC (MVDC/HVDC) distribution grid. In the following, first the conerter topology and especially the introduction of multi-phase ports is presented in section II. Then, section III shows the operating principle comprising the modulation and control strategy in terms of a mathematical analysis. Finally, simulation results of a prototype system are proided in section IV. II. CONVERTER TOPOLOGY Fig. 1 shows the proposed multi-port conerter in a wind energy generation system. The conerter topology consists of port switching networks S a, S b, S c which connect to a first three-phase system with phases a, b, c and the star point N abc and port switching networks S A, S B, S C which connect to a second three-phase system with phases A, B, C and the star point N ABC. The frequencies and the phases of the two three-phase systems do not hae to be equal. Furthermore, a DC port switching network S dc is attached to a DC-link. All of the switching networks are coupled with each other through six two-winding transformers T a, T b, T c, T A, T B, T C where the three secondary windings of the transformers belonging to one three-phase system form a series interconnection. The switching networks in Fig. 1 consist of half-bridges with a clamping switch (also known as T-type circuit) with bidirectional switching deices when coupled to a lowfrequency (LF) port and of full-bridges with unidirectional switching deices when coupled to a DC port. Each switching network applies a high-frequency () square-wae oltage with or without clamping interal to the corresponding winding or the series connection of windings with an amplitude equal to half of the oltage which occurs on the LF side or the full DC oltage on the DC side and a phase angle in relation to a chosen reference. The switching frequency is chosen to be well aboe the frequencies of the two three-phase systems and the capacitors assumed to be large enough, so that the amplitudes of the generated square-wae oltages can be considered as constant during one switching cycle. For simplicity, in the following inestigations, only one part of the multi-port conerter which is depicted in Fig. 2 and represents the connection of the generator side with the DC- a b c L f L f N abc L f LF LF LF S a S b S c i p,a p,a i p,b p,b i p,c p,c T a T b T c i Fig. 2. Bidirectional isolated multi-port conerter with one three-phase port and one DC port resulting from splitting up the multi-port conerter shown in Fig. 1. link is analyzed. The analysis of the grid side of the conerter can be done in a similar way. A. Multi-Phase Ports Bidirectional isolated multi-port conerters in literature usually exhibit three DC ports which are coupled through a threewinding transformer and are controlled by the phase-shifts between the square-wae oltages applied to the windings [11] [13]. To achiee high efficiency in terms of low switching and conduction losses, the DC port oltages are kept at mainly constant oltages or additional duty cycle control is introduced to compensate for oltage ariations [11], [12]. In [12], Zero- Voltage-Switching (ZVS) conditions are based on keeping the half-cycle oltage-second products (half-cycle oltage-time integrals) applied to the windings equal. In the case of an -DC two-port conerter employing a primary half-bridge, whose oltage aries in time, with the use of a secondary fullbridge, the oltage-second product applied to the secondary winding can be adjusted to the one of the primary side. If a multi-phase system with phase oltages, whose absolute alues add up to a nearly constant sum oer time, is considered, a multi-phase port is formed which can be coupled directly to a DC port through transformer structures shown in Fig. 3. An equialent circuit of the conerter is gien in Fig. 4a. Hence, for controlling the conerter, a nearly constant sum of the half-cycle oltage-second products applied to the primary windings is aailable oer the whole system period. By using this nearly constant control ariable, which is further described in section III, the conerter efficiency can be kept high. The term nearly constant in this context does not mean negligible ripple. In case of a symmetrical three-phase system, the sum of the absolute phase oltages with amplitude ˆV abc corresponds to a six-pulse wae form with a constant aerage alue 6 ˆV abc / oer the system period exhibiting a 2(/3 1) % = 9.4 % peak-to-peak oltage ripple. DC S dc L dc V dc + -

4 B. Transformer Structure for Port Coupling The use of the nearly constant sum of the absolute alues of the phase oltages requires a transformer structure where the oltages applied to the windings are added. Such a structure comprises for instance three two-winding transformers as depicted in Fig. 3a. There, the oltages p,a, p,b, p,c are summed up through the series interconnection of the secondary windings. Magnetically, the sum of the oltages corresponds with the sum of the winding fluxes. Therefore, the oltage sum can be translated into a winding flux sum which leads to a single four-winding transformer shown in Fig. 3b where the core winding fluxes are summed up through the secondary winding. In Fig. 3a, the oltages p,a, p,b, p,c applied to the primary windings impress the primary winding fluxes Φ p,a, Φ p,b, Φ p,c according to Faraday s Law. The oltage applied to the series connection of the secondary windings predefines the flux sum Φ s,a +Φ s,b +Φ s,c. The mismatch of the primary and secondary winding fluxes per core results in the leakage fluxes Φ σ,a, Φ σ,b, Φ σ,c which show the same alue for negligible magnetizing fluxes if all secondary windings hae the same number of turns. Hence, all of the windings exhibit the same leakage inductance current referred to a specific winding. The three transformers show the primary referred leakage inductances L σ,a, L σ,b, L σ,c. In case of the single four-winding transformer depicted in Fig. 3b, the oltages p,a, p,b, p,c applied to the primary windings impress the winding fluxes Φ p,a, Φ p,b, Φ p,c whereas the oltage applied to the secondary winding predefines the winding flux Φ s. Again, the mismatch of the impressed winding fluxes causes the leakage fluxes Φ σ,a, Φ σ,b, Φ σ,c which can be summed up to a total leakage flux. Also for this transformer structure, neglecting the magnetizing fluxes, all of the windings show the same leakage inductance current referred to a specific winding. The four-winding transformer exhibits a total leakage inductance L σ referred to the primary side. It is concluded, that the two transformer structures gien in Fig. 3 exhibit a comparable electrical behaiour seen from the windings for negligible magnetizing fluxes. An equialent electrical circuit is gien by Fig. 4a where the total leakage inductance L σ = L σ,a + L σ,b + L σ,c referred to the primary side is drawn. III. OPERATING PRINCIPLE Like the three-port DC-DC conerters discussed in [11] [13], the proposed multi-port conerter is operated by phaseshift control where the square-wae oltages applied to the transformer windings are phase-shifted against each other to control the power flow at the ports. The total leakage inductance L σ of the transformer structure acts as decoupling and energy transfer element between the square-wae oltages as can be seen from Fig. 4a. A. High-Frequency Square-Wae Voltage Summation The transformer structures depicted in Fig. 3 sum up the square-wae oltages p,a, p,b, p,c which are generated by the switching networks S a, S b, S c and phase-shifted towards the square-wae oltage by the angles φ a, φ b, φ c as can be seen in Fig. 5. There, also the resulting oltage sum p,s i Φ p,a p,a p,a p,b i p,b Φ p,b i Φ p,c p,c p,c Φ σ,a Φ σ,b Φ σ,c (a) Φ s,a Φ s,b Φ s,c i i Φ p,a Φ p,b Φ p,c i p,a i p,b i p,c Φ σ,a p,a Φ σ,b p,b Φ σ,c p,c Fig. 3. Possible transformer structures for port coupling in the multi-port conerter shown in Fig. 2 employing three two-winding transformers with leakage inductances L σ,a, L σ,b, L σ,c (a) and one four-winding transformer with leakage inductance L σ (b). p a p b p c p,c L σ Lσ p,a p,b p,s (a) p dc φs p,s φi (b) Lσ Fig. 4. Equialent circuit of the conerter in Fig. 2 for representing phaseshift control with a primary referred total leakage inductance L σ = L σ,a + L σ,b + L σ,c (a) and phasor diagram for the fundamental model in case of capacitie port currents. and the transformer leakage inductance current referred to the primary side oer two switching periods T s = 1/f s are drawn. The adaptation of the phase angles φ a, φ b, φ c within the period of the system enables keeping the half-cycle oltagesecond product of the oltage sum p,s (shaded area in Fig. 5) at a nearly constant alue. Therefore, a nearly constant ratio between the half-cycle oltage-second product of the oltage sum p,s and the oltage generated by switching network S dc is achieed. In other words, this means that the amplitude ratio of the fundamentals of p,s and which are shown in Fig. 5 is time-independent. Summarized, the aboe mentioned nearly constant control ariable of the sum of the half-cycle oltage-second products (b) Φ s

5 p,a p,b p,c φ a φ b Phase-Shift Control ference Ts φ c a /2 Ts δ abc c /2 b /2 δ abc δ abc phasors which are gien by p,a = 2 a cos (δ abc ) e jφa, (1) p,b = 2 b cos (δ abc ) e jφ b, (2) p,c = 2 c cos (δ abc ) e jφc, (3) = 4n abcv dc cos (δ dc ) e jφ dc, (4) p,s Fig. 5. Square-wae oltages p,a, p,b, p,c applied to the primary windings, the resulting oltage sum p,s, the primary referred square-wae oltage applied to the series interconnection of the secondary windings and the transformer leakage inductance current in case of b > a > c in -to-dc operation oer two switching cycles T s. enables to compose a oltage sum p,s = p,a + p,b + p,c with a nearly constant half-cycle oltage-second product and a constant amplitude of the fundamental, respectiely. B. High-Frequency Fundamental Model The multi-port conerter depicted in Fig. 2 can be modelled by an equialent circuit according to Fig. 4a by means of square-wae oltage sources for each switching network and a primary referred total leakage inductance L σ of the applied transformer structure. The leakage inductance current flows through all of the equialent sources and hence defines, together with the port oltage, the instantaneous power p a, p b, p c, p dc which has to be deliered to or drawn from the corresponding port. In the following, for the sake of simplicity, the analytical considerations are based on a fundamental model of the conerter where higher order harmonics of the squarewae oltages are neglected. The oltages considering the clamping interals 2δ abc, 2δ dc (see Fig. 5) and the resulting leakage inductance current are then described as complex V dc δ dc = î Lσ e jφi = p,a + p,b + p,c j L σ. (5) The phasors rotate with the angular frequency = 2f s. Additionally, p,a, p,b, p,c show a time-dependent amplitude caused by the oltage waeforms and a timedependent phase caused by the time-arying phase-shift control. The square-wae oltage is chosen to be the reference for the phase-shift control as shown in Fig. 5 and its phasor is placed on the real axis of the complex coordinate system, hence φ dc = (see Fig. 4b). C. Modulation and Control Strategy For the proposed multi-port conerter, a time-arying phaseshift control is applied, where the phase angles φ a, φ b, φ c according to (1)-(3) represent the control functions whose waeforms hae to assure sinusoidal phase currents i a, i b, i c with a gien amplitude Îabc and a gien phase-shift towards the phase oltages a = ˆV abc cos (ω abc t), (6) ( b = ˆV abc cos ω abc t 2 ), 3 (7) ( c = ˆV abc cos ω abc t 4 ). 3 (8) To get the required phase currents, the control functions are chosen in such a way that the corresponding instantaneous powers p a, p b, p c are drawn from or deliered to the ports which in turn are modelled by the equialent oltage sources. The aerage actie powers flowing out or into these sources oer one switching cycle T s can be determined with the model according to { } 1 p a = 2 î Lσ cos (δ abc ) p,a = a cos (φ a φ i ), { } (9) 1 p b = 2 î Lσ cos (δ abc ) p,b = b cos (φ b φ i ), { 1 p c = 2 p,c } = c î Lσ cos (δ abc ) cos (φ c φ i ). (1) (11) Taking the reactie power consumed by the capacitors into account, the reference alue of the control current can be represented by a phasor I ctrl = I abc I C (12)

6 I abc L f = φ i, (25) V abc I C I ctrl where δ abc, represent the control ariables. Inserting the control functions φ a, φ b, φ c according to (21)- (23) into (1)-(3) and adding the oltage phasors, the composed oltage sum phasor Fig. 6. Single-phase equialent circuit of LC filter input stage. The current I ctrl is controlled such that the desired phase current I abc results. as shown in Fig. 6 where the capacitor current is gien by I C = jω abc 2 (V abc jω abc L f I abc ). (13) The load is represented by a controlled current source I ctrl which corresponds to the drawn or injected power at the port in phase-shift operation. The phasor of the phase reference current is I abc = Îabce j. (14) Î abc denotes the reference amplitude whereas is the reference phase of the port currents i a, i b, i c. In the time domain, the reference alues of the control currents per phase are then gien by i a = Îctrl cos (ω abc t + ), (15) ( i b = Îctrl cos ω abc t 2 ) 3 +, (16) ( i c = Îctrl cos ω abc t 4 ) 3 +, (17) with amplitude Îctrl = I ctrl and phase = I ctrl. The control functions φ a, φ b, φ c can be determined using the nonlinear system of equations i a = i a = îlσ cos (δ abc ) i b = i b = îlσ cos (δ abc ) i c = i c = îlσ cos (δ abc ) cos (φ a φ i ), (18) cos (φ b φ i ), (19) cos (φ c φ i ), (2) while considering î Lσ = and φ i =. By comparing (15)-(17) with (18)-(2) and knowing that δ abc, δ dc [, /2], the most obious control functions which lead to sinusoidal port currents show the mathematical form φ a = ω abc t +, (21) φ b = ω abc t 2 3 +, (22) φ c = ω abc t (23) The amplitude and the phase of the reference alues of the control currents according to (15)-(17) are then gien by Î ctrl = îlσ cos (δ abc ), (24) p,s = p,a + p,b + p,c = ˆ p,s e jφs = 3 ˆV abc cos (δ abc ) e jφs (26) results, which shows a time-independent amplitude ˆ p,s and a time-independent phase angle. Applying the rotation operators e jφa, e jφ b, e jφc on the sinusoidal oltages a, b, c, a fixed space ector is formed, which is then rotated oer a switching cycle T s. Due to the time-independent phasors p,s,, also the transformer leakage inductance current shows a constant amplitude and phase oer time. Fig. 4b depicts a phasor diagram of the phasors p,s,, Lσ, in case of capacitie port currents. D. Accuracy of High-Frequency Fundamental Model In general, a periodic square-wae signal s(t) with clamping interal 2δ, amplitude h and angular frequency like the oltages drawn in Fig. 5 can be represented by its fourier series s(t) = 4h k=1 cos ((2k 1)δ) sin ((2k 1) t). (27) 2k 1 When adding the square-wae oltages p,a, p,b, p,c while considering their phase-shifts from (21)-(23) and their amplitudes according to the half of (6)-(8), it can be seen from the fourier series (27), that harmonics with an odd multiple of three of the fundamental frequency cancel out and are not present in the oltage sum p,s. Neertheless, if exhibits these orders of harmonics, corresponding current harmonics in the transformer leakage current will be drien. This causes power shares on the ports which lead to LF current harmonics exhibiting odd multiples of three of the fundamental system frequency ω abc. To suppress these LF current harmonics, δ dc = /6 can be chosen which in turn cancels out the corresponding harmonics in the square-wae oltage. Besides the fundamental, current harmonics of order 5, 7, 11, 13, 17,... remain in the transformer leakage current and contribute to a small current distortion. The control functions aboe are deried by setting p abc = p abc,(1) (28) where p abc denotes the reference alue of the instantaneous power of the considered phase and p abc,(1) the fundamental power. In a next step, also higher order power shares are considered by soling p abc = p abc,(k) (29) k=1 in order to get improed control functions φ a, φ b, φ c and a lower input current THD.

7 IV. PROTOTYPE SYSTEM For alidating the theoretical analysis, a prototype system of the multi-port conerter drawn in Fig. 2 is simulated in GeckoCIRCUITS [14] where the system oltage and frequency are fixed at 23 V rms and 5 Hz. The DC port is considered to be at a constant oltage of 4 V dc. The simulation model corresponds to Fig. 2 and applies three twowinding transformers, which show negligible high magnetizing inductances and primary referred leakage inductances of 3 µh each. The conerter is operated at a constant frequency of 2 khz. Table I summarizes the simulation parameters. A. Solutions of Control Variables In order to obtain the desired port phase currents with corresponding amplitude Îabc and phase, the control ariables δ abc, hae to be determined. This can be done by soling the nonlinear system of equations (18)-(2) while taking the control functions (21)-(23) into account. Solutions φ i p,s φ i p,s Lσ = /2 (a) = Lσ φ i φi p,s Lσ p,s = /4 (b) = /8 Lσ Control Variable (rad) δabc Control Variable (rad) φs = /2 = /4 = /2 = = = /4 = /4 φ.2 abc = /4 = ±/ Current Amplitude Iˆ abc (A) Fig. 7. Solutions of the control ariables δ abc, for the prototype system with parameters from Table I and δ dc = /6 for phase angles = { /2, /4,, /4, /2} ersus phase current amplitude Îabc. TABLE I SIMULATION PARAMETERS OF THE PROTOTYPE SYSTEM. system oltage V abc 23 V rms system frequency f abc 5 Hz DC port oltage V dc 4 V dc Switching frequency f s 2 khz Transformer turns ratios n abc 1/2 Transformer leakage inductances L σ,a, L σ,b, L σ,c 3 µh Transformer magnetizing inductances - neglected Total transformer leakage inductance L σ 9 µh Inductors L f 1 µh Capacitors 5 µf Inductor L dc 1 µh Capacitor C dc 5 µf φ i (c) = /4 (e) p,s Lσ ilσ φi (d) = /2 (f) p,s Lσ Fig. 8. Phasor diagrams for the fundamental model for phase control angles = /2 (a), = /4 (b), = (c), = /8 (d), = /4 (e) and = /2 (f). of the control ariables for the prototype system with parameters gien in Table I are depicted in Fig. 7 for phase angles = { /2, /4,, /4, /2}. For each phase angle, the port current amplitude is limited to a maximum which corresponds to the maximum transferrable apparent power at the specific phase angle. For small current amplitudes, the control ariable δ abc shows almost the same alue where the oltage sum phasor p,s exhibits approximately the same length as the DC oltage phasor. Fig. 8 shows the phasor diagrams for different phase control angles in -to-dc operation. In case of negligible reactie power consumed by the capacitors, equals the phase current angle. B. Simulation sults The multi-port conerter is simulated in steady-state with δ dc = /6 in -to-dc operation with reference alues Î abc = 2 A and = /8 for the phase currents. The control ariables are then gien by δ abc =.7543 and =.427. In the simulation model according to Fig. 2, the star point of the conerter N abc and the star point of the grid are connected to earth through 1 Ω earth resistances. The DC side is also grounded ia a 1 Ω earth resistance. The simulation

8 Voltage (V) a b c V dc Voltage (V) p,c p,b p,a Voltage (V) C,a C,b C,c C,dc Voltage (V) p,s Current (A) ia ib ic Time (ms) i dc Current (A) i p,a /i p,b /i p,c i Time (ms) Fig. 9. Simulated phase currents i a, i b, i c and DC current i dc of the proposed multi-port conerter with parameters gien in Table I for reference alues Îabc = 2 A and = /8 in -to-dc operation. The capacitor oltages C,a, C,b, C,c represent the oltages across the series connection of the two capacitors in each phase. C,dc is the oltage across the DC capacitor C dc. Fig. 1. Simulated transformer winding oltages p,a, p,b, p,c,, resulting oltage sum p,s and transformer winding currents i p,a, i p,b, i p,c, i of the proposed multi-port conerter with parameters gien in Table I for reference alues Îabc = 2 A and = /8 in -to-dc operation. results without feedback control applying the deried control functions are shown in Fig. 9 and Fig. 1. V. CONCLUSION An innoatie bidirectional isolated multi-port conerter with multi-phase ports and DC ports is presented, which is ideally suited for use in a wind energy generation system. The concept of multi-phase ports is introduced, which allows the direct coupling of a multi-phase system, whose absolute phase oltages add up to a nearly constant sum oer time, with one or more DC ports oer a single high-frequency transformer structure. The proposed conerter is operated by a time-arying phase-shift control to draw or inject sinusoidal currents with a corresponding amplitude and phase at the ports. The analytical description and simulation results of a prototype system for alidating the conerter topology are proided. KNOWLEDGMENT The authors would like to thank Swisselectric search and the Competence Center Energy and Mobility (CCEM) ery much for their strong financial support of the research work. REFERENCES [1] A. D. Hansen and L. H. Hansen, Wind Turbine Concept Market Penetration oer 1 Years ( ), Wind Energy, ol. 1, no. 1, pp , 27. [2] F. lo, M. Ciobotaru, and F. Blaabjerg, Power Electronics Control of Wind Energy in Distributed Power Systems, in Proc. 11th International Conference on Optimization of Electrical and Electronic Equipment (OPTIM), 28. [3] F. Blaabjerg, Z. Chen, R. Teodorescu, and F. Io, Power Electronics in Wind Turbine Systems, in Proc. CES/IEEE 5th International Power Electronics and Motion Control Conference (IPEMC), ol. 1, 26, pp [4] W. Li and G. Joos, A Power Electronic Interface for a Battery Supercapacitor Hybrid Energy Storage System for Wind Applications, in Proc. IEEE Power Electronics Specialists Conference (PESC), 28, pp [5] F. Blaabjerg, M. Liserre, and K. Ma, Power Electronics Conerters for Wind Turbine Systems, IEEE Transactions on Industry Applications, ol. 48, no. 2, pp , 212. [6] F. Z. Peng, A Generalized Multileel Inerter Topology with Self Voltage Balancing, IEEE Transactions on Industry Applications, ol. 37, no. 2, pp , 21. [7] A. Lesnicar and R. Marquardt, An Innoatie Modular Multileel Conerter Topology Suitable for a Wide Power Range, in Proc. IEEE Power Tech Conference, ol. 3, 23. [8] H. Akagi, Classification, Terminology, and Application of the Modular Multileel Cascade Conerter (MMCC), IEEE Transactions on Power Electronics, no. 99, 211. [9] M. Saeedifard and R. Iraani, Dynamic Performance of a Modular Multileel Back-to-Back HVDC System, IEEE Transactions on Power Deliery, ol. 25, no. 4, pp , 21. [1] H. Fan and H. Li, High-Frequency Transformer Isolated Bidirectional DC-DC Conerter Modules with High Efficiency oer Wide Load Range for 2 kva Solid-State Transformer, IEEE Transactions on Power Electronics, ol. 26, no. 12, pp , 211. [11] H. Tao, J. L. Duarte, and M. A. M. Hendrix, Three-Port Triple-Half- Bridge Bidirectional Conerter with Zero-Voltage Switching, IEEE Transactions on Power Electronics, ol. 23, no. 2, pp , 28. [12] H. Tao, A. Kotsopoulos, J. L. Duarte, and M. A. M. Hendrix, Transformer-Coupled Multiport ZVS Bidirectional DC-DC Conerter with Wide Input Range, IEEE Transactions on Power Electronics, ol. 23, no. 2, pp , 28. [13] C. Zhao, S. D. Round, and J. W. Kolar, An Isolated Three-Port Bidirectional DC-DC Conerter with Decoupled Power Flow Management, IEEE Transactions on Power Electronics, ol. 23, no. 5, pp , 28. [14] [Online]. Aailable: