High Boost Hybrid Transformer DC DC Converter for Photovoltaic Module Applications

Transcription

1 High Boost Hybrid Transformer DC DC Converter for Photovoltaic Module Applications K.Umadevi,Associate Professor Abstract This paper presents a nonisolated, high boost ratio hy-brid transformer dc dc converter with applications for low-voltage renewable energy sources. The proposed converter utilizes a hy-brid transformer to transfer the ductive and capacitive energy simultaneously, achievg a high boost ratio with a smaller sized magnetic component. As a result of corporatg the resonant operation mode to the traditional high boost ratio pulsewidth modulation converter, the turn-off loss of the switch is reduced, creasg the efficiency of the converter under all load conditions. The put current ripple and conduction losses are also reduced because of the hybrid learsusoidal put current waveforms. The voltage stresses on the active switch and diodes are mataed at a low level and are dependent of the changg put voltage over a wide range as a result of the resonant capacitor transfer-rg energy to the output of the converter. The effectiveness of the proposed converter was experimentally verified usg a 220-W prototype circuit. Utilizg an put voltage rangg from 20 to 45 and a load range of W, the experimental results show system of efficiencies greater than 96% with a peak efficiency of 97.4% at 35- put, 60-W output. Due to the high system effi-ciency and the ability to operate with a wide variable put voltage, the proposed converter is an attractive design for alternative low dc voltage energy sources, such as solar photovoltaic modules and fuel cells. ndex Terms energy sources with low dc voltage, European union (EU) effi-ciency, high boost ratio dc dc, high efficiency, hybrid transformer, photovoltaic (P) module.. NTRODUCTON Due to the risg costs and limited amount of nonrenew-able energy sources, there is an creasg demand for the utilization of renewable energy sources such as photovoltaic (P) modules. ntegratg the power from the P module to the existg power distribution frastructure can be achieved through power conditiong systems (PCS). Typical PCS can be accomplished usg a sgle-stage or a double-stage as shown Fig.. The double-stage PCS consists of a dc dc conversion stage that is connected to either a low-power dividual verter or a highpower centralized verter that multiple converters could connect to. The dc dc conversion stage of the PCS requires a high efficiency, high boost ratio dc dc converter to crease the low dc put voltage from the P panel to a higher dc voltage. This voltage has to be higher than the peak output voltage of the dc ac verter, nomally the range. The double-stage design can also suppress ac le double frequency by utilizg the active ripple cancellation technique. The high boost ratio dc dc converter for such systems can be isolated or nonisolated however, transformer-isolated converters tend to be less efficient and more expensive due to the creased manufacturg costs. A nonisolated dc dc converter with a high boost ratio would be advantageous for a two-stage PCS because it can be easily tegrated with current P systems while reducg the cost and matag a high system efficiency. Due to the different output voltages from the P panel, it would be beneficial to have a system with a high efficiency over the entire P voltage range to maximize the use of the P durg different operatg conditions. Another important function of the DC DC converter for P applications is beg able to implement maximum power pot trackg (MPPT). The ability to implement MPPT for an dividual P panel would ensure that a large cluster of P could mata maximum power output from each panel without terferg with the other panels the system. The major consideration for the ma power stage of the converter beg able to implement an accurate MPPT is that the put current ripple of the converter has to be low. The high boost ratio nonisolated dc dc converter the uses of coupled-ductor and switched-capacitor are attractive for use a simple high boost ratio converter due to the fact that only a sgle low voltage active switch is required. The reason that the primary side active switches of the high boost ratio converters have low voltage stress is because of the transformer effect from the coupled-ductors.sce there is a low voltage stress on the active switch, the circuits can then use low voltage MOSFETs that generally have a low Rds(on) and smaller switchg periods, decreasg both the conduction and switchg losses. An earlier paper on high boost ratio nonisolated dc dc converter presented a clamp-mode couple-ductor buck boost converter. The converter s leakage energy from the coupled-ductor was recycled reducg the losses of the system. The output diode stress for this converter was similar to that of a traditional Fig.. Typical double-stage PCS architectures with high boost ratio dc dc converters and PWM dc ac verters for P applications. (a) Two-state P module tegrated microverter. (b) Parallal P module tegrated microconverter with centralized verter

2 Fig. 2. High step-up dc dc converters usg coupled-ductor and switched-capacitor techniques. (a) High-step coupled-dcutor roboost dc-dc converter. (b) High step-up dc-dc converter with coupled-ductor and switched-capacitor. voltage operation. Because of the resonant capacitor transferrg energy to the output of the converter, all the voltage stresses of the diodes are kept under the output dc bus voltage and dependent of the put voltage. The efficiency of the proposed converter was verified experimentally utilizg a 220-W prototype circuit with an put voltage from 20 to 45. flyback converter, i.e., higher than the output dc bus voltage. Another drawback of the converter was that there was a highput current ripple due to the fact that there was no direct energy transfer path when the MOSFET was OFF. Further improvements creasg the boost ratio of a simple dc dc converter were accomplished by combg a boost converter with a flyback converter as shown Fig. 2(a). The boost ratio was improved as a result of the outputs of the boost converter and flyback converter beg connected series. By addg a switched-capacitor series with the energy transformer path, a new improved high boost ratio dc dc converter with coupledductor and switched-capacitor, as shown Fig. 2(b). With the switched-capacitor serted between the primary side and secondary side of the coupled-ductor, the boost ratio was creased and the output diode voltage stress was reduced closer to that of the output dc bus voltage. Light load efficiency of the converter is also reduced because switchg losses were more domant under light load conditions. n this paper, a high boost ratio dc dc converter with hybrid transformer is presented to achieve high system level efficiency over wide put voltage and output power ranges. By addg a small resonant ductor and reducg the capacitance of the switched-capacitor the energy transfer path, a hybrid operation mode, which combes pulsewidth modulation (PWM) and resonant power conversions, is troduced the proposed high boost ratio dc dc converter. The ductive and capacitive energy can be transferred simultaneously to the high-voltage dc bus creasg the total power delivered decreasg the losses the circuit. As a result of the energy transferred through the hybrid transformer that combes the modes where the transformer operates under normal conditions and where it operates as a coupled-ductor, the magnetic core can be used more effectively and smaller magnetics can be used. The con-tuous put current of the converter causes a smaller current ripple than that of previous high boost ratio converter topolo-gies that used coupledductors. The lower put current ripple is useful that the put capacitance can be reduced and it is easier to implement a more accurate MPPT for P modules. The conduction losses the transformer are greatly reduced because of the reduced put current RMS value through the primary side. The voltage stress of the active switch is always at a low voltage level and dependent of the put voltages. Due to the troduction of the resonant portion of the current, the turn-off current of the active switch is reduced. As a result of the decreased RMS current value and smaller turn-off current of the active switch, high efficiency can be mataed at light output power level and low-put Fig. 3. Proposed high step-up dc dc converter with hybrid transformer.. PROPOSED CONERTER TOPOLOGY AND OPERATON ANALYSS Fig. 3 shows the circuit diagram of the proposed converter. C is the put capacitor; HT is the hybrid transformer with the turns ratio :n; S is the active MOSFET switch; D is the clampg diode, which provides a current path for the leakage ductance of the hybrid transformer when S is OFF, C c captures the leakage energy from the hybrid transformer and transfers it to the resonant capacitor C r by means of a resonant circuit composed of C c, C r, L r, and D r ; L r is a resonant ductor, which operates the resonant mode; and D r is a diode used to provide an unidirectional current flow path for the operation of the resonant portion of the circuit. C r is a resonant capacitor, which operates the hybrid mode by havg a resonant charge and lear discharge. The turn-on of D r is determed by the state of the active switch S. D o is the output diode similar to the traditional coupled-ductor boost converter and C o is the output capacitor. R o is the equivalent resistive load. Fig. 4 illustrates the five steady-state topology stages of the proposed dc dc converter for one switchg cycle. Fig. 5 shows the key voltage and current waveforms for specific components of the converter over the switchg cycle. For the waveforms represented Fig. 5, g represents the driver signal for the active MOSFET switch S ; i s is the current of the MOSFET S ; i C r is the current of the resonant capacitor C r ; i C c is the current of clampg capacitor C C ; i is the primary side current of hybrid transformer; i o is the current through the output diode; v s and v D o are the voltage waveforms of the active switch MOSFET S and the output diode D o, respectively. For simplicity, we assume that the dc put voltage is a stiff voltage source with a constant

3 voltage, the load is a resistor and all the switch and diodes are ideal devices. The five operation modes are briefly described as follows. [t0,t ], [Fig. 3(a)]: n this period, MOSFET S is ON, the magnetizg ductor of the hybrid transformer is charged by put voltage, Cr is charged by Cc, and the secondary-reflected put voltage n of the hybrid transformer together by the resonant circuit composed of secondary side of the hybrid transformer, Cr, Cc, Lr, and Dr. The energy captured by Cc is transferred to Cr, which turn is transferred to the load durg the off-time of the MOSFET. The current MOSFET S is the sum of the resonant current and lear magnetizg ductor current as shown Fig. 5. There are two distctive benefits that can be achieved by the lear and resonant hybrid mode operation. The first benefit is that the energy is delivered from source durg the capacitive mode and ductive mode simultaneously. Compared to previous coupled-ductor high boost ratio dc dc converters with only ductive energy delivery, the dc current bias is greatly reduced, decreasg the size of the magnetics. Second, the turnoff current is decreased, which causes a reduction the turn-off switchg losses. [t,t2 ], [Fig. 4(b)]: At time t, MOSFET S is turned OFF, the clampg diode D is turned ON by the leakage energy stored the hybrid transformer durg the time period that the MOSFET is ON and the capacitor Cc is charged which causes the voltage on the MOSFET to be clamped. The conversion ratio is similar to the conventional boost converter except that the turns ratio term n is added, so the traditional duty ratio control method that is applied for a standard boost converter can also be applied to the proposed converter. [t2,t3 ], [Fig. 4(c)]: At time t2, the capacitor Cc is charged to the pot that the output diode Do is forwarded biased. The energy stored the magnetizg ductor and capacitor Cr is beg transferred to the load and the clamp diode D contues to conduct while Cc remas charged. [t3,t4 ], [Fig. 4(d)]: At time t3, diode D is reversed bi-ased and as a result, the energy stored magnetizg ductor of the hybrid transformer and capacitor Cr is simultaneously transferred to the load. Durg the steady-state operation, the charge through capacitor Cr must satisfy charge balance. The key waveform of the capacitor Cr current shows that the ca-pacitor operates at a hybrid-switchg mode, i.e., charged resonant style and discharged lear style. [t4,t0 ], [Fig. 4(e)]: The MOSFET S is turned ON at time t4. Due to the leakage effect of the hybrid transformer, the output diode current io will contue to flow for a short time and the output diode Do will be reversed biased at time t0 ; then the next switchg cycle starts. The boost ratio Mb can be obtaed by three flux balance crite-ria for the steady state. The first flux balance on the magnetizg ductor of hybrid transformer requires that steady state Second, accordg to flux balance on the resonant ductor durg on-time The last flux balance that governs the circuit is voltagesecond balance of the magnetizg ductor the hybrid transformer for the whole switchg period D = o C r + n ( D). (3) By substitutg (2) to (3), the boost conversion ratio can be obtaed M b = o = n + 2. (4) D Fig. 5. Key waveforms for different operation stages.

4 Fig. 4. Operation modes of the high boost ratio dc dc converter with hybrid transformer. (a) t 0 -t. (b) t -t 2. (c) t 2 -t 3. (d) t 3 -t 4. (e) t 4 -t 5.. ANALYSS AND ADANTAGES OF THE PROPOSED CONERTER A. Fixed oltage Stresses of the Power Devices Proper ratg and all the results are with respect to the output dc voltage. From the circuit diagram of t 0 to t and t to t 2 Fig. 4, respectively, the voltage stresses for MOSFET S and clampg diode D are obtaed o S = D = = D n. + 2 (5) From the circuit diagram of t 0 to t and t 2 to t 3 Fig. 4, one obtas the voltage stress of diode resonant diode D r and output diode D o ( + n) o D r = D o = o C c = o = D 2 + n. (6) From (5) and (6), it is obvious that all the voltage stresses of the switches are dependent of put voltage and load conditions. n other words, all the voltage stresses of the switches are optimized based on the output voltage and the turns ratio of the transformer. The resonant period T r and the resonant frequency are given by L r,total = L r + L lrs + n 2 L lrp (9) oltage stresses for all the power devices of the converter are determed this section to select power devices with the where L lrs is the secondary side leakage ductance and L lrp is the primary side ductance of the hybrid transformer. The T r =2π ₃ L r C r (7) f r =. (8) T r f the constant on-time control T on is used, choose T on = /2T r so that the resonant diode can turn OFF at zero-current condition and conduction loss can be mimized. n the experimental im-plementation of the hybrid transformer, the leakage ductance of the hybrid transformer should be considered, so that the total resonant ductance is expressed Fig. 6. as follows Waveforms for energy transfer analysis.

5 resonant capacitance C r is composed by C r and C c series. Normally, we choose C r ₃ C c so that the voltage stress of the MOSFET can be clamped well. The optimal operation mode is the constant PWM on-time T on control with variable frequency, however, traditional PWM control method is applicable to the proposed converter as described [26] and [27]. B. Analysis of Energy Transfer The simplified waveforms for energy transfer analysis are shown Fig. 6. n order to analyze the energy transfer feature from the low voltage dc energy source to the high-voltage dc bus, it is necessary to solve the equivalent circuit Fig. 3(a) subject to the itial conditions imposed by the previous PWM OFF-time terval given by i L r (0)= 0 (0) v C r (0)= v C r () where v C r is the ripple of the resonant capacitor C r. The resonant solutions are obtaed as i L i r (t) = L r s 2πfr t (2) where R N v C r (t) = v C r cos 2πf r t (3) i v C r = R N L r (4) is characteristic impedance given by RN = ₃ L r. (5) C r For PWM off-time terval, the discharge equations of the resonant capacitor C r are given by v Lm C r = Lm = sec T off 2C r o = P o (6) sec (7) D o D where Lm sec is the average lear magnetizg current referred to secondary side of the hybrid transformer, o is the average output current, P o is the output power, and o is the output voltage. The relationship between and lear magne-tizg current and susoidal resonant current can be expressed as i L r = π f r T s Lm sec ( D) = π f r T s o. Accordgly, the average primary side susoidal resonant current of hybrid transformer is given by L Lr pri = π n i L r Substitutg (8) to (9) yields. (8) (9) Substitutg (22) to (20) yields f r = 2 D T s. (22) L Lr pri = n o. (23) 2D The resonant contribution dex k r of energy transfer by susoidal resonant current can be defed as the ratio between the average put resonant current L Lr pri to the total put current k r = L Lr pri = n 2 D o ₃ ₃ n + D 2 o ₃ = ( D) n = n. (24) 2D n + 2 2D o n order to optimize the operation of the proposed converter, kr needs to be creased, this will reduce the turnoff losses of the MOSFET and decrease the size of the magnetic core used. The curve of the resonant energy transfer contribution dex Kr at different put voltage conditions is shown Fig. 7. The oper-atg conditions for the curve Fig. 7 are for when the output voltage o equals 400, the turns ratio of the hybrid transformer n equals 40:9, and an put voltage range from 20 to 45. For a given power and fixed output voltage, the resonant energy trans-fer contribution dex creases along with the crease put voltage. This feature helps improve the converter efficiency over a wide put voltage range by decreasg the conduction losses which are more domant at low-put voltages and reducg the switchg losses that are more domant at highput voltages. Fig. 7. K r versus curve. L Lr pri = n f r T s o. (20) The average put current can be obtaed from (4) by power balance n + 2 = D o. (2) Fig. 8. For the optimal mode operation, the relationship between the resonant frequency and the switchg period is nput current comparison between resonant mode and lear mode. (a) Resonant mode. (b) Lear mode.

6 Fig. 9. Two-phase extension for proposed converter. C. Advantages Over Conventional Nonresonant High Step-Up Converter Current popular methods used to achieve high boost ratio for nonisolated dc dc converters consist of usg coupledductor and switched-capacitor techniques]. The converter presented utilizes hybrid-switchg technique combg PWM and resonant power conversions to achieve a high boost ratio while matag a high efficiency. The put currents for the resonant susoidal charge mode and the PWM lear charge mode are comparatively illustrated Fig. 8. The proposed converter works usg the resonant susoidal charge mode, while a conventional nonresonant converter works usg the lear charge mode. For a fixed output power and given put voltage, the average put currents for these two converters shown Fig. 8 are equal. Areas A and B (see Fig. 8) show the capacitive energy transferred by the hybrid transformer of the proposed converter with resonant mode and lear energy transferred by the coupled-ductor of the converter. The switchg losses for a dc dc converter are directly proportional to the switchg current given by the fixed conversion voltages. As shown Fig. 5, the MOSFET is turned ON at time t = t 4, the raisg rate of the primary cur-rent is limited by the leakage ductor of the hybrid transformer alleviatg the turn-on losses. The ma switchg loss then becomes the turn-off switchg losses. For the resonant mode charge of the proposed converter, the turn-off switchg cur-rent, as shown Fig. 8(a), consists of only the magnetizg current as a result of resonant operation. For the conventional nonresonant converter,the turn-off switchg current 3, as shown Fig. 8(b), is the sum of the magnetizg current and the switchedcapacitor charge current, which is dependent on the leakage ductance of the coupled-ductor. For a given capacitor value of the switched-capacitor, creasg the leakage ductance can reduce the raisg rate of the primary side current to reduce the turn-off current; however, the conversion ratio will decrease because of the reduced couplg factor k of the coupled-ductor.as a result, the leakage ductance design of the coupled-ductor has a tradeoff between the conversion ratio and a higher turn-off switchg current. With the troduction of a resonant operation mode to the PWM converter for the proposed converter, the primary peak current 2, as shown Fig. 8(b), is smaller than the peak current 3 of its switched-capacitor counterpart. Sce the resonant mode is employed as opposed to the switchedcapacitor mode, the capacitance of the charge capacitor C r can be greatly reduced. Hence, utilizg the resonant mode allows the use of smaller sized magnetic components and lower profile charge capacitors for C r which can have a low capacitance. This is perfect for an application where a low profile P-moduletegrated dc dc converter is needed. The leakage ductance of the hybrid transformer can also be effectively utilized as part of the resonant mor loop simplifyg the design of the transformer for the proposed converter. D. Two-Phase nterleaved Extension n order for the proposed converter to be used higher power level conversion applications, the terleavg method applica-ble to the traditional high boost ratio PWM dc dc converter can be employed, as shown Fig. 9. This gives the advantages of standard terleaved converter systems such as low-put cur-rent ripple, reduced output voltage ripple, and lower conduction losses. The difference between standard terleaved convert-ers and the proposed terleaved converter is that the clampg capacitor C c can also be shared by the terleaved units reduc-g the total number of components the system. Usg the phase-shift method of control, the current ripple through the clampg capacitor C c is reduced as a result the capacitance needed for C c is also reduced... CONCLUSON A high boost ratio dc dc converter with hybrid transformer suitable for alternative dc energy sources with low dc voltage put is proposed this paper. The resonant conversion mode is corporated to a traditional high step-up PWM converter with coupled-ductor and switched-capacitor obtag the fol-lowg features and benefits: ) This converter transfers the capacitive and ductive

7 en-ergy simultaneously to crease the total power delivery reducg losses the system. 2) The conduction loss the transformer and MOSFET is re-duced as a result of the low-put RMS current and switch-g loss is reduced with a lower turn-off current. With these improved performances, the converter can mata high efficiency under low output power and low-put voltage conditions. 3) With low-put ripple current feature, the converter is suit-able for P module and fuel cell PCS, where, accurate MPPT is performed by the dc dc converter. A prototype-circuit-targeted P module power optimizer with put voltage range and 400- dc output was built and tested. Experimental results show that the MOSFET voltage was clamped at 60 and the output diode voltage was under 350. These results were dependent of the put voltage level. The conversion efficiencies from 30 to 220 W are higher than 96% and the peak efficiency is 97.4% under 35- put with 60-W output power. REFERENCES [] J.-S. Lai, Power conditiong circuit topologies, EEE nd. Electron. Mag., vol. 3, no. 2, pp , Jun [2] S. B. Kjaer, J. K. Pedersen, and F. Blaabjerg, A review of sglephase grid-connected verters for photovoltaic modules, EEE Trans. nd. Appl., vol. 4, no. 5, pp , Sep./Oct [3] F. Blaabjerg, Z. Chen, and S. B. Kjaer, Power electronics as efficient terface dispersed power generation systems, EEE Trans. Power Electron., vol. 9, no. 5, pp , Sep