An overview of key elements and characteristics of a radio. Power amplifiers & nonlinear distortions, low-noise amplifiers & noise figures, antennas,

Transcription

1 ECSE413B: COMMUNICATIONS SYSTEMS II Tho Le-Ngoc, Winter 2008 ELEMENTS OF A RADIO TRANSCEIVER An overview of key elements and characteristics of a radio transceiver: Power amplifiers & nonlinear distortions, low-noise amplifiers & noise figures, antennas,

2 A POINT-TO-POINT LINK & DIGITAL MICROWAVE TRANSCEIVER TRANSCEIVER Microwave cable TRANSCEIVER interface modulator up-converter PA IF RF interface demodulator down-converter LNA SWITCH or DUPLEXER PAGE 2

3 FREQUENCY TRANSLATION: UP & DOWN CONVERSION up-converter S(f) interface mod X BPF PA f A OSC or FREQ. SYNTHESIZER interface demod BPF X LNA down-converter Modulated signal s(t) has its spectrum S(f) centered at f 0. UP-CONVERVERTER: g(t)=s(t).cos(2πf A t) G(f)=0.5{S(f-f A )+ S(f+f A )} f D SWITCH or DUPLEX XER Select either S(f-f A ) or S(f+f A ) by filtering, e.g., S(f+f A ) centered at f B =f 0 +f A DOWN-CONVERVERTER: r(t)=g(t).cos(2πf D t) R(f)=0.5{G(f-f D )+ G(f+f D )} Select either G(f-f D D) or G(f+f D D) by filtering, e.g., G(f-f D D) centered at f c=f B-f D Signal is not distorted by frequency translation PAGE 3

4 ELECTROMAGNETIC SPECTRUM PAGE 4

5 POWER CAPABILITY: SOLID-STATE STATE POWER AMPLIFIER PAGE 5

6 POWER CAPABILITY: MICROWAVE TUBE PAGE 6

7 POWER CAPABILITY COMPARISON: SOLID STATE & TUBES PAGE 7

8 TWT: TRAVELING WAVE TUBE PAGE 8

9 Linear amplification of asingle single-input input frequency (a) linear amplification (b) time domain (c) frequency domain PAGE 9

10 Linear mixing with linear amplification (a) linear amplification (b) time domain (c) frequency domain PAGE 10

11 INPUT/OUPUT POWER CHARACTERISTICS P saturated OUTPUT POWER P out,dbm Ideal linear curve: P out,dbm = P in,dbm +G gain,db highest possible output power, P saturated OUTPUT BACKOFF [db] = P saturated -P out P out (X-1) db LINEAR REGION P out,dbm = P in,dbm +G gain,db Operating point X db INPUT BACKOFF [db] INPUT POWER, P in,dbm P in = P in,sat -P in P in,sat PAGE 11

12 NONLINEAR AMPLIFICATION OF ONE SINGLE-FREQUENCY INPUT (a) nonlinear amplification (b) time domain (c) frequency domain PAGE 12

13 NONLINEAR AMPLIFICATION OF TWO SINE WAVES (a) nonlinear amplification (b) time domain (c) frequency domain PAGE 13

14 OUTPUT SPECTRUM OF A NONLINEAR AMPLIFIER WITH 2 SINGLE-FREQUENCY INPUTS DESIRED UNWANTED HARMONICS DESIRED UNWANTED INTERMODULATION (IM) PRODUCTS PAGE 14

15 EFFECT OF NONLINEAR AMPLIFICATION ON MULTICARRIER INPUT Input to the amplifier has many carriers, e.g., 3 unmodulated carriers: x(t)=acos(ω 1 t)+bcos(ω 2 t)+ccos(ω 3 t) quasi-linear amplifier: y(t)=g 1 x(t)+ G 3 [x(t ) ] 3 y(t)=g 1 [Acos(ω 1 t)+bcos(ω 2 t)+ccos(ω 3 t)] + G 3 [Acos(ω 1 t)+bcos(ω 2 t)+ccos(ω 3 t)] 3 [Acos(ω 1 t)+bcos(ω 2 t)+ccos(ω 3 t)] 3 = A 3 cos 3 (ω 1 t)+b 3 cos 3 (ω 2 t)+c 3 cos 3 (ω 3 t) + 3A 2 Bcos 2 (ω 1 t)cos(ω 2 t)+ 3A 2 Ccos 2 (ω 1 t)cos(ω 3 t) + 3B 2 Acos 2 (ω 2 t)cos(ω 1 t)+ 3B 2 Ccos 2 (ω 2 t)cos(ω 3 t) + 3C 2 Acos 2 (ω 3 t)cos(ω 1 t)+ 3C 2 Bcos 2 (ω 3 t)cos(ω 2 t) + 6ABCcos(ω 1 t)cos(ω 3 t)cos(ω 2 t) PAGE 15

16 INTERMODULATION PRODUCTS cos 3 (ω n t)= 0.75cos(ω n t)+0.25 cos(3ω n t), remaining term after filtering: 0.75cos(ω n t) cos 2 (ω n t)cos(ω m t)=0.5 cos(ω m t)+0.25 cos(ω 2n-m t) cos(ω 2n+m t) remaining terms after filtering: 0.5 cos(ω m t)+0.25 cos(ω 2n-m t) where cos(ω 2n-m t) is an inband intermodulation interferer cos(ω 1 t)cos(ω 3 t)cos(ω 2 t)=0.25[cos(ω t)+cos(ω t)+cos(ω t+ cos(ω t)] remaining terms after filtering: 0.25[cos(ω t)+cos(ω t)+cos(ω t] where 0.25[cos(ω t)+cos(ω t)+cos(ω t are inband intermodulation ti interferers After filtering, y(t)=acos(ω 1 t)+bcos(ω 2 t)+ccos(ω 3 t)+ IM where IM= dcos(ω t)+ ecos(ω t)+fcos(ω t)+gcos(ω t)+hcos(ω t) +icos(ω t)+ jcos(ω t)+kcos(ω t)+lcos(ω t) are intermod interferers 3 RD ORDER IM PRODUCTS (IM3) HAS POWER RELATED TO INPUT SIGNAL POWER: P IM3, dbm =3P in, dbm +g db A2: Wireless Channels 2006 Tho Le-Ngoc PAGE 16

17 3 RD -ORDER IM PRODUCTS 2-TONE TEST (TEST WITH 2 SINGLE-FREQ. INPUTS 3 RD ORDER IM PRODUCTS (IM3) HAS POWER RELATED TO INPUT SIGNAL POWER: P IM3, dbm =3P in, dbm +g db (IM3 LINE with slope 3) RECALL: INPUT/OUTPUT POWER RELATIONSHIP FOR DESIRED SIGNAL: P out,dbm = P in,dbm +G gain,db (I/O LINE with slope 1) 3 rd -order intercept point in dbm is the assumed interception of IM3 and I/O LINES PAGE 17

18 EXAMPLE OF AMPLIFIER OUTPUTS FOR 9 INPUT TONES PAGE 18

19 Advanced PA Technologies Ultra Linear Power Amplifiers PAs are potentially 50-70% of future base station cost: Aim is to develop technologies for dramatically lower cost Digital Pre-distortion compensates for PA non-linearities: Enabled by accurate modelling of power devices RF Feed-Forward is replaced by Digital Pre-Distortion DSP-based algorithms for adaptive compensation Digital based correction implementation leads to lower cost & high efficiency PAGE 19

20 THERMAL NOISE IN RECEIVER signal + noise signal with noise Thermal noise produced by random motion of charged particles (e.g., electrons) has a Gaussian distribution and a power spectral density (PSD): S n (f)=h f /{exp(h f /kt)-1} f f W/Hz where k=1.38e-23 Joules/ K (Boltzman s constant) h=6.62e-34 62E Joules.sec sec (Plank s constant), K= 273+ C For f <0.1kT/h (about 1E12 room temperature (290 K) S n (f) kt=-174dbm/hz PSD [dbm/hz] THERMAL NOISE PSD E+02 1.E+02 1E+04 1.E+04 1E+06 1.E+06 1E+08 1.E+08 FREQUENCY [MHz] PAGE 20

21 CONCEPTS OF NOISE FACTOR, NOISE FIGURE: IDEAL, NOISELESS AMPLIFIER: Input: s i (t)+n i (t) Output: t a p {s i (t)+n i (t)} Output signal part: a p s i (t) Output noise part: a p n i (t) a n : voltage gain, A n =a n2 : power gain S i : input signal power N i : input noise power S o = A n S i : output signal power N o = A n N i : output noise power AMPLIFIER WITH INTERNAL NOISE: Input: s i (t)+n i (t) Output: t a p {s i (t)+n i (t)}+n d (t) Output signal part: a p s i (t) Output noise part: a p n i (t)+n d (t) a n : voltage gain, A n =a n2 : power gain S i : input signal power N i : input noise power, N d : internal noise power S o = A n S i : output signal power N o = A n N i +N d : output noise power S i /N i : input SNR in S o /N o = A n S i /A n N i : output SNR out = SNR in S i /N i : input SNR in S o /N o = A n S i /(A n N i +N d ): output SNR out < SNR in [S o /N o ]=[S i /N i ].[1+N d /N i A p ] -1 = S i /N i ].[F ] -1 where F=1+N d /N i A p =[S i /N i ]/ [S o /N o ], >1 F: noise factor indicating the factor of SNR degradation d at the amplifier output t noise figure NF db =10log 10 (F)= SNR in,db -SNR out,db PAGE 21

22 NOISE FACTOR, NOISE FIGURE OF CASCADED AMPLIFIERS: S o /N o [db] N 1 N 2 N n SIGNAL POWER S i A 1 S i A 2 A 1 S i S o =S i (A 1 A 2 A n ), NOISE POWER N i A 1 N i +N 1 A 2 (A 1 N i +N 1 )+N 2 N o N o =N n +N n-1 A n +N n-2 A n-1 A n + +N 2 A n A n-1 A 3 +N 1 A n A n-1 A 2 +N i A n A n-1 A 2 A 1 N o =N i (A n A 2 A 1 ){F 1 +(F 2-1)/A 1 +(F 3-1)/(A 1 A 2 )+ +(F n -1)/(A 1 A 2 A n-1 )}= N i (A n A 2 A 1 )F overall where F overall = {F 1 +(F 2-1)/A 1 +(F 3-1)/(A 1 A 2 )+ +(F+(F n -1)/(A 1 A 2 A n-1 )} F overall =[S i /N i ]/ [S o /N o ], and SNR out,db =SNR in,db -(NF overall,db ) where NF overall,db =10log 10 (F overall ) From F overall = {F 1 +(F 2-1)/A 1 + +(F n -1)/(A 1 A 2 A n-1 )} it is clear that if A 1 is sufficiently large then F 1 is dominant (i.e., contributions of F 2,,F n are small) Therefore, in a receiver, the front-end amplifier is a low-noise amplifier, i.e., with small F 1 and large A 1 Example: 3 stages with A 1=30dB, A 2=A 3=10dB and NF 1=3dB, NF 2=8dB, NF 3=10dB F overall = F 1 +(F 2-1)/A 1 +(F 3-1)/(A 1 A 2 )=10 3/10 +(10 8/10-1)/10 30/10 +(10 10/10-1)/10 40/10 =10 3/10 +(5.31E-3)+(9E-4)=2.001=3.013dB PAGE 22

23 NOISE PERFORMANCE OF LNA s PAGE 23

24 RECEIVER: OVERALL NOISE FACTOR From ANT BPF1 LNA MIXER1 BPF2 IF AMP DEMOD Loss Gain: G LNA loss: Loss Gain G IF L BPF1 F LNA L MIXER1 L BPF2 F DEMOD F IFAMP For a matched-impedance passive component, its noise factor = insertion loss Overall gain from input of BPF1 to input of DEMOD: G receiver = G LNA G IF /(L BPF1, L MIXER1 L BPF2 ) G receiver, db =-L BPF1,dB + G LNA,dB -L MIXER1,dB -L BPF2,dB +G IF,dB Overall noise factor (linear scale): F receiver = L BPF1 +(F LNA -1) L BPF1 + (L MIXER1-1) L BPF1 /G LNA +(L BPF2-1) L MIXER1 L BPF1 /G LNA +(F IFAMP -1) L MIXER1 L BPF1 L BPF2 /G LNA +(F DEMOD -1)L MIXER1 L BPF1 L BPF2 /G LNA G IF OR EQUIVALENTLY, WE CAN GROUP VARIOUS BLOCKS AS FOLLOWS: BPF1+LNA G 1 =G LNA /L BPF1 F 1 =L BPF1 F LNA MIXER1+BPF2+IF AMP G 2 =G IF /(L MIXER1 L BPF2 ) F 2 =L MIXER1 L BPF2 F IFAMP DEMOD F DEMOD F receiver = F 1 + (F 2-1) /G 1 +(F DEMOD -1)/G 1 G 2 PAGE 24

25 equivalent noise temperature For a receiver with an overall F overall, S o /N o =S i /(N i F overall ) where N i =kt and T is the input absolute temperature. The effective noise spectral density at the receiver input is N i F overall =kt F overall It can be expressed as: N i F overall =kt+kt(f overall -1) Since the NF and F are specified at the reference temperature T o =290 K, it is better to write N i F overall =k(t+t e ) where T is the actual input temperature and T e =T o (F overall -1) is a hypothetical value equivalent to an excessive noise temperature due to the excessive noise spectral density generated by the system. In summary, the additional noise generated by the system or device can be expressed in terms of noise factor (F), noise figure (NF) and equivalent noise temperature (T e ) where NF=10log 10 (F) and T e =T o (F-1) PAGE 25

26 SPHERICAL COORDINATES θ: POLAR or COLATITUDE ANGLE φ: LONGITUDE ANGLE FOR CONVENIENCE, CONSIDER SINGLE MAIN LOBE WITH AXIS ASSUMED TO LIE IN THE xy-plane θ: ELEVATION ANGLE φ: AZIMUTH ANGLE PAGE 26

27 RADIATION PATTERNS absolute (fixed power) radiation pattern relative (fixed distance) radiation pattern relative (fixed distance) radiation pattern in decibels relative (fixed distance) radiation pattern in decibels for an omnidirectional (point source) antenna PAGE 27

28 SIMPLIFIED EQUIVALENT CIRCUIT OF AN ANTENNA RF POWER TO ANTENNA: P RF(IN) =P D +P R ANTENNA EFFICIENCY: η=p R /P RF(IN) =R r /(R r +R e ) ANT DIRECTIVITY GAIN: D=P/P REF, P REF : POWER OF A REFERENCE (ISOTROPIC) ANTENNA ANT POWER GAIN: G ANT =ηd EIRP(EFFECTIVE ISOTROPIC RADIATED POWER) EIRP=P R D=P RF(IN) G ANT POWER DENSITY PER UNIT AREA AT A POINT WITH DISTANCE d FROM ANT: p d =(EIRP)/(4πd 2 ) PAGE 28

29 CAPTURE AREA & CAPTURED POWER P T Tx POWER P T Tx ANT POWER GAIN G T EIRP=P T G T T T G T Distance d POWER DENSITY PER UNIT AREA AT DISTANCE d: p d =(P T G T )/(4πd 2 ) P C Rx ANT POWER GAIN G R Rx (CAPTURED) POWER P C p d : AMOUNT OF POWER INCIDENT ON EACH UNIT AREA OF AN IMAGINARY SURFACE (PERPENDICULAR TO THE DIRECTION OF PROPAGATION OF THE ELECTROMAGNETIC WAVE). EFFECTIVE CAPTURE AREA OF THE Rx ANTENNA: A C =(G R λ 2 )/(4π) where λ=c/f: wavelength Rx CAPTURED POWER: P C =A C p C =(G R P T G T λ 2 )/(4πd )2 =P T (G T G R )/(4πdf/c )2 FREE-SPACE LOSS: L FREE-SPACE =(4πdf/c) 2 P CdB C,dBm =P TdB T,dBm +(G TdB T,dB +G RdB R,dB )- L FS, db L FS, db = 10log 10 (L FREE-SPACE ) = log 10 (f GHz )+20log 10 (d km ) PAGE 29

30 ANTENNA BEAMWIDTH Side lobes Side lobes PAGE 30

31 ANTENNA POWER GAIN AND BEAMWIDTH RELATIONSHIP PAGE 31

32 PARABOLIC REFLECTORS MAIN BEAM AND SIDE LOBES GEOMETRY & RADIATION DIRECTIONS (a) focal point outside the reflector (b) focal point inside the reflector PAGE 32

33 PARABOLIC ANTENNA WITH A HORN FEED: WAVEGUIDE HORN TYPES PAGE 33

34 PARABOLIC ANTENNAS PARABOLIC ANTENNA WITH A CENTER FEED PARABOLIC ANTENNA WITH A CASSEGRAIN FEED PAGE 34

35 ANTENNA NOISE -174 dbm/hz ~ -190 dbm/hz PAGE 35

36 Smart Antenna Principles $ Smart antenna (SA) systems r1 () t can be used for Rx and Tx. They exploit the spatial r2 () t Radio Unit dimension via spatial (IF down conversion sampling and coherent followed by A/D) processing of the EM wave field. r N (t) Four main system components (Rx mode): Control unit CTL (weight update) Antenna array: N elements, geometrical configuration. Radio unit: RF down-conversion, A/D conversion. Beam-forming (BF) network (BFN): signal weighting followed by summation. RF F-to-IF/AD DC Linear beamforming network: Control unit: adjusts BF weight to achieve desired d spatial response. Ideally, a set of weights is maintained and updated for each individual mobile user. SA can adapt to current radio conditions and tailor individual user beam- patterns so as to maximize i SIR: Communication link continually optimized. BFN sˆt () PAGE 36

37 examples of antenna patterns sector antenna N sets of four orthogonal beams from a four-column antenna array with a 0.5λ horizontal element spacing N=1 N=2 N=3 PAGE 37

38 EXAMPLE OF BEAMFORMING TO IMPROVE SIR PAGE 38

39 Smart Antenna Classification Switched beam (SB) systems: May be viewed as an extension to sectorization. Uses fixed set of pre-computed beams. Users assigned to different beams on the basis of received power. Requires beam switching as users roam around. Dynamically phased array (DPA): Ability to steer beams/nulls in arbitrary directions. Requires angle of arrival (AOA) estimation of signal and possibly interference (several approaches available). AOA info used to update BF weights so that SIR is maximized. As users change location, AOA and BF weights continually updated. Adaptive antenna (AA) systems: Uses fully adaptive scheme to optimize BF weights, based on available information: input/output signals, training sequence, etc. PAGE 39

40 Non-regenerative repeaters RF IF PAGE 40

41 BASEBAND REGENERATIVE REPEATER DEMOD MOD PAGE 41

42 Example: 8-channel high/low frequency plan (a) west to east PAGE 42

43 Example: 8-channel high/low frequency plan (b) east to west PAGE 43

44 protection switching arrangements: hot standby PAGE 44

45 protection switching arrangements: diversity to DEMOD to DEMOD to DEMOD to DEMOD REFERENCES: materials from various sources PAGE 45