R industrial and aerospace applications due to their numerous

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1 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 4. NO., JANUARY Integral Cycle Mode Control of the Series Resonant Converter GYU B. JOUNG, CHUN T. RIM, AND GYU H. CHO Abstract-A new control ethod for the series resonant converter (SRC) in which the device switching instants are always synchronized to the zero crossing points of the resonant current is proposed. Output voltage is controlled by proper selection of the switch odes by varying the duty ratio of powering ode and free resonant ode. Each switch ode is analyzed and the dc transfer function is obtained. It is also shown that the siulation results coincide well with the practical case. The SRC with the proposed control schee shows any advantages, such as low current switching stress, low switching loss, low electroagnetic interference, high input power factor, and wide control range. Soe of these characteristics are verified experientally. I. INTRODUCTION ESONANT converters are becoing widely used in R industrial and aerospace applications due to their nuerous erits [ -[9]. Conventional series resonant converters (SRC s) can be classified into two schees according to the control ethod. One is the frequency doain control schee, and the other is the phase doain control schee. In the frequency doain control schee, the output voltage is controlled by the ratio of the switching frequency and the circuit resonant frequency [ -[5], [7], [SI. On the other hand, in the phase doain control schee, it is achieved by the phase difference between two inverters the switching frequency of each converter is fixed to the resonant frequency [9]. In the frequency doain control schee, the output voltage depends largely on the load condition as well as the switching frequency, which akes it difficult to control the syste properly [7]. In particular, the control range of the output voltage becoes narrow for light load operation, and the device current switching stress becoes severe when the switching frequency deviates fro the resonant frequency [6]. The phase doain control schee is ore advantageous than the frequency doain control schee in ters of the load independency. However, this schee also undergoes severe current switching stress as phase difference becoes large [9]. It is well-known that the SRC operates optially when the switching frequency exactly coincides with the resonant frequency. The output voltage cannot be controlled, Manuscript received February 8, 988; revised August 8, 988. This paper was presented at the 988 IEEE Power Electronics Specialists Conference, Kyoto, Japan, April -4. The authors are with the Departent of Electrical Engineering, Korea Advanced Institute of Science and Technology, P.O. Box 50, Chongyang, Seoul , Korea. IEEE Log Nuber however, so far as the frequency and phase are fixed. Therefore, a tie doain control ethod is suggested in this paper to adjust the output voltage still satisfying its optiu operating conditions. This proposed control schee has inherent advantages such as zero current switching stress, low switching loss, low electroagnetic interference (EMI), and wide control range. Switch odes are analyzed for an SRC with MOS- FET, and a practical control schee is suggested and verified experientally.. OPERATING MODES AND CONTROL METHOD OF SRC A. Operating Modes The basic power circuit topology of the SRC is shown in Fig.. The power circuit has four switch odes as shown in Fig. 2: the powering ode, the free resonant ode, the regeneration ode, and the discontinuous ode. These odes are classified according to the direction of power flow. Switch on/off control always occurs in synchronization with the current zero crossing points. Denoting the circuit quantities as input voltage V,, output voltage u0, inductor current il, capacitor voltage vc, and the resonant tank circuit and load paraeters cor, 2, and Q which are the description of each ode is as follows. ) Powering Mode: S,S4 and S2,S3 pairs are turned on and off alternately in synchronization with the current zero crossings. The applied voltage U, across the tank circuit and the resonant current il are in phase as shown in Fig. 3(a). During this ode, the source power is delivered to the LC tank including the load. 2) Free Resonant Mode: D2,S4 and S2,D4 pairs (or, equivalently, Dl,S3 and S,D3 pairs) are turned on and off alternately. In this ode, power is not supplied to the tank circuit while the sinusoidal resonant current free-runs and gradually decreases as shown in Fig. 3(b). In this ode, the LC tank energy is delivered in the load. 3) Regeneration Mode: All of the forced coutation switches (Sl-S4) are turned off. Instead, the Dl,D4 and /89/ $0.OO O 989 IEEE

2 84 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 4. NO. I, JANUARY 989 vs +ll -T -.h L- AS4 D4 I CO I i I I IVo(t) + Fig.. Power circuit topography of SRC. DI,D4 D2,D3 DI.D4 D2.D3 S. D : on (C) (d) Fig. 2. Four odes of SRC operation. (a) Powering ode. (b) Free resonant ode. (c) Regeneration ode. (d) Discontinuous ode. D2,D3 pairs are alternately turned on and off in synchronization with the resonant current. During this ode, the applied voltage across the tank circuit and resonant current are out of phase, so the resonant current decreases faster than the current of the free resonant ode. As shown in Fig. 3(c), the LC tank energy is returned to the source while transferring part of the energy to the load. 4) Discontinuous Mode: As the previous free resonant or regeneration ode goes on, the tank resonant current decreases, and it can reach zero, as shown in Fig. 3(d), without any energy stored in the LC tank. Fro these characteristics, the operation odes are suarized in Table I. B. Mode Control Methods The output voltage of SRC can be controlled by oderate selection of the four odes. Mode selection is allowed only at current zero crossing points, thus the ode interval length should be an integer ultiple of half cycles of the resonant frequency. In this sense, it is called the integral cycle ode control schee. The diagra of the tie doain control schee is shown in Fig. 4. In this diagra, powering, free resonant, and regeneration odes are selectable by switches S-S4. Thus these odes are controllable except the discontinuous ode because the discontinuous ode occurs independently of the external switches. The powering ode increases the output voltage, as the free resonant and regeneration odes decrease it. Thus the output voltage can be controlled by properly cobining the powering ode and either the free resonant or the regeneration odes. Coparing the two passive odes, the regeneration ode deteriorates the input power factor, thus it is undesirable to include this ode in the control schee. Therefore, the powering and free resonant odes with (or without) the discontinuous ode are used for the output voltage control in this paper. Fig. 5 briefly illustrates the wavefors of the proposed control schee. In this control schee, the output voltage is regulated by the duty ratio of the powering ode and free resonant ode with (or without) discontinuous ode. As the duty ratio increases, the output voltage is also increased.. ANALYSIS OF THE PROPOSED CONTROL SCHEME The equations for analyzing the resonant operation are shown in the Appendix for the four respective odes. However, exact analysis takes a long tie, and understanding the tie doain control characteristics is very difficult. For siplicity, the operation odes of the SRC can be approxiately analyzed by the low ripple approxiation ethod [4] because the output filter capacitor CO is sufficiently larger than the tank circuit capacitor C in the practical case. To apply the low ripple approxiation ethod, all variables during the kth resonant period, kt/2 I t < (k + ) T/2 are denoted by index k. As indicated in Fig. 6, v,(k) is the output voltage, U,( k) is the tank capacitor voltage at the switching instant, and ip ( k) is the peak current during the kth event. Because the agnitude of the capacitor voltage is related to the stored energy of the capacitor and the agnitude of the inductor current is also related to the stored energy of the inductor, we can change the variables U, (k), ip (k) into new variables U,* (k), ip* (k), U,*(k) = (@)I (4) i,*(k) = I i,(k) I (5) and assuing that CO >> C, then the discrete state equa-

3 JOUNG er al.: INTEGRAL CYCLE MODE CONTROL TABLE I CHARACTERISTICS OF EACH MODE Tank Circuit Variables Operation Modes Voltage Current Input Power P, Output Power Po p, - Po U,*(k + ) = powering discontinuous - ode -.-- controllable Fig. 4. Diagra of tie doain control schee il vc vo(k + ) = + $(k) - - CO kt/2 + T/2 s kt/2 dt - R (7), T = 2adLC. Fro (6) and (7), the state equation during the powering ode can be given as powering ode powering ode free resonant ode free resonant ode Fig. 5. Typical inductor current and capacitor voltage wavefors in proposed integral cycle ode control. Vc(k+l) Vo(k+l) 2) Free Resonant Mode: In this ode, all conditions are equal to those of the powering ode except V, being set to zero. Therefore, we can easily obtain the following state equation fro (8) and (9): U,*@ + ) -2 L0(k + = [6-6 - **I k - th event (k+l) - th event Fig. 6. Representation of events tions of the four operating odes can be written using the low ripple approxiation ethod as follows. I) Powering Mode: During the kth resonant interval, the output voltage is assued to be nearly constant: that is, U,( t) 3 v0(k). Since the inductor current and the capacitor voltage change sinusoidally, the kth capacitor voltage and output voltage can be evaluated as follows: 3) Regeneration Mode: In this ode, all of the conditions are equal to those of the powering ode except that V, is changed into - V,. Therefore, we can obtain the following equations fro (8) and (9) as

4 86 IEEE TRANSACTIONS ON POWER ELECTRONICS. VOL. 4. NO. I, JANUARY 989 4) Discontinuous Mode: Fro (A4) and U,* (k + ) = U,* ( k) in event k, we obtain i" i,*(k) = 0. A. DC Transfer Function Assue that the interval length of the powering ode is ( T/2) out of the total period n (T/2) as shown in Fig. 7, the dc transfer function can be given as functions of and n. If the new variables are defined as x (k) = v 3k) (64 k, k+l,,... k+,... k+n,,,( events) X*(k) = U,(k) (6b) Fig. 7. Voltage (U,) and current (il) wavefors of tank circuit for duty ratio of /n. and if the converter is operated in the continuous ode, then the th state during the powering ode are found fro (8) as x(k + ) = Ax(k) + sv, ~ ( + k 2) = A2x(k) + (AB + B)Vx x(k + ) = Ax(k) + i 2 = (Ai-'B)Vs (7) During the free resonant ode, the state equations can be evaluated by (0) as x(k + n) = A"-"x(k + ). (9) Fro (7) and (9), we obtain ~ ( + k n) = A "x(~) + A"-" C (A~-~B)V~. In the steady state, it is reasonable to assue that r=l (20) x(k) = x(k + n). (2) Therefore, x (k) is deterined as follows: x(k) = (z - A")-'A"-" I = (A~-~B)v~ (22) Z is an identity atrix. Finally, the transfer function G, is given by G, = v,/v, = Cx(k) = C(Z - A")-' (A"-") C (A-lB). (23) i = l Fro (23), the transfer function G,, is found to be dependent on Q, C/Co,, and n. If Q >>, we see fro Fig. 8 that the output voltage is directly proportional to the duty ratio /n. Note that the output voltage can be controlled in a wide range fro zero to the axiu. However, the output voltage level is discrete and the nuber of discrete levels are n. To control the output voltage by, the step voltage should be given by Vs/n. Thus to reduce the increental step of the output voltage, the nuber of n should be increased. Fig. 9 shows the transfer function for different values of Q and n. We can see fro this curve that the dc transfer function deviates fro the straight line as Q and decreases and n increases. This is due to the discontinuous ode. When Q is sall, the LC tank energy also becoes sall and the tank current fastly reduces to zero and becoes discontinuous ode. When the duty ratio /n is sall, the energy dissipation interval of LC tank becoes longer than the energy storage interval, and thus the discontinuous ode appears. When the discontinuous ode appears, the dc transfer function curve becoes nonlinear. In practice, however, the nonlinearity can be eliinated by feedback control. Fig. 0 shows the dc transfer function as a function of load R for three selected values of for n = 5. We can see that the output voltage does not depend on the load so far as R is saller than a certain value. As R becoes larger than the critical value, the output voltage becoes high because the discontinuous ode appears. Fro this figure, the Q range can be deterined for a given linear region if the load and the output voltage are given. B. Tie Doain Siulation for the Feedback Control Schee Fig. shows the control schee used for the experient. In this figure, the duty ratio of the powering ode and the free resonant ode with (or without) the discontinuous ode is controlled by the tie doain selection of the switch odes fro the difference signal between

5 IOUNG et al.: INTEGRAL CYCLE MODE CONTROL 87 - ~ Fig.. Block diagra of output voltage feedback control schee Fig Gv a (a) i 0 Y " (b) odes Fig. 9. DC transfer function for different Q and n values. (a) n = 0. (b) n = Vo,n Vo,n Fig. 3. Diagra of steady state liit cycle on phase plane for noralized output error voltage V,,.,, V,,,, = (( V,, - Vrcl)/Vrcl), I Gv 06 =3 00 i ~ I r r Cl = wur Fig 0 DC transfer function as function of load and for n = 5 Fig. 4. Wavefors of voltage vs2 and current is? of switch S2. Upper trace (vs2): 50 V/div. Lower trace (isz): 2 A/div. Tie scale: 2 Fs/div.

6 88 IEEE Fig. 5. Wavefors of switch for various duty ratios. (a)l/ 0. (b) 2/ 0. (c) 3/0. (d) 4/0. (e) 5/0. (f ) 6/0. (9) 7/0. (h) 8/0. (i) 9/0. (j) 0/0. Upper trace (inductc sr current il): 5 A/div. Lower trace (switch voltage U=): 00 V/div. Tie scale: 20 ps/div. the output voltage and the coand voltage. Therefore, the output voltage can be effectively regulated by feedback control. In the iner loop, the peak resonant current is liited to a predeterined value to protect the power device during the powering ode operation. Especially in the transient or load dynaic state, the resonant current can be effectively liited to the predeterined value. Fig. 2 shows the siulation result of the control dia-

7 JOUNG et al.: INTEGRAL CYCLE MODE CONTROL 89 gra when the output voltage coand is given as a step function. The output voltage has no overshoot during the transient state and it is regulated during the steady state. The transient response tie is functions of filter size, load condition, and reference peak inductor current Zref. One deerit of such a control schee is the output voltage ripple during steady-state operation due to the liit cycle phenoenon [lo]. Fig. 3 shows the steady-state liit cycle on phase plane. The agnitude of the noralized output error voltage decreases as the capacitance ratio of C and CO decreases as shown in Fig. 3. This phenoenon occurs because of the low frequency control cycles due to the siple control pattern as shown in Fig.. Therefore, such a kind of liit cycle can be reduced further by adapting suitable control pattern or inserting an additional filter at the output side. IV. EXPERIMENTS In this experient, the full-bridge converter shown in Fig. is used. The converter is designed to have the axiu output power P,,, = 250 W at 00 khz of operating frequency. The paraeter values of the converter are given as follows: L = 258 ph C = pf Z = 56 Q CO = 470 pf. The voltage and current wavefors flowing on switch S2 are shown in Fig. 4. This oscillogra shows that the switching stress is low because the switching instants are synchronized to the current zero crossing points. The wavefors of the resonant current and switch voltage are shown in Fig. 5 the duty ratios are / 0, 2/0,. *., 0/ 0, respectively. These oscillogras show that the powering and free resonant odes are repeated alternately and the switching instants are always synchronized to the current zero crossing points. Fig. 6 shows the coparison of the easured data and the theoretical values for n = 0 and Q = (R = 56 Q), Q = 2 (R = 78 Q), and Q = 5 ( R = 3.2 Q), respectively. When Q = 5, the theoretical curve with conduction loss copensation has lower gain than the ideal one, as shown in Fig. 6(c), because of the FET series resistance ( 0.3 Q ), inductor series resistance ( 0.88 Q ), capacitor series resistance (0.79 Q), and rectified diode voltage drop (0.6 V). The transforer is eliinated in this experient. The easured values are nearly coincided with the theoretical values when the conduction losses are considered. In the light load ( Q = and Q = 2) condition, the conduction loss ter is sall, and the easured values nearly coincide with the theoretical ones, as shown in Fig. 6(a) and (b). Fig. 7 shows an experiental result of the transient response when the output voltage coand is given by a Gv theoretical value I experiental data +- theoretical value I experiental data / Gv (b) I I (C) Fig. 6. Coparisons of dc transfer functions between theoretical and experiental results. (a) Q = ( R = 56 0). (b) Q = 2 (R = 78 0). (c) Q = 5 (R = 3.2 0). step function. The result shows good regulation with no overshoot. V. CONCLUSION A new control schee for the SRC is suggested in this paper. The SRC operates at the circuit resonant frequency and the switching instants are always synchronized to the current zero crossing points. Therefore, the ajor device

8 90 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL 4. NO. I, JANUARY 989 Free Resonant Mode: 0-0 C L L o CO COR (A2) Fig. 7. Transient response of output voltage U,, for coand voltage Vrei. Upper trace (/*<,): 20 V/div. Lower trace: V,,,. Tie scale: 2 s/div. loss is conduction loss and the device capability is fully utilized under high-frequency operation. Converter switching frequency is not liited by the switching loss but by the device turn-on/off tie. The output voltage control is achieved by varying the interval length ratio of the switch odes. This control schee is analyzed in detail, and the characteristics are verified through the experient. The properties of the proposed schee are suarized as follows. ) An SRC always guarantees any advantages, such as zero device switching current, low switching loss, reliable high-frequency operation, and low EMI, because it always operates at optial condition. 2) For high Q, the dc transfer function is linear to the duty ratio of the powering ode and free resonant ode, and it is independent of the load. Despite a light load condition, the output voltage can be controlled fro zero to the axiu. 3) Two deerits of this schee are discrete output voltage levels and the output voltage ripple due to the liit cycle phenoenon. However, the output voltage can be controlled by feedback schee, and the output voltage ripple can be reduced further by a suitable control pattern. APPENDIX Fro Fig. 3, the differential equations of the syste for each switch ode are derived as follows. Powering Mode: u,*(t) = u,(t) u,*(t) = -v,(t) Regeneration Mode: = I u,*(t) = u,(t) v,*(t) = -u,(t)! + $(t) = il(t) for S4, D2 on iz(t) = -il(t) for S2, D4 on. 0-0 C 0 -- L L -- 0 [ Discontinuous Mode: K $(t) = i,(t) ford2, D3 on i;(t) = -il(t) for D, D4 on. u,*(t) = u,(t) il*(t) = i,(t) vf(t) = = -u,(t) if(t) = il,(t) for SI, S4 on $(t) = -i,(t) for S2, S3 on. REFEREN&S [l] S. W. H. De Hann, A new integral pulse odule for the seriesresonant converter. IEEE Trans. Ind. Electron., vol. IE-3, no. 3, pp , Aug [2] R. L. Steigenvald, High frequency resonant transistor dc-dc converters, IEEE Trans. Ind. Electron., vol. IE-3, no. 2, pp. 8-92, May F. C. Schwarz, An iproved ethod of resonant current pulse odulation for power converters, IEEE Trans. Ind. Electron., Contr. Intru., vol. IECI-23, no. 2, pp. 33-4, May 976. [4] R. J. King and T. A. Stuart, A large signal dynaic siulation for the series resonant converter, IEEE Trans. Aerosp. Electron. Syst., vol. AES-9, no. 6, pp , Nov [5] -, Inherent overload protection for the series resonant con-

9 IOUNG et al.: INTEGRAL CYCLE MODE CONTROL 9 verter, IEEE. Trans. Aerosp. Electron. Syst., vol. AES-9, no. 6, pp , NOV D. M. Divan, Design considerations for very high frequency resonant ode dcldc converters, in Conf. Rec. 986 leee Industry Applications Society Annu. Meering, pp V. Vorperian and S. Cuk, Sall signal analysis of resonant converters, in Proc. 983 leee PESC Rec., pp , June 983. A. S. Kislovski, A contribution for steady-state odeling of halfbridge series-resonant power cells, IEEE Trans. Power Electron., vol. PE-I, no. 3, pp. 6-66, July 986. [9 I. J. Pitel, Phase-odulated resonant power conversion techniques for high frequency link inverters, leee Trans. lnd. Appl., vol.-ia- 22, no. 6, pp , Nov [IO] K. J. Kuran, Feedback Control Theory and Design. Asterda, The Netherlands: Elsevier, 984. Gyu B. Joung was born in Korea in 96. He received the B.S. degree fro Ajou University, Suwon, Korea, and the M.S. degree fro Korea Advanced Institute of Science and Technology (KAIST), Seoul, in 984 and 986, respectively, both in electrical engineering. He is presently working towards the Ph.D. degree at KAIST. His research interests include all aspects of power electronics, particularly converters, and odeling of power converters. Chun T. Ri was born in Korea on Deceber 7, 962. He received the B.S. degree fro Kuoh Institute of Technology, Kui, Korea, and the M.S. degree fro Korea Advanced Institute of Science and Technology (KAIST), Seoul, in 985 and 987, respectively, both in electrical engineering. He is presently working towards the Ph.D. degree in the sae division. His research areas are the odeling of linear switching systes, the control of resonant converters, the design of SMPS and the design of linear IC s. Gyu H. Cho was born in Korea on April 9, 953. He received the M.S. and Ph.D. degrees fro Korea Advanced Institute of Science and Technology (KAIST), Seoul, in 977 and 98, respectively. During , he joined the Electronic Technology Division of Westinghouse R&D Center, Pittsburgh, PA, he worked on unrestricted frequency changer systes and inverters. Since 984, he has been an AssistantIAssociate Professor in the Electrical Engineering Departent of KAIST. His research interests are in the area of static power converters and drives, resonant converters and integrated linear electronic circuit design.