' r A MICROSTRIP-COUPLED SLOT-LOOP ANTENNA FOR INTEGRATED RECEIVERS IN THE MILLIMETER-WAVE BAND

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1 A MICROSTRIP-COUPLED SLOT-LOOP ANTENNA FOR INTEGRATED RECEIVERS IN THE MILLIMETER-WAVE BAND Pablo Otero, Alejandro Alvarez-Melcon, Jean-Françoi Zurcher, and Juan R. Moig Laboratoire d Electromagnetime et d Acoutique ( LEMA) Ecole Polytechnique Federale de Lauanne ( EPFL) CH-05 Lauanne, Switzerland Recei ed 3 December 997 ABSTRACT: A quare-loop lot antenna, printed at the back urface of an ended hemipherical len, i examined a a candidate for millimeter-wa e ( mm-wa e) integrated-circuit Schottky-diode mixer. The loop lot i etched in a microwa e ubtrate, and coupled to a microtrip line ection printed on the other ide of the ubtrate. A Schottky barrier diode hunted to ground i the mixing de ice. The input impedance of the antenna ha been optimized to conjugate match the Schottky diode. To deign and optimize the antenna, an efficient full-wa e method of moment ( MoM) technique, which ol e the mixed potential integral equation ( MPIE) in the tratified medium, ha been de eloped. Pattern ha e been computed uing the phyical optic approximation and diffraction theory. The propoed configuration how ery wideband performance. Simulation reult compare well with meaurement. The propoed configuration i uitable to be caled down for operation in the ubmillimeter-wa e ( ubmm-wa e) band. 998 John Wiley & Son, Inc. Microwave Opt Technol Lett 8: 9 95, 998. Key word: mixed potential integral equation ( MPIE ); method of moment ( MoM ); planar antenna; lot antenna; millimeter wa e. INTRODUCTION An increaed interet in planar integrated receiver for the mm- and ubmm-wave band ha been oberved recently. The main reaon i the deire to replace the waveguide-baed technology currently ued in paceborne application. In thi cont, previou work ha hown the feaibility of uing uniplanar technology Žcoplanar tranmiion-line-fed lot antenna., coupled to ended hemipherical dielectric lene, and integrating a mixing device. Thi deign i ought to operate a a front-end feed for reflector-type antenna. However, when caled to the ubmm-wave band, the location of the mixing device, a Schottky barrier diode, integrated with the coplanar waveguide, reult in an aymmetric configuration with too high idelobe level 2. To enure the ymmetry and to iolate the diode from the radiation emipace, an antenna configuration, already ued in the 980 in C- and X-band, ha been tried 3. Figure, where all unit are millimeter, how the propoed deign. The Schottky diode i only ymbolically repreented, a well a the intermediate frequency Ž IF. filter and the ended hemipherical len. The IF filter i a nine-ection, quarter-wavelength tranformer, low-pa filter with a cutoff frequency of about 40 GHz. The low relative permittivity ubtrate i mounted on the plane urface of the dielectric len. The radiu of the len i much larger than the one hown in Figure. The utilized ubtrate i RT Duroid 5870, r 2.33, tan 0.002, and h mm 4. The face which i in contact with the len ha a conductor plane where a quare lot loop ha been etched. On the other ubtrate face, a microtrip line ection couple to the lot loop at one end and feed the mixing diode at the other. The IF ignal i racted from the ame diode terminal, a hown in Figure. For our experi- ment, the local ocillator Ž LO. ignal ha been injected through the ame filter, o that ubharmonic mixing ha been performed. For a poible caled prototype operating at ubmm-wave frequencie, quaioptical LO ignal injection can alo be ued. The two main problem of the original deign, when uch high frequencie are ued, are the appearance of urface wave and the too high loe in microtrip line. The former i controlled by an appropriate choice of the ubtrate thickne and permittivity. A hown by Gupta et al. 5, the cutoff frequency fc of the firt higher order mode in a microtrip line i given approximately by 300 f Ž. c Ž 2W 0.8h. ' r where fc i in gigahertz, r i the ubtrate relative permittiv- ity, W i the microtrip width in millimeter, and h i the ubtrate thickne, alo in millimeter. For the RT Duroid 5870 ubtrate ued in the preent deign, to have fop f c, where f i the operating frequency, we need op 98 W Ž mm. 0.. Ž 2. f Ž GHz. op The width of the deigned microtrip line atify thi condition. Equation Ž. how that the maller r and h are, the le power conveyed into urface wave. Thi i a deign advantage becaue the larger the effective relative permittivity ratio at both ide of the lot loop plane, the larger the front-to-back power ratio of the antenna 6. The econd problem of the deign i due to the high ohmic loe of microtrip line at mm- and ubmm-wave frequencie. The preent olution ue the hortet poible length of microtrip at thee frequencie, a can be een in Figure. Beyond the diode, only the IF ignal propagate. A very important feature of the deign i that the lot loop perimeter and the microtrip length are quite independent. Thi fact ha two main advantage: firt, diode matching can be carried out with little concern about the lot loop ize and, econd, the frequency band can be widened compared to previou uniplanar deign 7. Indeed, it i poible that the antenna itelf conjugate matche the diode. Thi technique ha already been ued with ucce in the previouly mentioned deign. The ought input impedance of the antenna i therefore not 50, but rather Z diode. A typical Schottky diode impedance at 65 GHz, Z Ž 50 j50. diode, ha been ued. 2. ANALYTIC METHODS 2.. Theoretical Model. For the analyi of the radiating tructure propoed, a tandard integral equation formulated in the pace domain ha been derived. In the lot interface, Love equivalence principle i ued, and uitable magnetic current are defined to help impoe the continuity condition for the tranvere electric and magnetic field. The elf-interaction Ž electric electric and magnetic magnetic. are olved with the aid of the vector and calar potential Green function, epecially well uited due to their weak ingular behavior when the ource oberver ditance vanihe 8. All other interaction, nonexhibiting ingularitie, are treated directly uing the electric and magnetic field Green func- MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 8, No. 2, June

2 Figure Propoed deign. All unit in millimeter tion, thu leading to a very compact analytical treatment of the problem. Multilayered media potential and field Green function are firt obtained in the pectral domain uing the tranvere equivalent tranmiion line network repreentation of the tructure along the tratified axi. Following thi formalim, the pectral-domain Green function aociated to electric and magnetic dipole are eaily obtained by computing equivalent voltage and current in the reulting tranmiion line network 9. Once the pectral-domain Green function are computed, their patial-domain counterpart can be obtained through the well-known Sommerfeld integral, and the efficient numerical technique decribed in 8 have been ued. After efficient evaluation of the pace-domain Green function, the impoition of the boundary condition for the field in the pecific tructure hown in Figure yield the following vector coupled ytem of integral equation on the unknown induced electric Ž J. and magnetic Ž M. current: Že. E j HH GA J HH GV J j Se Se 0 j HH GFu GFl M S m ž / HH EM S m G M Ž 3. HH Ž G G. Wu Wl M j Sm HH HJ S e G J Ž 4. Že. where E i the exciting or impreed field due to the generator applied to the trip line, Se i the urface of the trip line, and Sm i the urface of the lot loop. The ubcript u and l of Green function tand for above and below, repectively, the conductor plane where the lot are etched Numerical Method. For the olution of the vector-coupled ytem of integral equation previouly derived, a Galerkin method of moment Ž MoM. algorithm baed on ubectional rooftop function ha been developed. The main difficulty in the implementation of the propoed approach i the efficient numerical evaluation of the fourfold overlapping integral between the patial-domain Green function and the rooftop bai and tet function. In the preent work, the cro-correlation integral of the rooftop function have been ued together with the ymmetry propertie of Green function to ytematically reduce all fourfold integral into double integral, thu coniderably reducing the computational effort. For the ingular cae, a further tranformation to the polar plane i performed in order to analytically ract the Ž. ingular behavior of the potential Green function. Thank to the analytical raction procedure applied, the remainder function behave very moothly, and tandard numerical integration technique can be applied, obtaining good accuracy with a limited number of integration point. In Figure 2, we preent the value obtained for a typical ingular MoM element a a function of the number of integration point. The computation i performed for the tructure hown in Figure and for the frequency of 70 GHz. We can oberve that good convergence i obtained with only Ž 4 4. integration point, the relative error being maller than 0.%. The developed oftware tool take only 4 per frequency point on an HP72 80 platform to complete the analyi of the antenna tructure hown in Figure, when only one cell i ued in the tranvere direction of the trip line and loop. Even when, looking for increaed accuracy, the dicretization i performed with three cell in the tranvere direction, the oftware till only take 8 per frequency point on the ame platform. Thee reult clearly indicate that the developed oftware can be ued a a real tool for the efficient optimization of thi type of antenna Len Analyi. The field in the far region from the len may be calculated by mean of the Love equivalence principle, which allow ubtitution of the len by a perfect conduc- 92 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 8, No. 2, June 5 998

3 Figure 2 Relative and abolute value of a typical ingular MoM element a a function of the number of integration point. Numerical tet i performed for the tructure hown in Figure and at a frequency of 70 GHz tor and equivalent electric and magnetic urface current. Phyical optic approximation, baed on image theory, allow the electric current to be conidered a hort circuited. The equivalent magnetic urface current i proportional to the electric field on the outer urface of the len. Thi field i computed accepting that the len radiu i much larger than the wavelength in the len dielectric, o that Frenel tranmiion coefficient may be ued. 3. RESULTS All imulated and meaured reult hown below correpond to the integrated antenna mixer depicted in Figure. Figure 3 how the module of the generalized load reflection coefficient, a defined by Collin, in decibel, that i, ant Zdiode Z 20 log. Ž 5. L Z Z diode ant Simulation preented in Figure 3 correpond to two different mehing reolution. The olid line i obtained when lot and trip width contain a ingle cell. To obtain the dotted line, width i divided into three cell. No ignificant change are obtained when further increaing the number of tranvere cell, which verifie the convergence of the numerical method. In addition, Figure 3 how the onet of a double-reonance phenomenon, expected from the dicuion in Section. One reonance i related to the lot loop ize, and the other one i determined by the coupling microtrip ection length. Bandwidth i then larger than in uniplanar deign. For a reflection coefficient of 0 db, the bandwidth i about 20%, which i better than previou deign 7. Figure 4 and 5 how the imulated and meaured radiation pattern in the E- and H-plane, repectively, at 65 GHz. Figure 6 how the E-plane pattern at 70 GHz. The H-plane pattern barely change with frequency, and therefore it i not hown. All of thee pattern are for an enion length of L 8.5 mm, which i between the hyperhemiphere Ž L 7.2 mm. and the yntheized ellipoid Ž L 9.6 mm.. Meaured pattern how reaonable agreement with Figure 3 Generalized load reflection coefficient between antenna and diode veru frequency for two different grid the imulated pattern. Moreover, the H-plane pattern how an remely low idelobe level, and it change very little with frequency. The E-plane idelobe level i not a good a the H-plane one. However, it i better than previou deign 2 and improve with frequency, a can be oberved in Figure 6. Even though the loop i perfectly ymmetric, the equivalent magnetic current that the microtrip create in the lot are not uniformly ditributed on both ide of the loop becaue feeding i done only on one ide. Thi fact i conidered by our in-houe imulation tool, but the predicted aymmetry i noticeably maller than the meaured one, a oberved on the left idelobe of the pattern in Figure 4. The left idelobe i therefore probably due to uncontrolled puriou propagation phenomena generated by the diode. Prediction for directivity of the len antenna at 65 GHz veru enion length are hown in Figure 7. An inflection MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 8, No. 2, June

4 Figure 4 Simulated and meaured E-plane pattern at 65 GHz Figure 6 Simulated and meaured E-plane pattern at 70 GHz Figure 5 Simulated and meaured H-plane pattern at 65 GHz point i oberved at the hyperhemipherical poition. The maximum directivity i obtained cloe to the yntheized ellipoidal point, a expected. The len antenna directivity, a well a the input impedance previouly hown, cannot be directly meaured, o far, at our laboratory. Intead, the iotropic converion gain Ž ICG. 2 ha been meaured, which i hown in Figure 8. The ICG i defined a the power delivered to the IF load from a copolarized incident plane wave by an antenna mixer or an antenna amplifier mixer, divided by the power that would have been received by an iotropic antenna with a matched load. The ICG i a magnitude that group antenna and mixer parameter uch a the antenna gain Ž directivity and loe., the antenna mixing device mimatch, the mixer converion lo, and the IF filter inertion lo. The ICG i a figure of merit of a frequency converter which may be ued when the mentioned parame- Figure 7 Simulated len antenna directivity veru enion length at 65 GHz ter cannot be directly meaured. In our prototype, the ICG performance i imilar to that obtained in previou deign, and good agreement i obtained with predicted bandwidth in Figure CONCLUSIONS A type of quare-loop lot antenna, coupled to a microtrip line ection, ha been examined a a candidate for a millimeter-wave integrated-circuit Schottky-diode mixer. The antenna ha been optimized to match the typical input impedance of uch diode in the millimeter-wave band. Optimization ha been performed by mean of an efficient MoM analyi oftware tool pecifically developed for thi tudy. The antenna how a very wide frequency band, clean and rotationally ymmetric pattern, and a good compromie be- 94 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 8, No. 2, June 5 998

5 0. G. V. Eleftheriade, J. F. Zurcher, and J. R. Moig, Pattern and Efficiencie of Slot-Fed mm-wave Gla-Ceramic Subtrate Len Antenna, Proc. ESA ESTEC Workhop mm-wa e Tech. Appl., Noordwijk, The Netherland, Nov. 995, pp R. E. Collin, Foundation for Microwa e Engineering, 2nd ed., McGraw-Hill, New York, G. M. Rebeiz, Millimeter-Wave and Terahertz Integrated Circuit Antenna, Proc. IEEE, Nov. 992, pp John Wiley & Son, Inc. CCC Figure 8 Iotropic converion gain veru frequency THE INPUT IMPEDANCE OF A THIN COAXIAL-LINE-FED PROBE IN A THICK SEMI-INFINITE COAXIAL LINE Z. M. Xie, Edward K. N. Yung, and R. S. Chen Department of Electronic Engineering City Univerity of Hong Kong Hong Kong Recei ed 5 December 997 tween directivity and Gauian coupling efficiency, depending on the hemipherical len enion length. Simulation reult agree well with meaurement. ACKNOWLEDGMENTS The author are grateful to Prof. F. Gardiol for hi encouraging upport. Pablo Otero i upported by the Spanih National Science and Technology Agency, CICYT Ž Comiion Interminiterial de Ciencia y Tecnologıa.. REFERENCES. D. F. Filipovic, S. S. Gearhart, and G. M. Rebeiz, Double-Slot Antenna on Extended Hemipherical and Elliptical Silicon Dielectric Lene, IEEE Tran. Microwa e Theory Tech., Oct. 993, pp P. Otero, G. V. Eleftheriade, and J. R. Moig, Slot-Loop Antenna on Subtrate Lene for Submillimeter-Wave Open Structure Mixer, Proc. 20th ESTEC Antenna Workhop, Noordwijk, The Netherland, June 997, pp J. P. Daniel, E. Penard, and C. Terret, Deign and Technology of Low-Cot Printed Antenna, in Handbook of Microtrip Antenna, J. R. Jame and P. S. Hall, Ed., Peter Peregrinu, London, 989, Vol., Chap., pp Roger Corporation, 00 South Rooevelt Ave., Chandler, AZ USA. 5. K. C. Gupta, R. Garg, I. Bahl, and P. Bhartia, Microtrip Line and Slotline, 2nd ed., Artech Houe, Norwood, MA, D. B. Rutledge, D. P. Neikirk, and D. P. Kailingam, Integrated-Circuit Antenna, in Infrared and Millimeter Wa e, K. J. Button, Ed., Academic Pre, New York, 983, Vol. 0, Chap., pp P. Otero, G. V. Eleftheriadeand, and J. R. Moig, Integrated Modified Rectangular Loop Slot Antenna on Subtrate Lene for mm- and Submm-Wave Frequencie Mixer Application, ubmitted to IEEE Tran. Antenna Propagat., Sept J. R. Moig, Integral Equation Technique, in Numerical Technique for Microwa e and Millimeter-Wa e Pai e Structure, T. Itoh, Ed., John Wiley & Son, New York, 989, Chap. 3, pp K. A. Michalki and J. R. Moig, Multilayered Media Green Function in Integral Equation Formulation, IEEE Tran. Antenna Propagat., Mar. 997, pp ABSTRACT: The Green function for determining the electromagnetic field in a emi-infinite coaxial line due to a radially directed infiniteimally hort electric dipole are deri ed. The coaxial line i horted at one end and terminated at a perfectly matched load at the other, and the probe i connected to the inner conductor of thi emi-infinite coaxial line and i fed by a maller coaxial line through a hole on the urface of the outer conductor. Baed on thee Green function, TEM, TE, and TM mode are conidered in our analyi, and the input impedance of a mall probe in front of the plunger i calculated. A a reult, our theoretical reult agree ery well with the experimental data o er a frequency band for e eral ditance between the probe and the plunger. The comparion i alo made between our reult and thoe calculated by uing traditional tranmiion line theory, which treat the probe a a tranmiion line branch. 998 John Wiley & Son, Inc. Microwave Opt Technol Lett 8: 95 00, 998. Key word: Green function; input impedance; coaxial line; mode expanion I. INTRODUCTION In ome application uch a antenna feeding or microwave meaurement, the connection mut be made between a thick coaxial line and a thin coaxial line. Uually, they are connected longitudinally, which i the cae in a coaxial line tep. However, when the difference between the diameter of the two coaxial line i very large, many tep impedance tranformer are needed to achieve good matching over a frequency band. In thi way, the tranition ha to be long enough, epecially for a low RF band. Alternatively, the maller coaxial line i ued a the feeding of a probe connected to the inner conductor of the thick coaxial line, jut like a T-junction with one end horted. The plunger can act a a tuning component. The geometry of the connection i hown in Figure. In a low enough frequency, the probe only excite the TEM mode in the large coaxial line, and the T-junction can be viewed a a tranmiion line branch. Therefore, the input impedance can be calculated by uing tranmiion line theory or a Smith chart. However, a the frequency increae, the effect of a high-order mode excited by the probe cannot be ignored. The above method i not rigorou for engineering deign, epecially in the cae where MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 8, No. 2, June