1.3W, Filterless, Stereo Class D Audio Power Amplifier

Transcription

1 9-3457; Rev 2; 2/7.3W, Filterless, Stereo Class D Audio General Description The stereo Class D audio power amplifier provides Class AB amplifier audio performance with the benefits of Class D efficiency, eliminating the need for a heatsink while extending battery life. The delivers up to.3w per channel into an 8Ω load while offering 87% efficiency. Maxim s next-generation, low- EMI modulation scheme allows the amplifier to operate without an external LC filter while still meeting FCC EMI emission levels. The offers two modulation schemes: a fixed-frequency (FFM) mode, and a spread-spectrum (SSM) mode that reduces EMI-radiated emissions. The oscillator can be synchronized to an external clock through the input, allowing synchronization of multiple Maxim Class D amplifiers. The sync output (_OUT) can be used for a master-slave application where more channels are required. The features a fully differential architecture, a full bridge-tied load (BTL) output, and comprehensive click-and-pop suppression. The device features internally set gains of db, 6dB, 2dB, and 8dB selected through two gain-select inputs, further reducing external component count. The features high 8dB PSRR, less than.% THD+N, and SNR in excess of 88dB. Short-circuit and thermal-overload protection prevent the device from being damaged during a fault condition. The is available in 24-pin thin QFN-EP (4mm x 4mm x.8mm), and 2-bump UCSP (2mm x 2.5mm x.6mm) packages. The is specified over the extended -4 C to +85 C temperature range. Cellular Phones Notebooks Handheld Gaming Consoles Docking Stations MP3 Players Applications Features Patented Spread-Spectrum Modulation Lowers Radiated Emissions Single-Supply Operation (2.5V to 5.5V).3W Stereo Output (8Ω, = 5V, THD+N = %) No LC Output Filter Required 87% Efficiency (, P OUT = mw) Less Than.% THD+N High 8dB PSRR Fully Differential Inputs Integrated Click-and-Pop Suppression Typical Low Quiescent Current (9mA) Typical Low-Power Shutdown Mode (.µa) Short-Circuit and Thermal-Overload Protection Available in Thermally Efficient, Space-Saving Packages 24-Pin Thin QFN-EP (4mm x 4mm x.8mm) 2-Bump UCSP (2mm x 2.5mm x.6mm) INR+ Ordering Information PART TEMP RANGE PIN- PACKAGE RIGHT MODULATOR AND H-BRIDGE PKG CODE EBP-T -4 C to +85 C 2 UCSP-2 B2- ETG+ -4 C to +85 C 24 TQFN-EP* T Denotes lead-free package. *EP = Exposed paddle. Block Diagram Pin Configurations appear at end of data sheet. GAIN GAIN2 GAIN INL+ INR- INL- LEFT MODULATOR AND H-BRIDGE OSCILLATOR _OUT UCSP is a trademark of Maxim Integrated Products, Inc. Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at , or visit Maxim s website at

2 ABSOLUTE MAXIMUM RATINGS to GND...6V to P...-.3V to +.3V P to PGND...6V GND to PGND...-.3V to +.3V All Other Pins to GND...-.3V to ( +.3V) Continuous Current In/Out of P, PGND, OUT_...±8mA Continuous Input Current (all other pins)...±2ma Duration of OUT_ Short Circuit to GND or P...Continuous Duration of Short Circuit Between OUT+ and OUT-...Continuous Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS Continuous Power Dissipation (T A = +7 C) 2-Bump UCSP (derate mw/ C above +7 C)...8mW 24-Pin Thin QFN (derate 2.8mW/ C above +7 C) mW Junction Temperature...+5 C Operating Temperature Range...-4 C to +85 C Storage Temperature Range C to +5 C Bump Temperature (soldering) Reflow C Lead Temperature (soldering, s)...+3 C ( = P = SHDN = 3.3V, GND = PGND = V, = V (FFM), gain = 6dB (GAIN =, GAIN2 = ), R L connected between OUT+ and OUT-, R L =, T A = T MIN to T MAX, unless otherwise noted. Typical values are at T A = +25 C.) (Notes, 2) GENERAL PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS Supply Voltage Range Inferred from PSRR test V = 3.3V, per channel Quiescent Current I DD = 5V, per channel 6.3 Shutdown Current I SHDN. µa Common-Mode Rejection Ratio CMRR f IN = khz 66 db Input Bias Voltage V BIAS V Turn-On Time t ON 4 ms T A = +25 o C ± ±3 Output Offset Voltage V OS T MIN < T A < T MAX ±55 = 2.5V to 5.5V, V IN = V 6 8 Power-Supply Rejection Ratio PSRR mv P-P ripple, f RIPPLE = 27Hz 72 V IN = V f RIPPLE = 2kHz 5 Output Power (Note 3) P OUT THD+N = %, T A = +25 o C Total Harmonic Distortion Plus Noise (Note 3) THD+N 46 = 3.3V RL = 4Ω 75 = 5V 3 R L = 4Ω 22 (P OUT = 3mW), f = khz.8 R L = 4Ω (P OUT = 4mW), f = khz.5 Signal-to-Noise Ratio SNR V OUT = V RMS A-weighted BW = 22Hz FFM 86 to 22kHz SSM 86 FFM 88.5 SSM 88.5 = GND = unconnected Oscillator Frequency f OSC = 2 ±6 Minimum On-Time t MIN 2 ns Frequency Lock Range f 6 khz 2 ma mv db mw % db khz

3 ELECTRICAL CHARACTERISTICS (continued) ( = P = SHDN = 3.3V, GND = PGND = V, = V (FFM), gain = 6dB (GAIN =, GAIN2 = ), R L connected between OUT+ and OUT-, R L =, T A = T MIN to T MAX, unless otherwise noted. Typical values are at T A = +25 C.) (Notes, 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS _OUT Capacitance Drive C _OUT pf Bridge-tied capacitance 2 Capacitive Drive C L Single ended 4 Click-and-Pop Level K CP A-weighted, 32 samples Peak reading, TH D + N = % per second (Note 4) Efficiency Input Resistance Gain Channel-to-Channel Gain Tracking η R IN A V Into shutdown 66.6 Out of shutdown = 3.3V, P OUT = 5mW per channel, f IN = khz, = 5V, P OUT = mw per channel, f IN = khz, GAIN =, GAIN2 = GAIN =, GAIN2 = 25 GAIN =, GAIN2 = 37.4 GAIN =, GAIN2 = 5 GAIN =, GAIN2 = 8 GAIN =, GAIN2 = 2 GAIN =, GAIN2 = 6 GAIN =, GAIN2 = pf db % kω db % Crosstalk DIGITAL INPUTS (SHDN,, GAIN, GAIN2) L to R, R to L, f = khz,, P OUT = 3mW 7 db Input-Voltage High V INH 2 V Input-Voltage Low V INL.8 V Input Leakage Current (SHDN, GAIN, GAIN2) Input Leakage Current () DIGITAL OUTPUTS (_OUT) V IN = GND, normal operation -5-7 V IN =, normal operation 2 25 ± µa Output-Voltage High V OH I OH = 3mA, = 3.3V 2.4 V Output-Voltage Low V OL I OL = 3mA.8 V Note : All devices are % production tested at +25 C. All temperature limits are guaranteed by design. Note 2: Testing performed with a resistive load in series with an inductor to simulate an actual speaker load. For R L = 4Ω, L = 33µH. For, L = 68µH. Note 3: When driving speakers below 4Ω with large signals, exercise care to avoid violating the absolute maximum rating for continuous output current. Note 4: Testing performed with 8Ω resistive load in series with 68µH inductive load connected across the BTL output. Mode transitions are controlled by SHDN. K CP level is calculated as: 2 x log[(peak voltage under normal operation at rated power level) / (peak voltage during mode transition, no input signal)]. Units are expressed in db. 3 µa

4 Typical Operating Characteristics ( = P = SHDN = 3.3V, GND = PGND = V, = (SSM), gain = 6dB (GAIN =, GAIN2 = )). TOTAL HARMONIC DISTORTION PLUS NOISE vs. FREQUENCY = 5V R L = 4Ω toc TOTAL HARMONIC DISTORTION PLUS NOISE vs. FREQUENCY = 5V toc2 TOTAL HARMONIC DISTORTION PLUS NOISE vs. FREQUENCY = 3.3V R L = 4Ω toc3 THD+N (%) OUTPUT POWER = 6mW THD+N (%) OUTPUT POWER = 5mW THD+N (%) OUTPUT POWER = 6mW... OUTPUT POWER = mw OUTPUT POWER = 3mW. k k k OUTPUT POWER = mw OUTPUT POWER = 25mW. k k k OUTPUT POWER = mw OUTPUT POWER = 3mW. k k k TOTAL HARMONIC DISTORTION PLUS NOISE vs. FREQUENCY = 3.3V toc4 TOTAL HARMONIC DISTORTION PLUS NOISE vs. FREQUENCY = 5V P OUT = 8mW toc5 TOTAL HARMONIC DISTORTION PLUS NOISE vs. OUTPUT POWER = 5V R L = 4Ω toc6 THD+N (%) OUTPUT POWER = 4mW THD+N (%) FFM THD+N (%) f IN = khz. OUTPUT POWER = mw OUTPUT POWER = 25mW. k k k. SSM. k k k.. f IN = khz f IN = 2kHz THD+N (%). TOTAL HARMONIC DISTORTION PLUS NOISE vs. OUTPUT POWER = 5V f IN = khz toc7 THD+N (%) TOTAL HARMONIC DISTORTION PLUS NOISE vs. OUTPUT POWER = 3.3V R L = 4Ω f IN = khz f IN = khz toc8 THD+N (%) TOTAL HARMONIC DISTORTION PLUS NOISE vs. OUTPUT POWER = 3.3V f IN = khz toc9. f IN = 2kHz f IN = khz. f IN = 2kHz. f IN = khz f IN = 2kHz OUTPUT POWER (mw) 4

5 Typical Operating Characteristics (continued) ( = P = SHDN = 3.3V, GND = PGND = V, = (SSM), gain = 6dB (GAIN =, GAIN2 = )). THD+N (%).. TOTAL HARMONIC DISTORTION PLUS NOISE vs. OUTPUT POWER = 5V f IN = khz SSM FFM toc EFFICIENCY (%) EFFICIENCY vs. OUTPUT POWER R L = 4Ω 2 = 5V f IN = khz toc EFFICIENCY (%) EFFICIENCY vs. OUTPUT POWER R L = 4Ω 2 V DD = 3.3V f IN = khz toc OUTPUT POWER vs. SUPPLY VOLTAGE R L = 4Ω A V = 2dB f IN = khz toc OUTPUT POWER vs. SUPPLY VOLTAGE A V = 2dB f IN = khz toc OUTPUT POWER vs. LOAD RESISTANCE = 5V f IN = khz toc THD+N = % THD+N = % THD+N = % THD+N = % THD+N = % THD+N = % SUPPLY VOLTAGE (V) SUPPLY VOLTAGE (V) LOAD RESISTANCE (Ω) 2..6 OUTPUT POWER vs. LOAD RESISTANCE f IN = khz toc POWER-SUPPLY REJECTION RATIO vs. FREQUENCY V RIPPLE = mv P-P toc COMMON-MODE REJECTION RATIO vs. FREQUENCY V CM = mv P-P toc8.2.8 THD+N = % THD+N = % PSRR (db) CMRR (db) = 5V = 3.3V LOAD RESISTANCE (Ω) - k k k - k k k 5

6 Typical Operating Characteristics (continued) ( = P = SHDN = 3.3V, GND = PGND = V, = (SSM), gain = 6dB (GAIN =, GAIN2 = )). CROSSTALK (db) CROSSTALK vs. FREQUENCY P OUT = 3mW LEFT TO RIGHT -3 k k k RIGHT TO LEFT toc9 CROSSTALK (db) CROSSTALK vs. INPUT AMPLITUDE f IN = khz RIGHT TO LEFT LEFT TO RIGHT INPUT AMPLITUDE (db) toc2 OUTPUT MAGNITUDE (dbv) OUTPUT FREQUENCY SPECTRUM FFM MODE V OUT = -6dBV f = khz UNWEIGHTED toc2 OUTPUT MAGNITUDE (dbv) OUTPUT FREQUENCY SPECTRUM FFM MODE V OUT = -6dBV f = khz A-WEIGHTED toc k k 5k 2k -4 5k k 5k 2k OUTPUT MAGNITUDE (dbv) OUTPUT FREQUENCY SPECTRUM SSM MODE V OUT = -6dBV f = khz UNWEIGHTED toc23 OUTPUT MAGNITUDE (dbv) OUTPUT FREQUENCY SPECTRUM SSM MODE V OUT = -6dBV f = khz A-WEIGHTED toc k k 5k 2k -4 5k k 5k 2k 6

7 Typical Operating Characteristics (continued) ( = P = SHDN = 3.3V, GND = PGND = V, = (SSM), gain = 6dB (GAIN =, GAIN2 = )). OUTPUT MAGNITUDE (db) k WIDEBAND OUTPUT SPECTRUM (FFM MODE) RBW = khz INPUT AC GROUNDED k k toc25 M OUTPUT MAGNITUDE (db) k WIDEBAND OUTPUT SPECTRUM (SSM MODE) RBW = khz INPUT AC GROUNDED k k toc26 M SHDN OUTPUT TURN-ON/TURN-OFF RESPONSE toc27 2V/div V 25mV/div SUPPLY CURRENT (ma) SUPPLY CURRENT vs. SUPPLY VOLTAGE BOTH CHANNELS SSM FFM toc28 ms/div SUPPLY VOLTAGE (V) SHUTDOWN CURRENT vs. SUPPLY VOLTAGE SHUTDOWN CURRENT (µa) BOTH CHANNELS toc SUPPLY VOLTAGE (V) 7

8 TQFN PIN UCSP NAME FUNCTION A2 SHDN Active-Low Shutdown. Connect to for normal operation. 2 B3 Pin Description Frequency Select and External Clock Input. = GND: Fixed-frequency mode with f S = khz. = Unconnected: Fixed-frequency mode with f S = 4kHz. = : Spread-spectrum mode with f S = 2kHz ±6kHz. = Clocked: Fixed-frequency mode with f S = external clock frequency. 3, 8,, 6 N.C. No Connection. Not internally connected. 4 A3 OUTL+ Left-Channel Amplifier Output Positive Phase 5, 4 A4, D4 P H-Bridge Power Supply. Connect to. Bypass with a.µf capacitor to PGND. 6, 3 B4, C4 PGND Power Ground 7 A5 OUTL- Left-Channel Amplifier Output Negative Phase 9, 22 B, B5 GND Analog Ground C5 _OUT Clock Signal Output 2 D5 OUTR- Right-Channel Amplifier Output Negative Phase 5 D3 OUTR+ Right-Channel Amplifier Output Positive Phase 7 C3 GAIN Gain-Select Input 8 D2 GAIN2 Gain-Select Input 2 9 D INR- Right-Channel Inverting Input 2 C2 INR+ Right-Channel Noninverting Input 2 C Analog Power Supply. Connect to P. Bypass with a µf capacitor to GND. 23 B2 INL+ Left-Channel Noninverting Input 24 A INL- Left-Channel Inverting Input EP EP Exposed Pad. Connect the exposed thermal pad to the GND plane (see the Supply Bypassing, Layout, and Grounding section). 8

9 µf.µf Functional Diagram P OSCILLATOR AND SAWTOOTH _OUT 47nF 47nF INL+ INL- R IN R IN V BIAS CLASS D MODULATOR AND H-BRIDGE OUTL+ OUTL- 47nF 47nF INR+ INR- R IN R IN CLASS D MODULATOR AND H-BRIDGE OUTR+ OUTR- V BIAS V BIAS GAIN GAIN2 SHDN GAIN CONTROL BIAS GENERATOR GND PGND 9

10 V IN- t SW V IN+ OUT- OUT+ V OUT+ - V OUTt ON(MIN) Figure. Outputs with an Input Signal Applied Detailed Description The filterless, stereo Class D audio power amplifier features several improvements to switch-mode amplifier technology. The offers Class AB performance with Class D efficiency, while occupying minimal board space. A unique, filterless modulation scheme, synchronizable switching frequency, and spread-spectrum switching mode create a compact, flexible, low-noise, efficient audio power amplifier. The differential input architecture reduces common-mode noise pickup, and can be used without input-coupling capacitors. The inputs can also be configured to accept a single-ended input signal. Comparators monitor the inputs and compare the complementary input voltages to the sawtooth waveform. The comparators trip when the input magnitude of the sawtooth exceeds their corresponding input voltage. Both comparators reset at a fixed time after the rising edge of the second comparator trip point, generating a minimum-width pulse (t ON(MIN) ) at the output of the second comparator (Figure ). As the input voltage increases or decreases, the duration of the pulse at one output increases while the other output pulse duration remains the same. This causes the net voltage across the speaker (V OUT+ - V OUT- ) to change. The minimum-width pulse helps the device to achieve high levels of linearity.

11 V IN_+ t SW t SW t SW t SW V IN_- OUT_- OUT_+ t ON(MIN) V OUT_+ - V OUT_- Figure 2. Outputs with an Input Signal Applied (SSM Mode) Operating Modes Fixed-Frequency (FFM) Mode The features two fixed-frequency modes. Connect to GND to select a.mhz switching frequency. Leave unconnected to select a.4mhz switching frequency. The frequency spectrum of the consists of the fundamental switching frequency and its associated harmonics (see the Wideband Output Spectrum (FFM Mode) graph in the Typical Operating Characteristics). Program the switching frequency so the harmonics do not fall within a sensitive frequency band (Table ). Audio reproduction is not affected by changing the switching frequency. Table. Operating Modes MODE GND FFM with f OSC = khz Unconnected FFM with f OSC = 4kHz SSM with f OSC = 2kHz ±6kHz Clocked FFM with f OSC = external clock frequency

12 AMPLITUDE (dbµv/m) VIN_ = V OUT_ FREQUENCY (MHz) OUT_+ Figure 3. with 76mm of Speaker Cable with TDK Common-Mode Choke: TDK ACM X Spread-Spectrum (SSM) Mode The features a unique, patented spreadspectrum mode that flattens the wideband spectral components, improving EMI emissions that may be radiated by the speaker and cables. This mode is enabled by setting = (Table ). In SSM mode, the switching frequency varies randomly by ±6kHz around the center frequency (.2MHz). The modulation scheme remains the same, but the period of the sawtooth waveform changes from cycle to cycle (Figure 2). Instead of a large amount of spectral energy present at multiples of the switching frequency, the energy is now spread over a bandwidth that increases with frequency. Above a few megahertz, the wideband spectrum looks like white noise for EMI purposes (Figure 3). A proprietary amplifier topology ensures this does not corrupt the noise floor in the audio bandwidth. Synchronous Switching Mode The input allows the to be synchronized to a user-defined clock, or another Maxim Class D amplifier, creating a fully synchronous system, minimizing clock intermodulation, and allocating spectral components of the switching harmonics to insensitive frequency bands. Applying a TTL clock signal between khz and 6kHz to synchronizes the. The period of the clock can be randomized, allowing the to be synchronized to another Maxim Class D amplifier operating in SSM mode. VOUT_+ - VOUT_- = V Figure 4. Outputs with No Input Signal _OUT _OUT allows several s as well as other Class D amplifiers (such as the MAX97) to be cascaded. The synchronized output minimizes interference due to clock intermodulation caused by the switching spread between single devices. Using _OUT, the modulation scheme remains the same and audio reproduction is not affected by changing the switching frequency. Filterless Modulation/Common-Mode Idle The uses Maxim s unique, patented modulation scheme that eliminates the LC filter required by traditional class D amplifiers, improving efficiency, reducing component count, conserving board space and system cost. Conventional Class D amplifiers output a 5% duty cycle, 8 out-of-phase square wave when no signal is present. With no filter, the square wave appears across the load as a DC voltage, resulting in finite load current, which increases power consumption especially when idling. When no signal is present at the input of the, the amplifiers will output an in-phase square wave as shown in Figure 4. Because the drives the speaker differentially, the two outputs cancel each other, resulting in no net idle mode voltage across the speaker, minimizing power consumption. Efficiency Efficiency of a Class D amplifier is due to the switching operation of the output stage transistors. In a Class D amplifier, the output transistors act as current-steering switches and consume negligible additional power. Any power loss associated with the Class D output stage is mostly due to the I 2 R loss of the MOSFET onresistance, and quiescent-current overhead. 2

13 EFFICIENCY (%) EFFICIENCY vs. OUTPUT POWER CLASS AB = 3.3V f = khz R L - 8Ω Figure 5. Efficiency vs. Class AB Efficiency The theoretical best efficiency of a linear amplifier is 78%, however that efficiency is only exhibited at peak output powers. Under normal operating levels (typical music reproduction levels), efficiency falls below 3%, whereas the still exhibits >8% efficiencies under the same conditions (Figure 5). Shutdown The has a shutdown mode that reduces power consumption and extends battery life. Driving SHDN low places the in a low-power (.µa) shutdown mode. Connect SHDN to for normal operation. Click-and-Pop Suppression The features comprehensive click-and-pop suppression that eliminates audible transients on startup and shutdown. While in shutdown, the H-bridge is in a high-impedance state. During startup, or power-up, the input amplifiers are muted and an internal loop sets the modulator bias voltages to the correct levels, preventing clicks and pops when the H-bridge is subsequently enabled. For 4ms following startup, a soft-start function gradually unmutes the input amplifiers. Applications Information Filterless Operation Traditional Class D amplifiers require an output filter to recover the audio signal from the amplifier s PWM output. The filters add cost, increase the solution size of the amplifier, and can decrease efficiency. The traditional PWM scheme uses large differential output swings (2 x (P-P) ) and causes large ripple currents. Any parasitic resistance in the filter components results in a loss of power, lowering the efficiency. The does not require an output filter. The device relies on the inherent inductance of the speaker coil and the natural filtering of both the speaker and the human ear to recover the audio component of the square-wave output. Eliminating the output filter results in a smaller, less costly, more efficient solution. Because the frequency of the output is well beyond the bandwidth of most speakers, voice coil movement due to the square-wave frequency is very small. Although this movement is small, a speaker not designed to handle the additional power can be damaged. For optimum results, use a speaker with a series inductance >µh. Typical 8Ω speakers, for portable audio applications, exhibit series inductances in the range of 2µH to µh. Output Offset Unlike a Class AB amplifier, the output offset voltage of a Class D amplifier does not noticeably increase quiescent current draw when a load is applied. This is due to the power conversion of the Class D amplifier. For example, an 8mV DC offset across an 8Ω load results in ma extra current consumption in a Class AB device. In the Class D case, an 8mV offset into 8Ω equates to an additional power drain of 8µW. Due to the high efficiency of the Class D amplifier, this represents an additional quiescent current draw of: 8µW/( / x η), which is on the order of a few µa. Selectable Gain The features four selectable gain settings, minimizing external component count. Gains of db, 3dB, 2dB, and 8dB are set through gain-select inputs, GAIN and GAIN2. GAIN and GAIN2 can be hard-wired or digitally controlled. Table 2 shows the suggested gain settings to attain a maximum output power from a given peak input voltage and given load at = 3.3V and THD+N = %. Table 2. Gain Settings GAIN GAIN2 GAIN (db) INPUT (V RMS ) R L (Ω) P OUT (mw)

14 SINGLE-ENDED LEFT AUDIO INPUT SINGLE-ENDED RIGHT AUDIO INPUT.47µF.47µF.47µF.47µF OUTL+ OUTL- OUTR+ CODEC INL+ INL- INR+ INR- OUTL+ INL+ INR+ INL- INR- OUTL- OUTR+ 2.5V TO 5.5V GAIN2 SHDN OUTR- GAIN GND 2.5V TO 5.5V GAIN2 SHDN OUTR- GAIN GND µf.µf P PGND µf.µf P PGND FFM MODE WITH f OSC = khz, GAIN = 6dB. Figure 6. Single-Ended Input Input Amplifier Differential Input The features a differential input structure, making it compatible with many CODECs and offers improved noise immunity over a single-ended input amplifier. In devices such as cellular phones, high-frequency signals from the RF transmitter can be picked up by the amplifier s input traces. The signals appear at the amplifier s inputs as common-mode noise. A differential input amplifier amplifies the difference of the two inputs, any signal common to both inputs is canceled. Single-Ended Input The can be configured as a single-ended input amplifier by capacitively coupling either input to GND, and driving the other input (Figure 6). DC-Coupled Inputs The input amplifier can accept DC-coupled inputs that are biased within the amplifier s common-mode range (see the Typical Operating Characteristics). DC coupling eliminates the input-coupling capacitors, reducing component count to potentially two external components (Figure 7). However, the highpass filtering effect of the capacitors is lost, allowing low-frequency signals to feed through to the load. FFM MODE WITH f OSC = khz, GAIN = 6dB CODEC BIASED TO /2 COMMON-MODE VOLTAGE. Figure 7. DC-Coupled Inputs Component Selection Input Filter An input capacitor, C IN, in conjunction with the input impedance (R IN ) forms a highpass filter that removes the DC bias from an incoming signal. The AC-coupling capacitor allows the amplifier to automatically bias the signal to an optimum DC level. Assuming zero-source impedance, the -3dB point of the highpass filter is given by: f 3dB = 2πRINCIN Choose C IN so f -3dB is well below the lowest frequency of interest. Use capacitors whose dielectrics have low-voltage coefficients, such as tantalum or aluminum electrolytic. Capacitors with high-voltage coefficients, such as ceramics, may result in increased distortion at low frequencies. Other considerations when designing the input filter include the constraints of the overall system and the actual frequency band of interest. Although high-fidelity audio calls for a flat-gain response between 2Hz and 2kHz, portable voice-reproduction devices such as cellular phones and two-way radios need only concentrate on the frequency range of the spoken human voice (typically 3Hz to 3.5kHz). In addition, speakers used 4

15 C IN 22pF C IN 22pF C IN 22pF C IN 22pF 5V INL+ INR+ INL- INR- OUTL+ OUTL- OUTR+ µf µf 8Ω 8Ω OUTR- _OUT R3 kω 5V R 2kΩ R2 2kΩ C2.µF R4 39kΩ C2 nf µf IN+ MAX97 OUT+.25V MAX4238 µf OUT- IN- 4Ω NOTE: VALUES SHOWN ARE FOR A LOWPASS CUTOFF OF 2Hz AND A BASS GAIN OF -V/V. FFM MODE WITH f OSC = khz. Figure Channel Application Circuit in portable devices typically have a poor response below 3Hz. Taking these two factors into consideration, the input filter may not need to be designed for a 2Hz to 2kHz response, saving both board space and cost due to the use of smaller capacitors. Output Filter The does not require an output filter. The device passes FCC emissions standards with 76mm of unshielded speaker cables. However, output filtering can be used if a design is failing radiated emissions due to board layout or cable length, or if the circuit is near EMI-sensitive devices. Use a ferrite bead filter when radiated frequencies above MHz are of concern. Use an LC filter or a common-mode choke when radiated emissions below MHz are of concern, or when long leads (>76mm) connect the amplifier to the speaker. 2. Channel Configuration The typical 2. channel application circuit (Figure 8) shows the configured as a mid-/high-frequency amplifier and the MAX97 configured as a mono bass amplifier. Input capacitors (C IN ) set the highpass cutoff frequency according to the following equation: f = 2π RIN CIN where R IN is the typical input resistance of the. The µf capacitors on the output of the ensure a two-pole highpass filter. 5

16 Low frequencies are summed through a two-pole lowpass filter and sent to the MAX97 mono speaker amplifier. The passband gain of the lowpass filter is unity for in-phase stereo signals, where R = R2 and R3 = R//R2. The cutoff frequency of the lowpass filter is set by the following equation: f = 2π 2 R3 R C C2 R3 R4 Supply Bypassing, Layout, and Grounding Proper layout and grounding are essential for optimum performance. Use large traces for the power-supply inputs and amplifier outputs to minimize losses due to parasitic trace resistance. Large traces also aid in moving heat away from the package. Proper grounding improves audio performance, minimizes crosstalk between channels, and prevents any switching noise from coupling into the audio signal. Connect PGND and GND together at a single point on the PC board. Route all traces that carry switching transients away from GND and the traces/components in the audio signal path. Bypass with µf to GND and P with.µf to PGND. Place the bypass capacitors as close to the as possible. Use large, low-resistance output traces. Current drawn from the outputs increases as load impedance decreases. High-output trace resistance decreases the power delivered to the load. Large output, supply, and GND traces allow more heat to move from the to the air, decreasing the thermal impedance of the circuit. The thin QFN-EP package features an exposed thermal pad on its underside. This pad lowers the package s thermal impedance by providing a direct heat conduction path from the die to the printed circuit board. Connect the exposed thermal pad to the GND plane. UCSP Applications Information For the latest application details on UCSP construction, dimensions, tape carrier information, printed circuit board techniques, bump-pad layout, and recommended reflow temperature profile as well as the latest information on reliability testing results, refer to Application Note: UCSP A Wafer-Level Chip-Scale Package available on Maxim s website at 6

17 System Diagram µf.µf.µf.µf P 2.2kΩ 2.2kΩ.µF AUX_IN V CC MAX46 BIAS OUT IN+ CODEC 47nF 47nF 47nF 47nF OUTR+ OUTR- OUTL- OUTL+ INR+ INR- INL- INL+.µF IN- GND GAIN GAIN2 SHDN GND PGND µf SHDN µf INL OUTL µcontroller µf MAX9722B INR CP CIN OUTR PV SS SV SS µf µf 7

18 TOP VIEW INL- SHDN OUTL+ P OUTL- INR- 9 GAIN2 GAIN N.C. OUTR+ PVDD PGND 3 2 OUTR- TOP VIEW (BUMPS ON BOTTOM) A Pin Configurations INR+ 2 N.C. 2 _OUT B GND INL+ PGND GND GND INL GND N.C. C INR+ GAIN PGND _OUT INL OUTL D INR- GAIN2 OUTR+ P OUTR- SHDN N.C. OUTL+ PVDD PGND UCSP TQFN Chip Information TRANSISTOR COUNT: 5688 PROCESS: BiCMOS 8

19 Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to 24L QFN THIN.EPS PACKAGE OUTLINE, 2, 6, 2, 24, 28L THIN QFN, 4x4x.8mm 2-39 F 2 PACKAGE OUTLINE, 2, 6, 2, 24, 28L THIN QFN, 4x4x.8mm F 2 Package Code: T

20 Package Information (continued) (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to 5x4 UCSP.EPS Package Code: B2- Pages changed at Rev 2:, 5, 2 Revision History Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 2 Maxim Integrated Products, 2 San Gabriel Drive, Sunnyvale, CA Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.