Huang-Jen Chiu. National Taiwan University of Science and Technology. Dept. of Electronic Engineering

Transcription

1 Huang-Jen Chiu Dept. of Electronic Engineering National Taiwan University of Science and Technology Office: EE502-1 Tel:

2 Textbook Power Electronics --Converters, Applications, and Design Third Edition Mohan / Undeland / Robbins 民全書局 Midterm: 50% Final: 50%

3 Outlines Power Electronic Systems Overview of Power Semiconductor Switches Switch-Mode DC/DC Converters Switch-Mode DC/AC Inverters Resonant Converters Switching DC Power Supplies Power Conditioners and Uninterruptible Power Supplies Practical Converter Design Considerations

4 Chapter 1 Power Electronic Systems

5 Power Electronic Systems

6 Linear Power Supply Series transistor as an adjustable resistor Low Efficiency Heavy and bulky

7 Switch-Mode Power Supply Transistor as a switch High Efficiency High-Frequency Transformer

8 Basic Principle of Switch-Mode Synthesis Constant switching frequency Pulse width controls the average L-C filters the ripple

9 Application in Adjustable Speed Drives Conventional drive wastes energy across the throttling valve to adjust flow rate Using power electronics, motor-pump speed is adjusted efficiently to deliver the required flow rate

10 Scope and Applications

11 Scope and Applications

12 Classification of Power Converters ac-dc converters (controlled rectifiers) dc-dc converters (dc choppers) dc-ac converters (inverters) ac-ac converters (ac voltage controllers)

13 Power Processor as a Combination of Converters Most practical topologies require an energy storage element, which also decouples the input and the output side converters

14 Power Flow through Converters Converter is a general term An ac/dc converter is shown here Rectifier Mode of operation when power from ac to dc Inverter Mode of operation when power from ac to dc

15 AC Motor Drive Converter 1 rectifies line-frequency ac into dc Capacitor acts as a filter; stores energy; decouples Converter 2 synthesizes low-frequency ac to motor Polarity of dc-bus voltage remains unchanged ideally suited for transistors of converter 2

16 Matrix Converter Very general structure Would benefit from bi-directional and bi-polarity switches Being considered for use in specific applications

17 Interdisciplinary Nature of Power Electronics

18 Chapter 2 Overview of Power Semiconductor Devices

19 Diodes On and off states controlled by the power circuit

20 Diode Turn-Off Fast-recovery diodes have a small reverse-recovery time

21 Thyristors Semi-controlled device Latches ON by a gate-current pulse if forward biased Turns-off if current tries to reverse

22 Thyristor in a Simple Circuit For successful turn-off, reverse voltage required for an interval greater than the turn-off interval

23 Generic Switch Symbol Idealized switch symbol When on, current can flow only in the direction of the arrow Instantaneous switching from one state to the other Zero voltage drop in on-state Infinite voltage and current handling capabilities

24 Switching Characteristics (linearized) Switching Power Loss is proportional to: switching frequency 1 P = V I turn-on and turn-off times d o s fs(tc(on) + 2 t c(off) )

25 Bipolar Junction Transistors (BJT) Used commonly in the past Now used in specific applications Replaced by MOSFETs and IGBTs

26 Various Configurations of BJTs

27 MOSFETs Easy to control by the gate Optimal for low-voltage operation at high switching frequencies On-state resistance a concern at higher voltage ratings

28 Gate-Turn Turn-Off Thyristors (GTO) Slow switching speeds Used at very high power levels Require elaborate gate control circuitry

29 GTO Turn-Off Need a turn-off snubber

30 Insulated Gate Bipolar Transistor (IGBT)

31 MOS-Controlled Thyristor (MCT) Simpler Drive and faster switching speed than those of GTOs. Current ratings are significantly less than those of GTOs.

32 Comparison of Controllable Switches

33 Summary of Device Capabilities

34 Rating of Power Devices

35 Chapter 3 Review of Basic Electrical and Magnetic Circuit Concepts

36 Sinusoidal Steady State P PF = = cosφ S

37 Three-Phase Circuit

38 Steady State in Power Electronics

39 f(t) = Fourier Analysis 1 F0 + fh(t) = a0 + h + h t) 2 h= 1 h= 1 { a cos(hωt) b sin(hω }

40 Distortion in the Input Current PF P I = = s1 cosφ1 = S I s DPF = Voltage is assumed to be sinusoidal I I s1 s THD 2 i DPF Subscript 1 refers to the fundamental The angle is between the voltage and the current fundamental

41 Phasor Representation

42 Response of L and C v L = L di dt L i c = C dv dt c

43 Inductor Voltage and Current in Steady State Volt-seconds over T equal zero.

44 Capacitor Voltage and Current in Steady State Amp-seconds over T equal zero.

45 Ampere s s Law H dl = i Direction of magnetic field due to currents Ampere s Law: Magnetic field along a path

46 Direction of Magnetic Field B = μh

47 B-H H Relationship; Saturation Definition of permeability

48 Continuity of Flux Lines φ + φ + φ =

49 Concept of Magnetic Reluctance Flux is related to ampere-turns by reluctance

50 Analogy between Electrical and Magnetic Variables

51 Analogy between Equations in Electrical and Magnetic Circuits

52 Faraday s s Law and Lenz s s Law dφ e = N = dt L di dt

53 Inductance L Inductance relates flux-linkage to current

54 Analysis of a Transformer

55 Transformer Equivalent Circuit

56 Including the Core Losses L ' = ( l2 N N N 1 2 ) 2 L 1 2 2' ( ) R2 N2 R = l2

57 Chapter 4 Computer Simulation

58 System to be Simulated Challenges in modeling power electronic systems

59 Large-Signal System Simulation Simplest component models

60 Small-Signal Signal Linearized Model for Controller Design System linearized around the steady-state point

61 Closed-Loop Operation: Large Disturbances Simplest component models Nonlinearities, Limits, etc. are included

62 Modeling of Switching Operation Detailed device models Just a few switching cycles are studied

63 Modeling of a Simple Converter Modeling of a Simple Converter 0 R v - dt dv C - i v v dt di L i r c c L oi c L L L = = + + oi c L L c L v 0 L 1 v i CR 1 - C 1 L 1 - L r - dt dv dt di + =

64 Modeling using PSpice Schematic approach is far superior

65 PSpice-based Simulation Simulation results

66 Simulation using MATLAB

67 Chapter 5 Diode Rectifiers

68 Diode Rectifier Block Diagram Uncontrolled utility interface (ac to dc)

69 A Simple Circuit Resistive load

70 A Simple Circuit (R-L L Load) Current continues to flows for a while even after the input voltage has gone negative

71 A Simple Circuit (Load has a dc back-emf) Current begins to flow when the input voltage exceeds the dc back-emf Current continues to flows for a while even after the input voltage has gone below the dc back-emf

72 Single-Phase Diode Rectifier Bridge Large capacitor at the dc output for filtering and energy storage

73 Diode-Rectifier Bridge Analysis

74 Diode-Rectifier Bridge Input Current

75 Current Commutation Assuming inductance in this circuit to be zero

76 Current Commutation

77 Current Commutation in Full-Bridge Rectifier

78 Current Commutation

79 Rectifier with a dc-side voltage

80 Diode-Rectifier with a Capacitor Filter Power electronics load is represented by an equivalent load resistance

81 Diode Rectifier Bridge Equivalent circuit for analysis on one-half cycle basis

82 Diode-Bridge Rectifier: Waveforms Analysis using PSpice

83 Input Line-Current Distortion Analysis using PSpice

84 Line-Voltage Distortion PCC is the point of common coupling

85 Line-Voltage Distortion Distortion in voltage supplied to other loads

86 Voltage Doubler Rectifier In 115-V position, one capacitor at-a-time is charged from the input.

87 A Three-Phase, Four-Wire System A common neutral wire is assumed

88 Three-Phase, Full-Bridge Rectifier Commonly used

89 Three-Phase, Full-Bridge Rectifier Output current is assumed to be dc

90 Three-Phase, Full-Bridge Rectifier: Input Line-Current Assuming output current to be purely dc and zero ac-side inductance

91 Rectifier with a Large Filter Capacitor Output voltage is assumed to be purely dc

92 Chapter 6 Thyristor Converters Controlled conversion of ac into dc

93 Chapter 6 Thyristor Converters Controlled conversion of ac into dc

94 Thyristor Converters Two-quadrant conversion

95 Primitive circuits with thyristors

96 Thyristor Triggering

97 Full-Bridge Thyristor Converters Single-phase and three-phase

98 Single-Phase Thyristor Converters

99 Average DC Output Voltage is ( ωt) = 2Is1sin( ωt - ) + 2Is3Is1sin[3( ωt - )] I = = P = 0.9cos π s1 2Id 0.9I d Assuming zero ac-side inductance

100 Input Line-Current Waveforms Harmonics, power and reactive power

101 1-Phase Thyristor Converter

102 Thyristor Converter

103 DC Voltage versus Load Current Various values of delay angle

104 Thyristor Converters: Inverter Mode Assuming the ac-side inductance to be zero

105 Thyristor Converters: Inverter Mode Family of curves at various values of delay angle

106 Thyristor Converters: Inverter Mode

107 Thyristor Converters: Inverter Mode

108 3-Phase Thyristor Converters

109

110

111 Chapter 7 DC-DC Switch-Mode Converters dc-dc converters for switch-mode dc power supplies and dc-motor drives

112 Block Diagram of DC-DC Converters Functional block diagram

113 Stepping Down a DC Voltage A simple approach that shows the evolution

114 Pulse-Width Modulation in DC-DC Converters

115 Step-Down DC-DC Converter ( V V ) T = V d o on o T off V V o d T = on = D T <1

116 Waveforms at the boundary of Cont./ Discont.. Conduction D(1-1 t T V I I on (V -V ) s d LB = L,peak = d o = D(1- D) = 2 2L 2L 4I LB, max D) Critical current below which inductor current becomes discontinuous

117 Step-Down DC-DC Converter: Discontinuous Conduction Mode V V o d = D D ( I 2 I o LB, max ) Steady state; inductor current discontinuous

118 Limits of Cont./ Discont. Conduction V V o = d D : CCM V V o d = D D ( I 2 I o LB, max ) : DCM

119 Output Voltage Ripple ΔQ Δ Vo = = C ΔI LT 8C s

120 Step-Up DC-DC Converter V d T on V o 1 = (Vo Vd ) Toff = > 1 V 1 D Output voltage must be greater than the input d

121 Limits of Cont./ Discont. Conduction 1 t T V I I on V s o LB = L,peak = d = D(1- D) = 4I 2 2L 2L TsVo 2 27 I ob = (1- D)ILB = D(1- D) = D(1- D) 2L 4 LB, max 2 D(1- D) I ob,max

122 Discont.. Conduction D = 4 27 V V o d V ( V o d -1) I I o ob, max

123 Limits of Cont./ Discont. Conduction V V o d 1 = 1 D : CCM D = 4 27 V V o d V ( V o d -1) I I o ob, max : DCM

124 Output Ripple ΔV I t o on o = = C V o R DT C s

125 Step-Down/Up DC-DC Converter V T = d on V o T off V V o d = 1 D D The output voltage can be higher or lower than the input voltage

126 Limits of Cont./ Discont. Conduction 2 ob ob,max(1- D) 1 t T V I = I on V s o L,peak = d = (1- D) 2 2L 2L TsVo 2 I = (1- D)ILB = (1- D) = I 2L LB = I LB, max (1- D)

127 Discontinuous Conduction Mode V D = V o d I I o ob, max This occurs at light loads

128 Limits of Cont./ Discont. Conduction V o V d = D V I : CCM D = o o : DCM 1 D V I d ob, max

129 Output Voltage Ripple ΔV I t o on o = = C V o R DT C s ESR is assumed to be zero

130 The output voltage can be higher or lower than the input voltage Cuk DC-DC Converter

131 Converter for DC-Motor Drives

132 Converter Waveforms

133 Output Ripple in Converters for DC-Motor Drives

134 Switch Utilization in DC-DC Converters It varies significantly in various converters

135 Reversing the Power Flow in DC-DC Converters

136 Chapter 8 Switch-Mode DC-AC Inverters Converters for ac motor drives and uninterruptible power supplies

137 Switch-Mode DC-AC Inverter

138 Switch-Mode DC-AC Inverter

139 V m = a m = f Synthesis of a Sinusoidal Output ^ control ^ Vtri f f s 1 by PWM

140 Details of a Switching Time Period Small m f (m f 21): Synchronous PWM Large m f (m f >21): Asynchronous PWM

141 Harmonics in the DC-AC Inverter Output Voltage Harmonics appear around the carrier frequency and its multiples

142 Harmonics due to Over-modulation These are harmonics of the fundamental frequency

143 Square-Wave Mode of Operation Harmonics are of the fundamental frequency Less switching losses in high power applications The DC input voltage must be adjusted

144 Half-Bridge Inverter Capacitors provide the mid-point

145 Single-Phase Full-Bridge DC-AC Inverter Consists of two inverter legs

146 PWM to Synthesize Sinusoidal Output

147 Analysis assuming Fictitious Filters Small fictitious filters eliminate the switching-frequency related ripple

148 DC-Side Current

149 Uni-polar Voltage Switching

150 DC-Side Current in a Single-Phase Inverter

151 Sinusoidal Synthesis by Voltage Shift Phase shift allows voltage cancellation to synthesize a 1-Phase sinusoidal output

152 Square-Wave and PWM Operation PWM results in much smaller ripple current

153 Push-Pull Pull Inverter Only one switch conducts at any instant of time High efficiency for low-voltage source applications

154 Three-Phase Inverter Three inverter legs; capacitor mid-point is fictitious

155 Three-Phase PWM Waveforms

156 Three-Phase Inverter Harmonics

157 Three-Phase Inverter Output

158 Square-Wave and PWM Operation PWM results in much smaller ripple current

159 DC-Side Current in a Three-Phase Inverter The current consists of a dc component and the switching-frequency related harmonics

160 Effect of Blanking Time Results in nonlinearity

161 Effect of Blanking Time ΔV o 2t T = s 2t - T Δ Δ s V d V d,i o,i o > 0 < 0 Voltage jump when the current reverses direction

162 Effect of Blanking Time Effect on the output voltage

163 Programmed Harmonic Elimination Angles based on the desired output

164 Tolerance-Band Current Control Results in a variable frequency operation

165 Fixed-Frequency Frequency Operation Better control is possible using dq analysis

166 Chapter 9 Zero-Voltage or Zero-Current Switchings converters for soft switching

167 Hard Switching Waveforms The output current can be positive or negative

168 Turn-on and Turn-off Snubbers

169 Switching Trajectories Comparison of Hard versus soft switching

170 Undamped Undamped Series Series-Resonant Circuit Resonant Circuit L c r d c L r i dt dv C V v dt di L = = + ) t t ( sin I Z ) t - (t )cos - V (V - V (t) v ) t t ( sin Z - V V ) t - (t cos I (t) i o o Lo o o o co d d c o o o co d o o Lo L + = + = ω ω ω ω V d

171 Series Series-Resonant Circuit Resonant Circuit with Capacitor with Capacitor-Parallel Load Parallel Load o L c r c d c L r I - i dt dv C i V v dt di L = = = + ) t t ( sin ) I - (I Z ) t - (t )cos - V (V - V (t) v ) t t ( sin Z - V V ) t - (t )cos I - (I I (t) i o o o Lo o o o co d d c o o o co d o o o Lo o L + = + + = ω ω ω ω

172 Impedance of a Series-Resonant Circuit Q = ω L R 1 C o r = = ω o r R Z R o The impedance is capacitive below the resonance frequency

173 Undamped Undamped Parallel Parallel-Resonant Circuit Resonant Circuit dt di L v I dt dv C i L r c d c r L = = + ) t t ( cos V ) t - (t )sin I - (I Z (t) v ) t t ( sin Z V ) t - (t )cos I - (I I (t) i o o co o o Lo d o c o o o co o o d Lo d L + = + + = ω ω ω ω

174 Impedance of a Parallel-Resonant Circuit R Q = ω o RC r = = ω L o r R Z o The impedance is inductive at below the resonant frequency

175 Series-Loaded Resonant (SLR) Converter 2ω s <ω o Turn off with ZVS and ZCS Turn on with ZCS Thyristors used Large peak current, high conduction losses ZCS ZVS, ZCS

176 SLR Converter Waveforms 1/2ω o <ω s <ω o ZVS, ZCS Turn off with ZVS and Thyristors used Large turn - on switching ZCS losses

177 SLR Converter Waveforms ω s >ω o Turn on with ZVS and ZCS Large turn - off switching losses Controllable switches used ZVS, ZCS

178 Lossless Snubbers in SLR Converters The operating frequency is above the resonance frequency

179 SLR Converter Characteristics The operating frequency is varied to regulate the output voltage

180 SLR Converter Control The operating frequency is varied to regulate the output voltage

181 Parallel-Loaded Resonant (PLR) Converter No turn - on and turn - off losses ZVS, ZCS ZCS ω s 1 ω 2 o

182 PLR Converter Waveforms No turn - off losses ZVS, ZCS 1 ω o < ω s < 2 ω o

183 PLR Converter Waveforms No turn - on losses ZVS

184 PLR Converter Characteristics Output voltage as a function of operating frequency for various values of the output current

185 Hybrid-Resonant DC-DC Converter Combination of series- and parallel-loaded resonances A SLR offers an inherent current limiting under short-circuit conditions and a PLR regulating its voltage at no load with a high-q resonant tank is not a problem

186 Parallel-Resonant Current-Source Converter Resistive Induction Coil Capacitive Basic circuit to illustrate the operating principle at the fundamental frequency

187 Parallel-Resonant Current-Source Converter Using thyristors; for induction heating

188 Class-E E Converters Single-switch Used for high - frequency electronic ballasts Sin-wave Current ZCS Turn-on No switching losses ZVS Turn-off High peak volatge and current

189 Class-E E Converters

190 Resonant Switch Converters

191 ZCS Resonant-Switch Converter Voltage is regulated by varying the switching frequency ZCS Turn-on ZCS Turn-off

192 ZCS Resonant-Switch Converter Accelerating diode ZCS Turn-off ZCS Turn-on Discharge slowly at light load

193 ZVS Resonant-Switch Converter ZVS Turn-off ZVS Turn-on

194 MOSFET Internal Capacitances ZVS is preferable over ZCS at high switching frequencies These capacitances affect the MOSFET switching

195 ZVS-CV DC-DC Converter ZVS Turn-on The inductor current must reverse direction during each switching cycle

196 ZVS-CV DC-DC Converter

197 ZVS-CV Principle Applied to DC-AC Inverters

198 Three-Phase ZVS-CV DC-AC Inverter Very large ripple in the output current

199 Output Regulation by Voltage Control Each pole operates at nearly 50% duty-ratio

200 ZVS-CV with Voltage Cancellation Commonly used

201 Resonant DC-Link Inverter ZVS Turn-on The dc-link voltage is made to oscillate

202 Three-Phase Resonant DC-Link Inverter Modifications have been proposed

203 High-Frequency Frequency-Link Inverter Basic principle for selecting integral half-cycles of the high-frequency ac input

204 High-Frequency Frequency-Link Inverter Low-frequency ac output is synthesized by selecting integral half-cycles of the high-frequency ac input

205 High-Frequency Frequency-Link Inverter Shows how to implement such an inverter

206 Chapter 10 Switching DC Power Supplies One of the most important applications of power electronics

207 Linear Power Supplies Very poor efficiency and large weight and size

208 Switching DC Power Supply High efficiency and small weight and size

209 Switching DC Power Supply: Multiple Outputs In most applications, several dc voltages are required, possibly electrically isolated from each other

210 Transformer Analysis Needed to discuss high-frequency isolated supplies

211 PWM to Regulate Output

212 Flyback Converter Derived from buck-boost; very power at small power (> 50 W ) power levels

213 Flyback Converter Switch on and off states (assuming incomplete core demagnetization)

214 Flyback Converter Switching waveforms (assuming incomplete core demagnetization)

215 Other Flyback Converter Topologies

216 Forward Converter Derived from Buck; idealized to assume that the transformer is ideal (not possible in practice)

217 Forward Converter: in Practice Switching waveforms (assuming incomplete core demagnetization)

218 Forward Converter: Other Possible Topologies Two-switch Forward converter is very commonly used

219 Push-Pull Pull Inverter Leakage inductances become a problem

220 Half-Bridge Converter Derived from Buck

221 Full-Bridge Converter Used at higher power levels (> 0.5 kw )

222 Current-Source Converter More rugged (no shoot-through) but both switches must not be open simultaneously

223 Ferrite Core Material Several materials to choose from based on applications

224 Core Utilization in Various Converter Topologies At high switching frequencies, core losses limit excursion of flux density

225 Control to Regulate Voltage Output Linearized representation of the feedback control system

226 + = + = s d s d T d B v A x x dt Bv Ax x ),(1, = = s o s o T d C x v dt C x v ),(1, = = x d C C d v v d B Bd x d A Ad x o d )] (1 [ )] (1 [ )] (1 [ V d d D B d D B x X d D A d D A x X )] ( [1 ) ( [ ) )]}( ( [1 ) ( { ~ 2 ~ 1 ~ ~ 2 ~ 1 ~ = + d V d B D B B d B D x X d A D A A d AD ] ) (1 [ ) ]( ) (1 [ ~ 2 2 ~ 1 1 ~ ~ 2 2 ~ = ~ ~ 2 1 ~ 2 1 ~ ) ( )] (1 [ ] ) ( ) [( )] (1 [ )] (1 [ d x A A x D A AD d V B B X A A V D B B D X D A D A d d = Linearization of the Power Stage Linearization of the Power Stage

227 Linearization of the Power Stage Linearization of the Power Stage ~ ~ ~ ] ) ( ) [( d V B B X A A Ax BV AX x X d d ~ ~ ~ ] ) ( ) [( d V B B X A A Ax x d + + = BV d AX X + = = 0 Θ ~ ~ 2 1 ~ 2 1 ~ ~ ~ 2 ~ 1 ~ ) ( )] (1 [ ] ) [( )] (1 [ ] )][ ( [1 ) ( { xd C C x D C C D d X C C X D C C D x X d D C d D C v V o o = = + ~ ~ 2 1 ~ ] ) [( x C d X C C CX v V o o ΘV o =CX ~ 2 1 ~ ~ ] ) [( d X C C Cx v o + =

228 BV d AX X + = = 0 Linearization of the Power Stage Linearization of the Power Stage CX V and o = B CA V V d o 1 = Steady-state DC voltage transfer ratio ~ ~ ~ ] ) ( ) [( d V B B X A A Ax x d + + = ) ( ] ) ( ) [( ) ( ) ( ~ ~ ~ s d V B B X A A s Ax s x s d + + = ) ( ] ) ( ) [( ] [ ) ( ~ ~ s d V B B X A A A si s x d + = X C C V B B X A A A C si s d s v s T d o p ) ( ] ) ( ) [( ] [ ) ( ) ( ) ( ~ ~ + + = = ~ 2 1 ~ ~ ] ) [( d X C C Cx v o + =

229 Forward Converter: An Example Forward Converter: An Example = + + = ) ( 0 ) ( C x x R x Cr x C x x R x r Lx V c L d d c c c c L c L c V L x x r C R r C R R r R L R r R L r r Rr Rr x x = 0 1 ) ( 1 ) ( ) ( ) ( A 1 =A 2 B 1 B 2 =0

230 + + = = ) ( x x r R R r R Rr C x x R v c c c o C 1 =C ,, C C B D B A A = = = + = = CR C L L r r A A A L c [ ] c r C C C = = + >> ) ( L r C r R D L B D B = = 0 1/ 1

231 + + + = L r r C L CR R r r LC A L c L c )/ ( 1 1 D r r R r R D V V L c c d o = ) ( { } ~ ~ 2 1/ ] )/ ( [1/ 1 ) ( ] ) ( ) [( ] [ ) ( ) ( ) ( o o z z o d L c c d d o p s s s V LC L r r CR s LC s src V X C C V B B X A A A C si s d s v s T ω ξω ω ω ω = = =

232 Forward Converter: Transfer Function Plots T p ( s) = V d 2 ωo ω z s 2 s+ ωz + 2ξω s+ ω o 2 o

233 Flyback Converter: Transfer Function Plots T p ( s) (1+ s/ ωz )(1 s/ ωz 2 = Vd f ( D) 2 as + b s+ c 1 ) o

234 Linearizing Linearizing the PWM Block the PWM Block ^ ~ ~ 1 ) ( ) ( ) ( r c m V s v s d s T = = ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ~ ~ ~ ~ ~ ~ s T s T s v s d s d s v s v s v s T m p c o c o l = = =

235 Typical Gain and Phase Plots of the Open-Loop Transfer Function Definitions of the crossover frequency, phase and gain margins

236 A General Amplifier for Error Compensation Can be implemented using a single op-amp

237 Type-2 2 Error Amplifier Shows phase boost at the crossover frequency

238 Feedback-Loop Stabilization

239 Feedback-Loop Stabilization F F co K = = z F F p co

240 Feedback-Loop Stabilization F F co K = = z F F p co θ total lag = 270 tan K + tan K

241 L C o o = Compensator Design Example 3V I o on T = = 6 V o 5V I o(nom) 10A I o(min) 1A Switching frequency 100kHz Minimum output ripple 50mV P-P di V or = Fo = = π L C F o 1 2πR o Hz 6 6 = 15 μh = 2600μF = = 2. esr esr C 5 6 o = π 1 khz

242 Compensator Design Example G 0.5( V 0.5 (11 1) 3 sp m = = = ) db 2.5 G m + G s = log( ) = = 1. 5dB 5 R = R 100(40dB) = 1k 100 = 100kΩ 2 1

243 Compensator Design Example 1 Fco = Fs = 20 khz 5 F F co esr 20 k = = 8 lag = k EA lag = = 218 K = 4 Fco 20 1 Fz = = = 5kHz C1 = = 318 pf K 4 2π (100 k )(5k ) 1 F p = K Fco = 4 20 = 80 khz C 2 = = 20 2π (100 k )(80 k ) pf

244 Voltage Feed-Forward Forward Makes converter immune from input voltage variations

245 Voltage versus Current Mode Control

246 Various Types of Current Mode Control

247 Peak Current Mode Control Slope compensation is needed

248 A Typical PWM Control IC

249 Current Limiting

250 Implementing Electrical Isolation in the Feedback Loop

251 Implementing Electrical Isolation in the Feedback Loop

252 Input Filter Needed to comply with the EMI and harmonic limits

253 ESR of the Output Capacitor ESR often dictates the peak-peak voltage ripple

254 Chapter 11 Power Conditioners and Uninterruptible Power Supplies Becoming more of a concern as utility de-regulation proceeds

255 Distortion in the Input Voltage The voltage supplied by the utility may not be sinusoidal

256 Typical Voltage Tolerance Envelope for Computer Systems This has been superceded by a more recent standard

257 Typical Range of Input Power Quality

258 Electronic Tap Changers Controls voltage magnitude by connecting the output to the appropriate transformer tap

259 Uninterruptible Power Supplies (UPS) Block diagram; energy storage is shown to be in batteries but other means are being investigated

260 UPS: Possible Rectifier Arrangements The input normally supplies power to the load as well as charges the battery bank

261 UPS: Another Possible Rectifier Arrangement Consists of a high-frequency isolation transformer

262 UPS: Another Possible Input Arrangement A separate small battery charger circuit

263 Battery Charging Waveforms as Function of Time Initially, a discharged battery is charged with a constant current

264 UPS: Various Inverter Arrangements Depends on applications, power ratings

265 UPS: Control Typically the load is highly nonlinear and the voltage output of the UPS must be as close to the desired sinusoidal reference as possible

266 UPS Supplying Several Loads With higher power UPS supplying several loads, malfunction within one load should not disturb the other loads

267 Another Possible UPS Arrangement Functions of battery charging and the inverter are combined

268 UPS: Using the Line Voltage as Backup Needs static transfer switches

269 Chapter 16 Residential and Industrial Applications Significant in energy conservation; productivity

270 Inductive Ballast of Fluorescent Lamps Inductor is needed to limit current

271 Rapid-Start Fluorescent Lamps Starting capacitor is needed

272 Electronic Ballast for Fluorescent Lamps Lamps operated at ~40 khz

273 Induction Cooking Pan is heated directly by circulating currents increases efficiency

274 Industrial Induction Heating Needs sinusoidal current at the desired frequency: two options

275 Welding Application

276 Switch-Mode Welders Can be made much lighter weight

277 Chapter 17 Electric Utility Applications These applications are growing rapidly

278 HVDC Transmission There are many such systems all over the world

279 Control of HVDC Transmission System Inverter is operated at the minimum extinction angle and the rectifier in the current-control mode

280 HVDC Transmission: AC-Side Filters Tuned for the lowest (11 th and the 13 th harmonic) frequencies

281 Effect of Reactive Power on Voltage Magnitude

282 Thyristor-Controlled Inductor (TCI) Increasing the delay angle reduces the reactive power drawn by the TCI

283 Thyristor-Switched Capacitors (TSCs( TSCs) Transient current at switching must be minimized

284 Instantaneous VAR Controller (SATCOM) Can be considered as a reactive current source

285 Characteristics of Solar Cells The maximum power point is at the knee of the characteristics

286 Photovoltaic Interface This scheme uses a thyristor inverter

287 Harnessing of Wing Energy A switch-mode inverter may be needed on the wind generator side also

288 Active Filters for Harmonic Elimination Active filters inject a nullifying current so that the current drawn from the utility is nearly sinusoidal

289 Chapter 18 Utility Interface Power quality has become an important issue

290 Various Loads Supplied by the Utility Source PCC is the point of common coupling

291 Diode-Rectifier Bridge

292 Typical Harmonics in the Input Current Single-phase diode-rectifier bridge

293 Harmonic Guidelines: IEEE 519 Commonly used for specifying limits on the input current distortion

294 Harmonic Guidelines: IEEE 519 Limits on distortion in the input voltage supplied by the utility

295 Reducing the Input Current Distortion use of passive filters

296 Power-Factor Factor-Correction (PFC) Circuit For meeting the harmonic guidelines

297 Power-Factor Factor-Correction (PFC) Circuit Control generating the switch on/off signals

298 Power-Factor Factor-Correction (PFC) Circuit Operation during each half-cycle

299 Switch-Mode Converter Interface Bi-directional power flow; unity PF is possible

300 Switch-Mode Converter Control DC bus voltage is maintained at the reference value

301 Switch-Mode Converter Interface

302 EMI: Conducted Interefence Common and differential modes

303 Switching Waveforms Typical rise and fall times

304 Conducted EMI Various Standards

305 Conducted EMI Filter

306 Turn-off Snubber + i D F D f I o Turn-off snubber D f I o V d - S w D s R s C s i C s V d i sw I o - i C s sw C s = I o t fi 2V d, t on >2.3R s C s, V d /R s <0.2I o

307 Turn-on Snubber + V d D f L s I o R Ls D Ls D f Snubber circuit + V d L s D f R Ls D Ls I o I o i sw With snubber Without snubber L s di sw dt - S w - S w V d v sw Δv sw = L s I o t ri t off >2.3L s /R s Pr=1/2L s I o^2f s

308 Aspects of EMC (EMI EMS) EMC is concerned with the generation, transmission, and reception of electromagnetic energy EMI occurs if the received energy causes the receptor to behave in an undesired manner

309 EMI Sources and Sensors

310 Three Ways to Prevent Interference Suppress the emission at its source Make the coupling path as inefficient as possible Make the receptor less susceptible to the emission

311 Four Basic EMC Problems

312 Other Aspects of EMC

313 EMC Requirements Those required by governmental agencies Those imposed by the product manufacturer

314 Frequency Range of EMC Requirements

315 National Regulations Summary

316 Federal Communications Commission (FCC) Class A for use in a commercial, industrial or business environment Class B for use in a residential environment

317 FCC Emission for Class B

318 FCC Emission for Class A

319 Comparison of the FCC Class A and Class B Radiated Emission Limits

320 Open Area Test Site

321 Chamber for Measurement of Radiated Emissions

322 Radiated EMI Test Setup

323 Antennas

324 Conducted EMI Test Setup

325 Line Impedance Stabilization Network (LISN)

326 Conducted Emissions Test Layout

327 Conducted Emissions Test Layout

328 CISPR Bandwidth Requirements

329 Three Detection Modes Envelope Detector Quasi-Peak Detector Average Detector

330 Design Constraints for Products Product Cost Product Marketability Product Manufacturability Product Development Schedule

331 Advantages of EMC Design Minimizing the additional cost required by suppression elements or redesign Maintaining the development and product announcement schedule Insuring that the product will satisfy the regulatory requirements

332 Effects of Component Leads

333 Resistors

334 1000Ω, Carbon Resistor having 1/4 Inch Lead Lengths

335 Capacitors

336 470 pf Ceramic Capacitor with Short Lead Lengths

337 470 pf Ceramic Capacitor with 1/2 Inch Lead Lengths

338 0.15 μf Tantalum Capacitor with Short Lead Lengths

339 0.15 μf Tantalum Capacitor with 1/2 Inch Lead Lengths

340 Inductors

341 1.2μH Inductor

342 Common-Mode Choke

343 Common-Mode Choke

344 Frequency Response of the Relative Permeabilities of Ferrite

345 Ferrite Beads

346 Multi-Turn Ferrite Beads

347 Driver Circuit of the DC Motor

348 The Periodic, Trapezoidal Pulse Train Representing Clock and Data Signals The key parameters that contribute to the highfrequency spectral content of the waveform are the rise-time and fall-time of the pulse.

349 The Spectra of 1V, 10MHz, 50% Duty Cycle Trapezoidal Pulse Trains for Rise-/Fall-time of 20ns/5ns

350 Spectrum Analyzer

351 The Effect of Bandwidth on Spectrum

352 The Effects of Differential-Mode Current and Common-Mode Currents Common-mode current often produce larger radiated emissions than the differential-mode currents

353 Differential-Mode Current Emission E,max = I D Kf D 2 A

354 Radiated Emission due to the Differential-Mode Currents

355 Common Mistakes that Lead to Unnecessarily Large DM Emissions

356 Common-Mode Current Emission E C,max I C = Kf L

357 Radiated Emission due to the Common-Mode Currents

358 Susceptibility Models

359 10V/m, 100MHz Incident Uniform Plane Wave

360 Measurement of Conducted Emissions

361 Line Impedance Stabilization Network (LISN)

362 Differential-Mode and Common-Mode Current Components

363 Methods of Reducing the Common-Mode Conducted Emissions

364 Definition of the Insertion Loss of a Filter

365 Four Simple Filters IL = 20 log 10 V ( V L, wo L, w ) = 20 log 10 ( ω L R S + R L )

366 Insertion Loss Tests

367 Conducted EMI Filter

368 Common-Mode Choke

369 The Equivalent Circuit of the Filter for Common-Mode Currents

370 The Equivalent Circuit of the Filter for Differential-Mode Currents

371 The Dominant Component of Conducted Emission ^ I Total = ^ I C ± ^ I D

372 A Device to Separate the CM and DM Conducted Emissions

373 Measured Conducted Emissions without Power Supply Filter

374 Measured Conducted Emissions with 3300pF Line-to to-ground Cap.

375 Measured Conducted Emissions with a 0.1μF F Line-to to-line Cap.

376 Measured Conducted Emissions with a Green Wire Inductor

377 Measured Conducted Emissions with a Common-Mode Choke

378 Nonideal Effects in Diodes

379 Construction of Transformers

380 The Effect of Primary-to to-secondary Capacitance of a Transformer

381 The Proper Filter Placement in the Reduction of Conducted Emissions

382 Crosstalk The unintended EM coupling between wires and PCB lands that are in close proximity. Crosstalk between wires in cables or between lands on PCBs concerns the intrasystem interference performance of the product.

383 Three-Conductor Transmission Line illustrating Crosstalk

384 Wire-type Line illustrating Crosstalk

385 PCB Transmission Lines illustrating Crosstalk

386 The Equivalent Circuit of TEM Wave on Three-Conductor Transmission Line

387 The Simple Inductive-Capacitive Coupling Model

388 Frequency Response of the Crosstalk Transfer Functions ^ NE ^ V S V = jω( R NE R + NE R FE R S L + m R L + R R NE NE R + R FE FE R R S C m + R L L ) = IND j ω( M NE + M CAP NE ) ^ FE ^ V S V = jω( R NE R + FE R FE R S L + m R L + R R NE NE R + R FE FE R R S C m + R L L ) = IND j ω( M FE + M CAP FE )

389 Effect of Load Impedance

390 Common-impedance Coupling ^ V ^ V ^ V ^ V NE S FE S = jω( M + M ) + IND NE IND FE CAP NE = jω( M + M ) + CAP FE M M CI NE CI FE

391 Time-Domain Crosstalk for R=50Ω

392 Time-Domain Crosstalk for R=1KΩ

393 The Capacitance Equivalent for the Shielded Receptor Wire

394 The Lumped Equivalent Circuit for Capacitive Coupling ^ CAP V NE V R R ^ CAP NE FE RS GS = FE jω VG DC RNE + RFE C RS + CGS C C

395 Illustration of Placing a Shield on Inductive Coupling

396 The Lumped Equivalent Circuit for Inductive Coupling ^ V IND NE = R NE R + NE R FE jωl GR ^ I G R SH R + SH jωl SH SF = R SH R + SH jωl SH

397 Explanation of the Effect of Shield Grounding

398 Twisted Wires

399 The Inductive-Capacitive Coupling Model

400 Terminating a Twisted Pair

401 A Model for the Unbalanced Twisted Receptor Wire Pair

402 Explanation of the Effect of an Unbalanced Twisted Pair

403 The Three Levels of Reducing Inductive Crosstalk

404 A Coupling Model for the Balanced Termination

405 The Effect of Balanced and Unbalanced Terminations

406 Purposes of a Shield To prevent the emissions of the electronics of the product from radiating outside the boundaries of the product To prevent radiated emissions external to the product from coupling to the product s electronics

407 Degradation of Shielding Effectiveness

408 Termination of a Cable Shield to a Noisy Point The cable shield may become a monopole antenna, if the ground potential is varying Peripheral cables such as printer cables for PC tend to have lengths of order 1.5m, which is a quarterwavelength at 50MHz Resonances in the radiated emissions of a product due to common-mode currents on these types of peripheral cables are frequently observed in the frequency range of MHz

409 Shielding Effectiveness R represents the reflection loss A represents the absorption loss M represents the additional effects of multiple reflections SE = R + A + db db db M db / transmissions

410 Reflection Loss R db 20 log 10 ( ηo ) 4η 20 log 10 ( 1 4 σ ωμ r ε o ) By referring to copper, R db = log 10 ( σ μ r r f ) The reflection loss is larger at lower frequencies and high-conductivity metals

411 Absorption Loss A db = 20 log 10 e t / δ = t fμ r σ r The absorption loss increases with increasing frequencies as f

412 Shielding Effectiveness

413 Shielding Effectiveness Reflection loss is the primary contributor to the shielding effectiveness at low frequencies At the higher frequencies, ferrous materials increase the absorption loss and the total shielding effectiveness

414 Shielding Effectiveness of Metals

415 The Methods of Shielding against Low-Frequency Magnetic Fields The permeability of ferromagnetic materials decreases with increasing frequency The permeability of ferromagnetic materials decrease with increasing magnetic field strength

416 The Frequency Dependence of Various Ferromagnetic Materials

417 The Phenomenon of Saturation of Ferromagnetic Materials

418 The Bands to Reduced the Magnetic Field of Transformer Leakage Flux

419 Effects of Apertures Since it is not feasible to determine the direction of the induced current and place the slot direction appropriately, a large number of small holes are used instead

420 ESD Events Typical rise times are of order 200ps-70ns, with a total duration of around 100ns-2μs The peak levels may approach tens of amps for a voltage difference of 10kV The spectral content of the arc may have large amplitudes, and can extend well into the GHz frequency range

421 Effects of the ESD Events The intense electrostatic field created by the charge separation prior to the ESD arc The intense arc discharge current

422 Three Techniques for Preventing Problems Caused by an ESD Event Prevent occurrence of the ESD event Prevent or reduce the coupling (conduction or radiation) to the electronic circuitry of the product (hardware immunity) Create an inherent immunity to the ESD event in the electronic circuitry through software (software immunity)

423 Preventing the ESD Event Electronic components such as ICs are placed in pink polyethlene bags or have their pins inserted in antistatic foam for transport Some products can utilize charge generation prevention techniques For example, printers constantly roll paper around a rubber platen. This causes charge to be stripped off the paper, resulting in a building of static charge on the rubber platen. Wires brushes contacting the paper or passive ionizers prevent this charge building

424 Hardware Immunity Secondary arc discharges Direct conduction Electric field (Capacitive) coupling Magnetic field (Inductive) coupling

425 Preventing the Secondary Arc Discharges

426 Single-point Ground

427 Use of Shielded Cables to Exclude ESD Coupling

428 The Methods of Preventing ESD-induced Currents

429 Reduction of Loop Area in Power Distribution Circuits

430 Reduction of Loop Areas to Reduce the Pickup of Signal Lines

431 Software Immunity Watchdog routines that periodically check whether program flow is correct The use of parity bits, checksums and errorcorrecting codes can prevent the recording of ESD-corrupted data Unused module inputs should be tied to ground or +5V to prevent false triggering by an ESD event

432 Packaging Consideration A critical aspect of incorporating good EMC design is an awareness of these nonideal effects throughout the functional design process Another critical aspect in successful EMC design of a system is to not place reliance on brute force fixes such as shielding and grounding

433 Common-impedance Coupling

434 The Effect of Conductor Inductance on Ground Voltage

435 Segregation of Grounds

436 Ground Problems between Analog and Digital Grounds

437 The Generation and Blocking of CM Currents on Interconnect Cables

438 Methods for Decoupling Subsystems

439 Interconnection and Number of PCBs It is preferable to have only one system PCB rather than several smaller PCBs interconnected by cables The PCBs can be interconnected by plugging their edge connectors into the motherboard

440 Use of Interspersed Grounds to Reduce Loop Areas

441 PCB and Subsystem Placement Attention should be paid to the placement and orientation of the PCBs in the system

442 Decoupling Subsystems Common-mode currents flowing between subsystems can be effectively blocked with ferrite, common-mode chokes Another method of decoupling subsystems is insert a filter in the connection wires or lands between the subsystems. This filter can be in the form of R-C packs, ferrite beads, or a combination High-frequency signals on the power distribution system between subsystems can be reduced by the use of decoupling capacitors

443 Splitting Crystal/ Oscillator Frequencies The 16 th harmonics (32MHz and MHz) are separated by 304kHz, so that they will not add in the bandwidth of the receiver The 100 th harmonic of the 2MHz signal (200MHz) and the 101 st harmonic of the 1.981MHz signal ( MHz) will be within 81kHz of each other and will add in the bandwidth of the receiver

444 Component Placement

445 Component Placement

446 A Good Layout for a Typical Digital System

447 Creation of a Quiet Ground where Connectors Enter a PCB

448 Unintentional Coupling of Signals between Chip Bonding Wires Placing a small inductor in series with that pin to block the high-frequency signal Ferrite beads could also be used, but their impedance is typically limited to a few hundred ohms

449 Use of Decoupling Capacitors

450 Decoupling Capacitor Placement

451 Minimizing the Loop Area of the Power Distribution Circuits