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1 A Ciculaly Polaized 6 GHz Micostip Antenna Hee is a mm-wave antenna suitable fo WLAN systems By V. A. Volkov, M. D. Panes Asco, and V. D. Koolkov and R. G. Shifman Resonance This aticle pesents the design of a micostip adiato-based ciculaly polaized antenna fo the opeation ange of 59 to 61 GHz. Chaacteistics of the antenna ae appended. This antenna, owing to its inheent simplicity, cheapness and small size, can be used in tansceives fo 6 GHz wieless local netwoks. High-speed and high-capacity inoffice wieless local netwoks have ecently become in lage demand. In addition, since the centimete waves ange has been congested, intenational ecommendations appeaed to bing some of the millimete wavelength subanges to commecial employment. Figue 1 shows the 6 GHz wieless local netwok that is intended fo indoo use, allowing data tansfe at the ate of 1 Mbit/s. Two micostip antennas (eceiving and tansmitting) ae connected to a high fequency module. Micostip antennas ae built fom a numbe of standad units, at the use station and fom a single unit, in the opeation sites distibuto. They adiate and eceive ciculaly polaized waves to minimize the influence of multiple eflections in the oom. In the opeation sites distibuto, the antenna beamwidth fom 6 to 8 degees would suffice to cove a athe lage teitoy. This aticle discusses the design sequence of a single-unit ciculaly polaized 6 GHz micostip antenna. A ciculaly polaized electomagnetic wave can be obtained using two mutually pependicula, equi-amplitude, linealy Ethenet Teminal Pinte Antenna Tansceive Seve Figue 1. A 6 GHz wieless local netwok. Notebook polaized waves that ae in phase quadatue elative to each othe. Then, if a micostip patch (MSP a squae-shaped micostip adiato) is excited on two sides by equi-amplitude oscillations with the phase diffeence of 9 degees between them, the total field adiated by all MSPs will be ciculaly polaized. The idea of cicula polaization esulting fom the excitation of two spatially othogonal modes in the antenna is the basis fo all otay polaization MSPs. Vaious appoaches can make the idea a eality. Pesented hee is the simplest technique: a quadatue hybid is used to divide the powe into two equi-amplitude pats in phase quadatue. Calculation and design Figue illustates the antenna design and method of mating with the UG-358/U standad 34 APPLIED MICROWAVE & WIRELESS

2 waveguide flange. A boad (1) (h =.5 mm in thickness), including both the adiato and the bidge, togethe with the.5 mm-thick dielectic laye () and the uppe sceen (3), fom a balanced stipline with b =.5 mm. The antenna pedestal is designed as a special configuation waveguide flange (4) with a.5 mm deep slot to eceive both the stip cicuit and the waveguideto-stipline adapte. A quate-wavelength waveguide shot (5) povides the optimal connection between the balanced stipline and the standad-size waveguide ( mm). The stip boad incopoates the micostip patch (adiato) and the quadatue hybid. In this case, the adiato and the matching quate-wave tansfome ae constucted on an unbalanced micostip line, while the hybid, the pill and the waveguide-to-stip adapte ae based on a balanced stipline. The Roges Cop. cladded sheet dielectic (R33, h =.5 mm, e = 3. ±.1) was selected as mateial fo the antenna. Fist, both the geomety and the fequency esponse of the micostip pat of antenna wee defined. Then the MSP with the quadatue hybid and calculation of polaization chaacteistics of the antenna was integated. Figue 3 depicts the antenna configuation and the six-pole equivalent cicuit whee 1 and 1' ae the input teminals and and ' and 3 and 3' ae outputs of highesistance pobes. The electic models do not take into account physical dimensions of the pobes consideing them as lumped elements. With the supposition of negligible spacing between exciting points compaed to the fee-space wavelength, it is obvious that complex amplitudes of the esulting spatially-othogonal electic field components E x and E y vay in popotion with the tansmission gains s 1 and s 13, espectively. That is, E x ~ s 1 and E y ~ s 13 o P x ~ s 1 and P y ~ s 13, whee P x and P y ae the aveaged densities of the powe fluxes tansfeed by the othogonal components E x and E y, espectively). Accodingly, estimation of MSP polaization popeties is educed to calculation of the full multipot scatteing matix, using the micowave device analysis pogam simila to that descibed in [1]. The pogam pemits calculation of the full multipot scatteing matix fom known s-matices of its components. Wheeas the pogam leaves oom only fo editing to allow new element definition subpogams to be enteed. We have coected the available desciption of the element the open end of micostip line, i.e. MSP, accoding to the teminology adopted in the pape. It is known that MSP can be modeled by the equivalent cicuit in the fom of a tansmission line section with the length that diffes fom the actual length L by the value l, which is the MSP extension on both ends, due to the edge capacitances (such as with allowance made fo existence of the edge fields). The tansmission line is loaded by the eal esistance R that is equivalent to the ohmic loss by enegy adiation. The dispesion influence on the quasi-static expansion l st value was studied in [], whee the authos have shown that, fo the millimete ange of wavelengths, the dispesion expansion l f appeas to be substantially shote than the l st value obtainable using a quasi-static appoximation. As shown in [3] and [4], the fequency depended wave impedance Z (f) and the effective dielectic constant vs. fequency e e (f) can be epesented as follows: Z ( f)= Z ε e whee t ( f )= ε G = Zt Z 1 + G f f p ε εe + G f 1 f p 5. Z Z 6 Metal sceen with opening Figue. Configuation of the antenna. Quate-wavelength waveguide shot Dielectic laye Pinted-cicuit boad with adiato Special-pupose flange Figue 3. Pinted-cicuit boad topology. (1) () (3) 36 APPLIED MICROWAVE & WIRELESS

3 Z f p = h In these equations, f p is measued in GHz, h is measued in millimetes, and Z and Z t ae measued in ohms. Z and Z t ae the wave impedance of the stipline of width (w), height (h and h), espectively. e e is the effective dielectic constant. Both Z and e e ae the known quasi-static values [1]. In the final analysis, the input eflectivity of the L + l (f) long egula line, when loaded by the eal esistance R, was computed fom the known fomula with egad to imaginay inhomogenities and to the effect of dispesion. A novel expession has been obtained fo R fom numeous expeimental obsevations: s 11 whee π β = λ 1 j tg L 1f = β( + ) 1+ j tg β L+ 1 ε ( ) e f [ ] [ ( f )] R = 1 6λ 9 Z f + ( ) ω ω 8π 1 λ e R R 1 1, fo ω < 35. λ + 1 (4) (5) (6) (7) lf h S11, db Figue 4. Theoetical etun loss fo the micostip patch, mm in size, with the matching quate-wave tansfome. ε e( f ) + 3. ω +. 64h =. 41 f ε ( ). 58 ω + 8. h e Ignoing dispesion Including dispesion Figue 4 gives estimated fequency dependencies of s 11 fo the micostip patch, mm in size, with the matching quate-wave tansfome, eithe with o without egad to dispesion effects. Thee ae two typical design altenatives fo the quadatue hybid. One topology povides a matched load available in the fouth isolated am; the othe has no load in the fouth am. Figues 5 and 6 pesent both the gain-fequency and the phase-fequency chaacteistics of the equivalent six-pole cicuit tansfe coefficients s 1 and s 13 fo both altenative quadatue hybid designs. (8) Powe, db S1 S13 Powe, db S1 S (ags1-ags13), deg 9 8 (ags1-ags13), deg Figue 5. The tansfe coefficient gain- fequency (a) and phase fequency (b) chaacteistics of the equivalent sixpole cicuit incopoating a matched load quadatue hybid. Figue 6. The tansfe coefficient gain-fequency (a) and phase-fequency of the equivalent six-pole cicuit incopoating a quadatue hybid with no load. 38 APPLIED MICROWAVE & WIRELESS

4 S11, db (a) Cicumfeence of adius = P max = W max Figue 7. Antenna s etun loss vs. fequency. The vital distinction between the show loaded cicuit fequency esponse and that of the unloaded cicuit povide clea evidence in suppot of the athe appaent but sometimes fogotten statement that any uneasonable idealization of the device to be developed is inadmissible at the simulation stage. al esults The topology vesion pesented in Figue 3 was accepted fo implementation as the best suitable to fit equiements of cicula polaization. It was supposed to use a dielectic wedge made of 16 m thick tantalum film as a matched load in the bidge cicuit. Shown in Figue 7 is the MSP fequency chaacteistic that was measued with the following waveguide-tostip adapte paametes: depth of the stip immesion in the waveguide.6b and the shot cicuit plane is to be offset by.1 l fom the stip, whee b is the width of the naow waveguide wall and l is the waveguide wavelength that coesponds to the cental fequency of the given ange (6 GHz). Polaization chaacteistics wee measued using a hon otating aound the longitudinal axis. The measuing eo due to the coss-polaization loss was about db. The peak powe esponse W was obseved unde such conditions when the hon plane E was paallel to the majo axis of the polaization ellipse and the minimum esponse W, when E was pependicula to that axis. Just as the amplitude atio = E x /E y, so the phase diffeence d between E x and E y wee detemined fom the expeimentally found values of W, W and t (the polaization inclination) using the elationships [5] that coelate two foms of the polaization ellipse desciption (canonic and paametic). It follows then that a polaization ellipse can be epesented paametically using the field components and basing on the measued values, as in the calculation: Ex E x sin ω t E = E sin ω t+ y = ( ) ( ) y 8 (9) (1) (b) (c) Figue 8. Polaization ellipses fo MSP waves adiated with fequencies (a) 59. GHz, (b) 6. GHz and (c) 61. GHz. Paametes of Polaization Cicumfeence of adius = P max = W max Cicumfeence of adius = P max = W max q, db t, degeed p c Table 1. Polaization ellipse paametes measued at the fequencies of 59, 6 o 61 GHz. whee E x and E y ae amplitudes of othogonal electic field components. Figue 8 shows both theoetical and expeimental polaization ellipses fo MSP adiated waves such as 59, 6 and 61 GHz. Table 1 gives polaization ellipse paametes, such as the axial atio q, the inclination t and the cicula polaization efficiency p c, that wee measued with the fequency of 59, 6 o 61 GHz. The cicula polaization efficiency was detemined fom the expession: 4 APPLIED MICROWAVE & WIRELESS

5 W W P c = W + W 11 1 (11) The micostip antenna patten is shown in Figue 9. Conclusion The poposed micostip antenna povides the equied beam width and cicula polaization paametes at the fequency 6 GHz. The stipline antenna design, including a waveguide input, pomises high manufactuability and low cost in mass poduction. The analytic elationships used fo computing eal and imaginay components of the MSP input impedance have been effective in estimation, with a high degee of pecision (> pecent), both of the adiato esonant fequency and of the adiato-to-fee space matching fequency band, allowing to avoid an exta iteation in designing pinted antenna topology. This aticle has demonstated essential diffeences in diectional couple fequency chaacteistics between eal devices with a matched load available in the isolated am and those with the missing fouth am. It is hoped that this finding will seve as a eminde of the typical designe's delusion a pioi abstaction of the device that is to be developed. -18 Ghz Dispesion Measuements on 19-1 Ohm Micostip Line on Sapphie, IEEE Tans. Micowave Tech., Vol. MTT- 4, August J. D. Kaus, Radio Astonomy, McGaw-Hill Book Company, Autho infomation V.A. Volkov and M.D. Panes ae employed with Asco, and V.D. Koolkov and R.G. Shifman ae employed with Resonance; both companies ae headquateed in St. Petesbug, Russia. All authos may be eached by phone/fax at M. Panes may be eached via at asco@ panes.spb.u, and M. Shifman may be eached at shifman@ esonance.spb.u. Asco s web addess is Refeences 1. K. C. Gupta, R. Gag, R. Chadha, Compute-Aided Design of Micowave Cicuit, Atech House, T. Itoh, Analysis of Micostip Resonatos, IEEE Tans. Micowave Tech., Vol. MTT-, Novembe B. Banco, et al., Fequency Dependence of Micostip Paametes, Alta Fequenza, Vol. 43, T. C. Edwads and R. P. Owens, NOVEMBER