Power electronic converters for motors with bifilar windings

Transcription

1 Power electronic converters for motors with bifilar windings Thesis submitted for the degree of Doctor of Philosophy at the University of Leicester by Weng Kwai Thong Department of Engineering University of Leicester February 2005

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3 CONTENTS List of figures...vi List of tables...x List of abbreviations...xi List of main symbols used... xii Acknowledgements...xiii Declaration of authenticity... xiv Abstract... xv Chapter 1 Background Introduction Review of motor technology Brushed motors Brushed dc m otors Universal motors Brushless motors Induction motors Brushless dc motors Doubly salient electronically commutated m otors Structure of thesis...14 Chapter 2 Power electronics Introduction Unipolar converters Clamped inductive load Split d c C dump topology Bifilar topology Asymmetric half-bridge... 26

4 2.2.6 Shared switch asymmetric half-bridge Bipolar converters Full-bridge and half-bridge converters Three-phase inverter Bipolar bifilar converter Power semiconductors Gate driver design Isolated gate drivers Non isolated gate drivers Summary...39 Chapter 3 Bifilar converter Operation and construction Practical considerations Winding design Snubber requirements Operation of an RCD snubber Summary Chapter 4 Bifilar converter for dual voltage applications - Part Introduction Dual voltage application Existing dual voltage powered drives Switched reluctance dual voltage powered drive The dual voltage lawnmower application Bifilar converters for dual voltage applications RCD Snubber C dump circuits Operation stages of a dual voltage drive with snubbers Stage one Stage two Stage three Mains RCD snubber, battery C dump with buck-boost configuration...64

5 4.7 Discussion of modes of operation Mains powered motoring Battery powered motoring Phase demagnetisation and energy recovery circuit Battery charging circuit Mains RCD snubber, battery C dump with buck configuration Discussion of modes of operation Mains powered motoring Battery powered motoring Phase demagnetisation and energy recovery circuit Battery charging circuit Implementation Motor construction Winding design Comparison between C dump subcircuits Power electronic converter Gate drive circuits Software design Experimental results Comparison of measured flux-linkage characteristics against design Determination of flux-linkage curves over a range o f rotor angles Dynamometer test results Lawnmower test results Battery powered drive with integral charging capability Conclusions Chapter 5 Bifilar converter for dual voltage applications - Part Introduction Power electronic converter Configuration of the bifilar windings Operation stages of a dual voltage drive with freewheel capability Stage one Stage two iii

6 5.4.3 Stage three Discussion of modes of operation Mains powered motoring Battery charging Battery powered motoring Implementation Control circuits Battery charging system Software design Experimental results Dynamometer test results Lawnmower test results Converter comparison Conclusions Chapter 6 Bifilar converter for bipolar excited motor drives Introduction RCD snubber Novel RCD snubber Discussion of modes of operation Comparison between conventional and novel RCD snubber Power dissipation Rate of transfer Size of capacitor Dependence of snubber performance on motor speed Experimental results Summary of novel RCD snubber Novel LCD snubber Description of the LCD snubber Discussion of modes of operation Implementation Sizing the snubber components Experimental results iv

7 Load test comparison between LCD and RCD snubbers Dynamic performance of LCD Snubber Comparison between LCD and RCD snubbers Summary of novel LCD snubber Conclusions Chapter 7 Conclusions and further work Conclusions Further work References V

8 List of figures Figure 1-1: Construction of a 4/2 flux switching m otor...12 Figure 1-2: Changing the polarity of armature current in FSM...12 Figure 2-1: R dump converter Figure 2-2: Dual decay converter...20 Figure 2-3: Split dc converter...21 Figure 2-4: C dump converter...22 Figure 2-5: Series connected C dump Figure 2-6: Modified C dump...24 Figure 2-7: Bifilar converter...25 Figure 2-8: Asymmetric half-bridge converter for single-phase and two-phase windings...27 Figure 2-9: Shared switch asymmetric half-bridge converter Figure 2-10: Half-bridge converter...29 Figure 2-11: Bipolar operation of half-bridge converter Figure 2-12: Full-bridge converter...30 Figure 2-13: Shared switch full-bridge converter [32]...30 Figure 2-14: Three-phase full-bridge inverter...31 Figure 2-15: Upgraded bifilar converter with bipolar capability Figure 2-16: High side device Figure 2-17: Opto-isolator with floating gate drive supply Figure 2-18: Pulse transformer gate driver...37 Figure 2-19: Bootstrap circuit Figure 3-1: Bifilar converter and its transformer equivalent circuit...40 Figure 3-2: Bipolar bifilar converter with secondary reflected through the transformer...42 Figure 3-3: Bifilar converter with RCD snubber...45 Figure 3-4: Principal operating modes of a bifilar wound single-phase inverter with a RCD snubber Figure 3-5: Experimental plots of RCD snubber at turn off of the switching device..49 Figure 4-1: Connection sense of bifilar windings vi

9 Figure 4-2: RCD snubber subcircuit Figure 4-3: C dump subcircuits...61 Figure 4-4: Dual voltage converter...61 Figure 4-5: Dual voltage converter with mains RCD and battery C dump buck-boost configuration Figure 4-6: Mains powered operating modes (motoring and charging) Figure 4-7: Battery motoring operating modes Figure 4-8: Dual voltage converter with mains RCD and battery C dump buck configuration Figure 4-9: Mains powered operating modes (motoring and charging) Figure 4-10: Battery powered operating modes Figure 4-11: Energy recovery modes...74 Figure 4-12: Stator and rotor laminations...76 Figure 4-13: Experimental waveforms for buck and buck-boost configurations 80 Figure 4-14: Circuit diagram of the power electronic converter...81 Figure 4-15: Schematic layout of the control circuit diagram Figure 4-16: Bootstrap start-up circuit for battery applications...86 Figure 4-17: Gate pulse transformer conditioning circuit...88 Figure 4-18: PWM duty cycles...89 Figure 4-19: Load algorithm...90 Figure 4-20: Power supply configuration...93 Figure 4-21: Comparison of measured flux-linkage characteristics against design...94 Figure 4-22: Mains winding flux-linkage characteristics for (a) rising inductance and (b) falling inductance...95 Figure 4-23: Static magnetisation curves over one rotor pole-pitch o f the mains winding...95 Figure 4-24: Experimental test results from dynamometer...97 Figure 4-25: Waveforms of dynamometer experimental results...99 Figure 4-26: Power flow analysis Figure 4-27: Comparison of flux linking the bifilar windings Figure 4-28: Breakdown of power flow in static iron loss test Figure 4-29: Hysteresis losses in the motor Figure 4-30: Battery acceleration Figure 4-31: Mains acceleration...105

10 Figure 4-32: Experimental waveforms for motor drive testing in the lawnmower deck 108 Figure 4-33: Waveforms during single pulse mode at full speed with no load Figure 5-1: Complete circuit diagram for the power electronic converter Figure 5-2: Forward conversion paths through discrete diodes during mains PVL..118 Figure 5-3: Forward conversion paths through anti-parallel diode of switches during mains PV L Figure 5-4: Principal operating modes of voltage doubler split dc converter Figure 5-5: Operating modes of the charging algorithm Figure 5-6: Timing diagram for instruction channel codes Figure 5-7: Schematic layout of the control circuit diagram Figure 5-8: Switching algorithms for battery charging Figure 5-9: Charging current and voltage profile of 3 battery charging system Figure 5-10: Experimental waveforms using 8/4 motor to charge the battery Figure 5-11: Battery discharge profile for 3 charging system Figure 5-12: PWM duty cycles Figure 5-13: Acceleration profile using main motoring Figure 5-14: Load algorithms Figure 5-15: Experimental results for 8/4 m otor Figure 5-16: Waveforms showing battery motoring for 8/4 m otor Figure 5-17: Waveforms showing mains motoring with battery present for 8/4 motor Figure 5-18: Waveforms showing different operating m odes Figure 5-19: Waveforms showing mains switch on with no battery present Figure 5-20: Comparison between converters presented in Chapter 4 and Figure 6-1: Single-phase inverter for bifilar winding with novel RCD snubber Figure 6-2: Principal operating modes of a bifilar wound single-phase inverter with a novel RCD snubber Figure 6-3: Experimental plots at turn off for novel RCD snubber Figure 6-4: RCD snubbers Figure 6-5: Experimental results at 2500 r/m in Figure 6-6: Experimental results at 5500 r/m in Figure 6-7: Experimental results showing snubber capacitor voltages at 5500 r/min

11 Figure 6-8: Single-phase inverter for bifilar winding with novel LCD snubber 175 Figure 6-9: Principal operating modes of a bifilar wound single-phase inverter with a novel LC snubber Figure 6-10: Experimental plots showing the LCD snubber operation in single pulse mode Figure 6-11: Load test comparison between LCD and RCD snubbers Figure 6-12: Output torque against motor speed curve Figure 6-13: Output torque against efficiency curve Figure 6-14: Output torque against input power curve Figure 6-15: Output torque against output power curve Figure 6-16: Snubber performance during start-up Figure 6-17: Snubber performance during acceleration Figure 6-18: Photograph comparing the size of the components used in the RC and proposed snubber circuits ix

12 List of tables Table 4-1: Target specification for motor design...57 Table 4-2: Winding specification...78 Table 4-3: Experimental results for buck and buck-boost configurations Table 4-4: Component list for the power electronic converter Table 4-5: Operating modes of the lawnmower Table 4-6: Inputs to the microcontroller...85 Table 4-7: Outputs from the microcontroller Table 4-8: Power supply requirements...92 Table 4-9: Experimental results at different output torque levels Table 5-1: Component list for the power electronic converter Table 5-2: Description of the optocoupler functions Table 5-3: Operating modes of the lawnmower system Table 5-4: Description of the optocouplers in each operating m odes Table 5-5: Description of communication and synchronisation between LV and HV microcontrollers Table 5-6: LV microcontroller inputs Table 5-7: LV microcontroller outputs Table 5-8: HV microcontroller inputs Table 5-9: HV microcontroller outputs Table 5-10: Switching algorithm for battery charging system Table 5-11: Comparison of 3 battery charging system Table 5-12: Mains switching algorithm Table 5-13: Converter comparison - components Table 5-14: Converter comparison - performance and functionality Table 6-1: Characteristics of novel and conventional RCD snubbers Table 6-2: Experimental results of novel and conventional RCD snubbers Table 6-3: Component rating and cost comparison between novel and conventional RCD snubbers Table 6-4: Component size and cost comparison between LCD and RCD snubbers 189 Table 6-5: Comparison between novel RCD and LCD snubbers x

13 List of abbreviations EMC : Electro-Magnetic Compatibility EMF : Electro-Motive Force FEA : Finite Element Analysis FSM : Flux Switching Motor HV : High Voltage IGBT : Insulated Gate Bipolar Transistor LCD : Inductor, capacitor and diode LV : Low Voltage MMF : Magneto-Motive Force MOSFET : Metal Oxide Silicon Field Effect Transistor NVL : Negative Voltage Loop PCB : Printed Circuit Board PVL : Positive Voltage Loop RCD : Resistor, capacitor and diode RPM : Revolutions per minute SRM : Switched Reluctance Motor ZVL : Zero Voltage Loop xi

14 List of main symbols used Symbol Description Units * Magnetic flux V e p CO Magnetic flux linkage Rotor angle Resistivity Radian frequency Weber, Wb Volt seconds, Vs Degree, 0 Ohm metre, Qm Radians per second, rads"1 B e f i L m m f P R S t T V Magnetic flux density Electro motive force Rate of frequency Current Inductance Magneto motive force Power Resistance Speed Time Torque Voltage Tesla, T Volt, V Hertz, Hz Ampere, A Henry, H Ampere turns Watt, w Ohm, Q Revolutions per second, rpm Second, s Newton metre, Nm Volt, V

15 Acknowledgements I would like thank my supervisor, Prof. Charles Pollock, without whom this PhD would not have been a success. I would also like to thank friends and colleagues of the power electronics research group at the University of Leicester and all industrial members of Centre of Advanced Electronically Commutated Drives. In particular, I would like to acknowledge Black and Decker, which has sponsored many of my research projects and has developed many prototypes throughout the research. Last but not least, I would like to extend my deepest gratitude and heartfelt thanks to my wife, Chloe and my son, Ethan, for their endless support all the way through my PhD. xiii

16 Declaration of authenticity The material of which this thesis is composed is the original work of the author except where reference has been made to prior publications. No part of this thesis has been submitted for a degree at any other institution. xiv

17 Abstract There is a high demand to improve performance and efficiency of motors used in domestic applications but there is constant downward pressure on manufacturing cost. This thesis identifies switched reluctance motors and flux switching motors as being low cost brushless motor technology and shows that bifilar windings can lead to lower cost electronic converters. The thesis develops innovative power electronic converters for motor incorporating bifilar windings. Novel circuits for reliable capture, dissipation or recovery of energy associated with leakage inductance of the closely coupled bifilar windings are presented. Two applications identified are dual voltage drives and bipolar excited motor drives. Several bifilar converter drives are presented in detail in this thesis. These include converters with dissipative snubbers, converters with energy recovery snubbers, converters with nondissipative snubbers and converters with inherent freewheel capability. Each of the presented converters offers advantages in cost and/or performance over existing converters. xv

18 Chapter 1 Background Chapter 1 Background 1.1 Introduction As the capabilities of power electronics have improved and their cost has decreased, it has become both possible and economical to produce converters providing variable voltage and variable frequency for brushless motors that provides high performance and economical variable speed drives. Many applications that have conventionally used fixed speed motors, such as pumps and vacuum cleaners, are now being converted to adjustable speed drives so that optimum speed can be chosen for each operating conditions, thus achieving higher system efficiency and energy conservation [1-3]. Other applications that require variable speed operations, such as washing machines and food mixers, often use the universal motors. The universal motor is a widely used brushed motor that is capable of variable speed operations with simple electronic control using triac and a few passive components [4], With the use of power electronics, brushless motors are capable of variable speed operation and can offer practical alternatives to the brushed motors. The variable speed drives have a higher initial capital cost and so must offer the user a significant improvement in performance in return [5]. Variable speed drives are used to drive loads at variable speeds and allow a motor to be accelerated in an efficient and controlled way. In all drives where the speed and position are controlled, a power electronic converter is needed as an interface between the input power and the motor. The power converter can be something simple, as in triacs used in universal motors, or something more complex, as in variable frequency, variable voltage, inverter used in induction motors. In any case, at full power variable speed drives will always have lower efficiency than the fixed speed counterpart 1

19 Chapter 1 Background operated directly from the power supply. This is due to the losses incurred in the power converter. However at other operating points the capability of variable speed drives to adjust speed and maintain it under different operating points will lead to energy savings and is essential in most applications of variable speed drives. The efficient use of electricity with the help of power electronics helps in reducing power consumption and correspondingly improving the environmental impact. Saving electricity not only gives financial benefit and cleaner environment, but prevents growth of new power generating capacity [6]. The market for low cost variable speed drive systems is huge as such systems power the majority of domestic appliances, from lawnmowers, electric showers, portable power tools to food mixers. Domestic appliances represent very cost conscious applications for drives where the emphasis is on achieving the optimum compromise between cost and performance. As the capabilities of power electronics have improved and as their cost has decreased, it has become both possible and economical for them to be used in domestic electrical drives markets. These markets are at the time of writing being dominated by brushed motors, or where low noise is a critical factor, by the induction motors. The brushed motors dominate the current market for low cost variable speed drive for various reasons: it can be manufactured relatively cheaply, it can be operated with only a few additional components and, it is an established well understood motor technology. Power electronics has reached an advanced level of development where it can be considered for use in motor technologies where previously only simple drives were considered economic. Each major development in electrical switching technology stimulates a matching development in drives as the characteristics of the new switch are exploited, from thyristors to bipolar power transistors to insulated gate bipolar transistors (IGBTs). Combining power electronics with microelectronic control circuitry also enables the use of motors with specific or complex control requirements to be considered. A number of different drive technology options have emerged for variable speed drives prompting an evaluation of the available motor technologies to identify opportunities to realise the best solution for a low cost drive. 2

20 Chapter 1 Background 1.2 Review of motor technology Many factors influence the choice of the most suitable variable speed drive for a particular application. Variable speed drives can be subdivided to two main classes, namely the brushed and brushless drives. Brushless drives are a class of motors where the excitation of the rotor does not require an electrical contact. This is in contrast with brushed motors where carbon brushes and commutator in the motor is used to switch current to different windings to provide the rotor excitation required. The resultant rotor field then interacts with the stator field to create a continuous rotating motion. 1.3 Brushed m otors Brushed dc motors The dc motor is a rotating electric machine designed to operate from a source of direct voltage. A dc motor consists of 3 mains parts: Stator: permanent magnets or field windings are used to generate the field flux. Rotor: composed of armature windings that are supplied in current via the carbon brushes Carbon brushes and commutator: mechanical commutators linking dc power supply and armature windings In a dc motor, the opposite magnetic polarities of the energised armature and stator winding attract and the rotor rotates to reduce the distance between stator and rotor fields. As the rotor rotates, the brushes move across the commutator contacts and energise the next set of armature windings, thereby achieving a continuous motion. 3

21 Chapter 1 Background There are three main types of dc motors: permanent magnet, series wound and shunt wound dc motors. These motors use a similar rotor with brushes and a commutator. A series wound dc motor has the stator windings in series with the armature windings on the rotor. A shunt wound dc motor has the stator windings in parallel with the armature winding. A permanent magnet dc motor uses permanent magnets instead of field windings to provide the stator excitation, thereby achieving higher efficiency [7]. There are two ways to adjust the speed of a dc motor using field or armature control. Combinations of the two are sometimes used to achieve speed control of the dc motor. Varying the field current allows the speed range of the dc motor to be increased by field weakening. Reducing the field current of the dc motor increases speed and reduces output torque for a given armature current. The armature voltage, and hence the armature current, can be controlled by a single switch dc chopper. The circuit can be easily extended to provide four-quadrant operation using a full-bridge consisting of four power switches. The armature voltage and polarity can then be adjusted using the Pulse Width Modulation (PWM) technique, enabling the speed and direction of the motor to be controlled Universal motors A series wound motor is also called a universal motor. It is universal in the sense that it will run equally well using either an ac or a dc voltage source. When run from an ac source, both the field flux and armature current reverse with each crossover of the ac supply. This produces a unidirectional though pulsating torque [8]. In terms of construction the motor is similar to the brushed dc motor, but laminations are required for the static magnetic circuit on the stator to reduce the iron losses due to the alternating ac supply. The rotor of both universal and dc motors has to withstand alternating flux and therefore has to be laminated. The universal motor can be controlled either as a phase-angle drive or as a chopper drive. In the phase-angle application, the phase-angle control technique is used to adjust the ac voltage applied to the motor. A phase shift of the gate pulses allows the 4

22 Chapter 1 Background effective voltage, seen by the motor, to be varied. The phase-angle drive requires just a triac. In the chopper application, PWM technique is used to adjust the voltage applied to the motor. Modulation of the PWM duty cycle allows the effective voltage, seen by the motor, to be varied. Compared to a phase-angle drive, a chopper drive requires a more complicated power stage with an input rectifier, a switch and a diode. The advantage is higher efficiency, less acoustic noise and better Electro-Magnetic Compatibility (EMC) behaviour [9]. Universal motors are often found in vacuum cleaners, portable power tools, food processors, mixers and other small devices operating over a speed range of 3,000 to 30,000 rpm. Its high speed capability gives the universal motor a high power to weight ratio making it attractive in portable appliances. The series universal motor and brushed dc motor are currently used in many applications as a low cost variable speed drives. The brushes and commutators make them unreliable but the speed can be controlled simply using a triac controller or a single switch dc chopper. 1.4 Brushless motors In a brushless motor, rotor excitation is achieved by an electromagnetic field alone, which can be either induced or supplied by a permanent magnet. The absence of brushes to provide rotor excitation eliminates the friction between the brushes and commutator. This provides the benefits of low mechanical wear, high reliability and high speed capability. Additionally, the lack of mechanical switching action in a brushless motor means that there is no electrical arcing, hence radiated noise is low and operation is safe in hazardous environment. Robustness, high torque and speed, and lower maintenance of the brushless motors therefore make them attractive compared to the brushed motor drives [10], 5

23 Chapter 1 Background Induction motors The stator of an induction motor has sinusoidally distributed windings and there are two types of rotor construction. A wound rotor has winding of the same number of poles as the stator winding. A cage rotor has a distributed winding in which all the conductors are connected together at each end by end rings. Simplicity and low cost of rotor construction have made the squirrel cage induction motor a popular construction. A revolving magnetic field is produced by the alternating currents in the distributed winding of the stator. Induced currents in the shorted conductors of the rotor cage also produce a rotating magnetic field. The interaction of the rotor field with the stator field generates torque. The induction motor is said to be an asynchronous motor as the rotor must rotate at a lower speed than the field generated by the stator to produce an output torque. The speed of the induction motor primarily depends on the frequency of the ac supply while the torque produced by the induction motor is determined by the amount of slip, which is determined by the difference in rotation between the rotor and stator fields. The speed of induction motor has traditionally been altered by having additional sets of coils or poles in the motor that can be switched on and off to change the speed of magnetic field rotation [9]. However, developments in power electronics mean that the frequency of the power supply can also now be varied to provide a smoother control of the motor speed. This method of speed control is however more expensive as the electronics required to control a variable frequency ac inverter drive are considerably more expensive than those required to control a dc motor. Single-phase induction motors The single-phase induction motor has a main winding carrying most of the load current and an auxiliary winding for starting. This is because if a single alternating voltage were applied to a single-phase distributed winding, two equal rotating fields, one in the forward direction and one in the reverse direction, would be produced. The 6

24 Chapter 1 Background rotor will carry current induced by the stator field, but there will be two equal and counter rotating torque fields. This will cause the rotor to vibrate but not to rotate. In order to rotate, there must be a resultant torque field rotating in one direction only. Starting is therefore achieved with an auxiliary winding, phase shifted from the main winding to produce a net starting torque. The phase shift between the auxiliary and main windings is produced using a high resistance winding or series capacitance. In a split phase motor, the auxiliary winding has a higher ratio of resistance to reactance to achieve the phase shifting required for starting. In capacitor start motors, the series capacitance is used only for starting and is cut of by a centrifugal switch once the speed is near the synchronous speed. In other cases, two capacitors are used, one for starting and one for running. The capacitor in the capacitor type motors contributes significantly to the overall cost of the motor. Starting capacitors which are switched out are normally short-time rated whilst the running capacitors are full-time rated and consequently more expensive. The ease of operating the single-phase induction motor drives at fixed speed makes the drive a good low cost option for many domestic appliances. However the presence of the backward rotating field has complicated the variable speed control, resulting in single-phase induction motors not being widely used in variable speed drives [11]. In addition, the interaction between forward and backwards fields produces a pulsating torque at twice the line frequency, which produces a humming noise that is not present in polyphase machines [9]. Three-phase induction motors A three-phase induction motor has three-phase stator winding connected to a threephase supply. The phase differences between the three phases of the polyphase electrical supply is utilised to create a rotating electromagnetic field in the motor. Unlike single-phase induction motors, three-phase induction motors therefore selfstart and use no capacitor, start winding, centrifugal switch, or other starting device. 7

25 Chapter 1 Background As part of a power electronic drive, a three-phase induction motor is not required to operate from a three-phase supply, but the three-phase supply can rather be derived using a three-phase inverter running from a dc bus voltage rectified from the widely accessible single-phase supply. This releases the three-phase induction motors from the requirement of three-phase supplies, and enables three-phase induction motors to be considered as a viable option for domestic and commercial drives. Three-phase inverters would be used to provide variable voltage variable frequency supply to achieve speed control of the three-phase induction motors. Three-phase induction motor has been the workhorse of the industry and these motors are readily available at competitive prices. Although the cost of a three-phase inverter is much higher than a single-phase inverter, the reducing cost of power electronics and increasing importance of other factors such as noise and efficiency has seen threephase induction motor being adopted for a washing machine application [12] Brushless dc motors The brushless dc motor can be described as an inverted permanent magnet brushed dc motor. Although they do not operate from a dc voltage source, their name comes from the fact they operate in a similar way as brushed dc motors but inverted inside out. The rotor carries the field flux producing permanent magnets while the stator holds the normally rotating armature windings stationary. In the brushed dc motor, the current polarity is altered by the commutator and brushes. In contrast, in the brushless dc motor, the polarity reversal is performed by electronic commutation of the stator windings in synchronism with the rotor position. The requirement of the rotor position sensing and electronic commutation is the reason why brushless dc motors need power electronic controller for proper operation. The expensive cost of the permanent magnet rotors is the principal disadvantage of brushless dc motors but this effect is diminishing with the reducing cost of permanent magnets [7].

26 Chapter 1 Background Doubly salient electronically commutated motors Another subclass of the brushless motor family is the doubly salient electronically commutated motors. Doubly salient motors develop torque by the movement of the rotor, with respect to the stator, to a position of minimum reluctance to the flux set up by the current on the stator windings. The doubly salient laminated structure consists of stator and rotor with copper windings on the stator. The rotor construction is simple and robust, as it contains pure laminated steel with neither permanent magnets nor electrical windings. The stator can be easily manufactured as the windings are not distributed but concentrated around salient poles. The concentrated windings around salient pole structure are a simpler and more costeffective structure than distributed windings in an ac machine and allow higher efficiency to be achieved [13]. Switched reluctance and flux switching motors are two examples of doubly salient electronically commutated motors. The doubly salient, singly excited requirements lend the motors the simplest and lowest cost motor option available to date. These motors appear to be the ideal candidates for low cost drives and would be explored in further detail. Switched reluctance motors Locomotives in the early 19th century were the first machines to use Switched Reluctance Motors (SRM). They did not perform satisfactorily and have faded into obscurity. These motors reappeared in the early 1980s, with the development of electronic controllers for brushless motors. Construction and Operation The SRM consists a rotor and stator with a coil winding in the stator. The rotor, which consists of a laminated permeable material with teeth, is a passive device with no coil winding or permanent magnets. The stator typically consists of slots containing a 9

27 Chapter 1 Background series of coil windings, the energisation of which is electronically switched to generate a moving field. For the most part, only a single coil set is activated at any one time. Torque is produced by the reluctance alignment principle. When one stator coil set is on, a magnetic flux path is generated around the coil and the rotor. The reluctance property causes the rotor to be attracted in order to align itself to a minimum reluctance position with an applied magnetic field. Therefore, the iron rotor of a switched reluctance motor rotates to remain aligned with the rotating magnetic field, produced by electronically commutated current through consecutive stator winding. Since the torque is produced by minimisation of the reluctance in the magnetic circuit, the direction of excitation of either phase coil is not important. The freedom in choosing the current flow direction makes it possible to use a unipolar converter to drive the switched reluctance motor. Unipolar converter is generally simpler and requires fewer switches than the corresponding bipolar drive. It is also more faulttolerant as there are no shoot through paths if the switches fail. This reduces the converter cost because of the reduced complexity of the power electronic switching circuits and improves the reliability of the drive because of the inherent shoot-through fault immunity. The fault tolerant feature is applicable not only to the converter, but also to the motor. The motor is inherently resistant to overload and immune to singlepoint failure because of the absence of internal excitation or permanent magnet and the independent nature of torque production in the various phases. The SRM offers better performance than many other types of motors. The production of torque by reluctance alignment principle is efficient and provides good dynamic response [13]. A SRM does not require sinusoidal exciting waveforms for efficient operation, so it can maintain higher torque and efficiency over broader speed ranges than is possible with other advanced variable-speed systems. The optimal waveforms needed to excite a switched reluctance motor are typically the result of a fixed voltage applied to the motor coils at predetermined rotor angles. With the appropriate switching and energisation of the stator coils, such waveforms can be achieved at virtually any speed and they can be programmed to precisely match the loads they serve. 10

28 Chapter 1 Background In addition, as long as the commutation can be accurately controlled with respect to the rotor angle, the motor will operate at its predicted high efficiency. It is the injection of current pulses to the motor winding at the correct time that produces torque. The position of the rotor relative to the stator must therefore be known and this is most simply achieved by using a position sensor, like slotted optical sensor, encoder, Hall effect sensor, etc. It should be noted that there is continuing work on the sensorless control of the switched reluctance motor, as the implementation of sensors increases costs and decreases system reliability [14]. Flux switching motors The Flux Switching Motor (FSM) [15] is a new class of electric motor, and is a unique combination of an inductor alternator [16, 17], and a two-phase switched reluctance motor with fully pitched windings [18]. It uses the reluctance principle for rotation, but exhibits motor characteristics similar to those of a dc motor. Construction and Operation The FSM is shown in Figure 1-1 with four stator teeth and two rotor teeth. There are two windings in this machine. These shall be termed field winding, labelled F, and armature winding, labelled A. The two windings are wound over two stator teeth, creating a fully pitched winding, as opposed to conventional switched reluctance motor with short pitched winding wound concentrated around one stator tooth. The two-phase fully pitched winding arrangement is similar to that used in inductor alternators [16]. 11

29 Chapter 1 Background Figure 1-1: Construction of a 4/2 flux switching motor The field winding is fed with direct current at all times. This establishes a two-pole magnetic field. The other two slots contain an armature winding. The direction of current in the armature winding determines which two of four stator poles carry flux and hence the position of the rotor. With the aid of Figure 1-2 the principle of operation of the FSM will be explained. > \ i v I t! OF > (a) F ield p ositive, armature positive Figure 1-2: Changing the polarity of armature current in FSM 12

30 Chapter 1 Background In Figure 1-2 (a) the field winding, F, is carrying positive current and the armature winding, A, is also carrying positive current. The top and bottom stator pole pair carry the flux and will pull the rotor into alignment as shown. In Figure 1-2 (b) the field winding is still carrying positive current, but the armature winding is reversed and carries negative current. The quadrature set of stator poles carries the flux and pulls the rotor into the other aligned position as shown. Flux is always present in the FSM since the motor always carries dc current in the field winding. The direction of the armature current, simply switches the flux between two sets of alternate stator teeth. It is for this reason that the motor is being termed the flux switching motor. Even though the structure of the machine is very similar to a switched reluctance topology, the method of flux control is very different. The FSM shares many benefits of the switched reluctance motor, namely the doubly salient, singly excited structure with no conductors or permanent magnets on the rotor. The unique benefit of the flux switching motor, as with the inductor alternator it is derived from, is that one of the two windings can carry dc current leaving only one winding requiring electronic control. This dc field winding can be energized continuously with dc current, providing the magnetizing energy to the motor at all times without increasing the kva requirements of the power electronic controller. However the armature winding has a requirement for bipolar phase current as the current needs to be reversed to give the required resultant flux orientation for rotor rotation. As with the brushed dc motor, the flux switching can be shunt or series configured. The field winding can be connected either in series with the armature winding or with a higher number of turns, in shunt, across the field supply. In the shunt mode, the voltage applied to the field can be controlled by a single switch dc chopper which offers the signifcant advantages of field control (e.g. field weakening) during normal operation. 13

31 Chapter 1 Background The flux switching drive is seen as an advancement in low cost brushless machines, particularly because of its low cost converter topologies. It retains many of the benefits of the switched reluctance machine in terms of high power to weight ratio, variable speed, rugged simple construction and low manufacturing cost. One of the mains drawbacks of the switched reluctance motor is the cost of the power converter. Current has to be injected at the beginning of the motoring stroke and withdrawn at the end of the motoring stroke. A substantial magnetising energy results in the high kva ratings of the power devices in the switched reluctance drives. In flux switching motors, the field winding is continuously energised with dc current it can be connected to a dc source without the need for electronic control. The energisation of the field winding without requiring power electronics, leads to a dramatic reduction in the rating and the cost of the power electronic converter relative to a switched reluctance motor. 1.5 Structure of thesis This chapter has highlighted the huge demand of low cost variable speed drives in the domestic appliance market. Brushless drives offer many advantages over brushed drives, namely robustness, high torque and speed, and lower maintenance. The brushless drives however require more sophisticated control for variable speed operation than the brushed drives, resulting in brushed drives dominating the current domestic market. With the advent of power electronics, opportunities exist for many brushless drives to penetrate the position currently held by brushed motor in the domestic appliance market. Switched reluctance and flux switching motors have a simple, low cost construction and are therefore ideally suited for applications in the mass-produced domestic appliance market. If a suitably low cost power converter solution can be identified to match the low manufacturing cost of the motors, the complete drive can offer a very low cost solution to rival the brushed drives in the domestic appliances market. 14

32 Chapter 1 Background A review of all the relevant converter topologies is presented in Chapter 2. The optimum compromise between cost and complexity of a domestic appliance drive depends on the target application. The issues surrounding the selection and the gate driving requirements of the power semiconductor switches are also discussed. The bifilar converter is identified as the best candidate for producing low cost drives for domestic appliances and is investigated in detail in Chapter 3. Several bifilar converter drives were constructed and are presented in detail in Chapter 4, 5 and 6. Each drive offering advantages in cost and performance over existing drives. Chapter 4 presents bifilar converters with energy recovery snubbers suitable for dual voltage applications. The search for suitable bifilar converter for dual voltage applications continues in Chapter 5 with the presentation of bifilar converters with inherent freewheel capability. Chapter 6 presents low cost bifilar converters with novel snubbers for motors requiring bipolar excitation such as brushless dc and flux switching motors. Chapter 7 highlights the main conclusions of the thesis. The author s contribution is described and some areas of future work are proposed. 15

33 Chapter 2 Power electronics Chapter 2 Power electronics 2.1 Introduction Electrical machines have been controlled through power electronic circuits since the early days of mercury-arc-type rectifiers. The development of power semiconductors has increased such application enormously. In general, the function of the power converter is to provide some means of increasing and decreasing the supply of current to the phase winding, and hence, the electromagnetic torque produced by the motor. Efficient energy conversion requires that the semiconductor devices in the power converter be operated as switches rather than in their linear range. This is desirable when processing power as ideally the converter should not consume energy. In reality this is not the case and any losses must be kept to a minimum as the losses would not only incur the cost of wasted energy but also the cost of removing the heat generated from the losses. Power converters are made up of standard elements such as capacitors, resistors, inductors, diodes and the power devices. A large number of combinations of standard elements are possible resulting in a large number of different topologies being proposed. Each topology has advantages and drawbacks which must be evaluated against the demands of the target application and its benefits over other topologies. The choice of converter for a certain application is usually based on finding the lowest cost arrangement that satisfies the requirements o f the drive specification. For a given winding, the applied voltage determines the rate of change of current. The converter manipulates power flow by modifying the voltage applied across the motor phase windings and thereby allowing the converter to regulate the magnitude and even the waveshape of the current. The converter achieves this by using three modes of operation, namely positive, zero and negative voltage loops. 16

34 Chapter 2 Power electronics Positive Voltage Loops Positive voltage loops occur when switching devices associated with a phase winding are closed. The supply voltage is connected across the phase winding and the current in the phase winding increases rapidly, supplying energy to the motor. Zero Voltage Loops Zero voltage loops occur if the power devices are closed to shunt the phase winding while current is still flowing in the a phase winding. In this case the current continues to flow through the power devices. Energy is neither taken from or returned to the dc supply. The voltage across the phase winding during this time is equal to the on-state voltages of the power devices devices. This voltage is very small compared to the supply voltage and so the current in the phase winding decays very slowly. Negative Voltage Loops The final mode of operation is a negative voltage loop. The power devices are turned off resulting in a negative voltage being applied to the phase winding, decreasing the current in the phase winding. Not all modes are applicable to all topologies. Positive and negative voltage loops are present in all converters to increase and decrease current in the phase windings while zero voltage loop is only available in some converters. The capability of zero voltage loops is very desirable in the topologies as it allows soft chopping. A soft chopping scheme employs a zero voltage loop to maintain the phase current close to the reference level once a positive or negative voltage loop has been used to reach the level. The small on state voltages across the power devices allow a certain rate of change of current under zero voltage loop conditions, but this level is much smaller than that experienced under a full voltage loop. Without zero voltage loops, the converter would only be able to regulate the phase winding current using just the positive and negative voltage loops. This is known as a 17

35 Chapter 2 Power electronics hard chopping scheme, which forces large changes in the current over a chopping cycle. The large current ripple in a hard chopping scheme exacerbates the acoustic noise and core losses within the motor. Many different topologies have emerged with reduced number of switches and faster commutation time through continued research [19]. In general, the alternative topologies result from attempts to minimise drive cost by reducing the number of power switching devices per motor phase. The fewer power switching device, the lower the cost, not only in terms of device cost, but also in terms of the required gate drives, protection, heat sinks and snubbing associated with each device. However there has always been a trade-off between gaining some of the advantages and losing some with each topology. Only voltage fed converter topologies will be considered in this thesis. Current source converter topologies use large inductor in the dc link, which makes the input appear as a current source to the converter. These large and costly inductors preclude the current source converters from consideration for the low cost drives. The resonant converters are not considered in this thesis as the increased complexity of resonant converters both in terms of components and controls makes them unsuitable for low cost drives. The selection of a power electronic converter topology and its control depends on the type of motor drive. The vast majority of power converter topologies can be split into two main categories, unipolar and bipolar [20]. Unipolar types can only control current flow in one direction whereas bipolar types have bi-directional control of the current. Unipolar converters are more fault-tolerant due to the absence of shootthrough currents. In general, unipolar converters are simpler and use fewer switching devices compared to the bipolar converters and therefore have often been proposed for low cost drive. As described in Chapter 1, switched reluctance drives can operate with unipolar excitation, but flux switching drives require bipolar excitation. The excitation requirements have a large influence over the choice of appropriate power converter for a drive. 18

36 Chapter 2 Power electronics 2.2 Unipolar converters Clamped inductive load The simplest form of unipolar topology is to use a switch and a diode connected to the motor winding as shown in Figure 2-1. sw Figure 2-1: R dump converter Current will build up in the motor winding when the switch is on, and current will decay through the freewheel diode when the switch is off. The time constant of the winding,, where R is the winding resistance, will determine how fast the current can be increased and decreased. Additional suppression resistance, R, is often added in series with the diode to shorten the commutation period, by reducing the time constant to + R') during commutation. This increases the suppression voltage across the winding during turn off, but this is inefficient and the suppression voltage decays with the current, decreasing the rate of fall of current during commutation. Furthermore extra losses are dissipated away from the motor resulting in poor efficiency [21]. A capacitor can be added to form an RCD (Resistor, capacitor and diode) snubber to avoid the long current tail, as the capacitor voltage rises as it is being charged by the commutation energy. 19

37 Chapter 2 Power electronics A further variation in an attempt to improve the efficiency involves the addition of a switch in parallel to the suppression resistor in the so-called dual decay converter, shown in Figure 2-2 [22]. The inclusion of the switch adds zero voltage loop capability to the dual decay converter. When the switch is closed, current flows through it instead of the suppression resistor. Closing the switch during the chopping keeps the energy in the winding and prevents it from being dumped in the suppression resistor, thereby reducing the energy dissipation in the suppression resistor. Figure 2-2: Dual decay converter Split dc Adding resistors in the freewheeling path reduces the commutation period but the amount of power wasted is not insignificant. The energy in the resistor is dissipated as heat and even in medium sized appliance drives, the cost of heat extraction may be unbearable. In the split dc topologies, instead of dumping the energy into resistors at turn off, energy freewheels into a capacitor, creating a second voltage rail [10]. The two capacitors are then connected in series to form a split dc topology as shown in Figure 2-3. The motor windings are connected to the central tap point of the split dc link. The other ends of the windings are each connected to a switching device and associated freewheeling diode. During energisation the phase energy is supplied by

38 Chapter 2 Power electronics one of the capacitor while during commutation the phase demagnetises into the other capacitor. The energy flow to and from the capacitors must be balanced to maintain the common point voltage. This results in some loss of phase winding switching independence. C2 sw P h a sel P hase2 C1 Figure 2-3: Split dc converter The split dc topology achieves the minimum of one switch per phase without adding extraneous passive components or sacrificing control flexibility or efficiency. However, the phase number must be even, and the converter does not tolerate a phase imbalance or fault in any phase because any such fault results in voltage build-up in the capacitors. Zero voltage loops are not possible which means that hard chopping would be used to regulate the phase winding current. This raises the switching frequency for a given current ripple. In addition, each switching device and freewheel diode must be rated to withstand the complete supply voltage but only half the amplitude of the supply voltage is applied to each phase during conduction, resulting in poor dc bus voltage utilisation C dump topology C dump topology uses a capacitor to capture the commutation energy as in split dc topology. This creates a second voltage rail by a capacitor into which the 21

39 Chapter 2 Power electronics commutation energy flows, but instead of energising another phase to discharge the capacitor, it uses a dc chopper to return the energy captured in the capacitor back to the source [23]. The C dump therefore requires a dc chopper consisting of a high side switch, diode and inductor, as shown in Figure 2-4. sw chopper G dump P hase v d c A SW Figure 2-4: C dump converter During the phase demagnetisation, the voltage across the phase winding is the difference between the dc source voltage and the capacitor voltage. The capacitor voltage is generally maintained at twice the supply voltage, in order to supply the full reverse voltage to the off-going phase. The dump capacitor voltage regulation make the control system complex; however, the availability of the low cost microprocessors solves this problem to a great extent [24]. The drawbacks of this converter are the high switching device voltage ratings, the expense of the additional switch, the dump capacitor and inductor, and the losses associated with the dc chopper. Various forms of the C dump converter have appeared in the literature [23, 25, 26]. The common factor in all these C dump topologies is that the energy from the offgoing phase is dumped into a capacitor to achieve fast demagnetisation. The energy is then either returned to the source from the capacitor [26] or recirculated into another 22

40 Chapter 2 Power electronics phase, speeding up the current build-up in the next winding [25]. The dump capacitor can also be connected in series as shown in Figure 2-5, for which the dump capacitor voltage has to be maintained at the supply voltage, not twice the supply voltage as the negative voltage loop does not flow via the dc link capacitance. C2 S2 L2 Phase D2 VD C C1 S1 Figure 2-5: Series connected C dump Modified C dump converter, shown in Figure 2-6 [25], is a topology derived from basic C dump by eliminating the inductor in the dump circuit. The winding commutation is first being captured in the dump capacitor as in C dump circuit, but instead of recovering the energy to the source, it is being transferred into the next phase, providing a faster magnetisation than is achievable from the dc source. As in C dump circuit, the capacitor voltage is maintained at twice the supply voltage in order to supply the full reverse voltage to achieve fast phase demagnetisation. The converter is capable of zero voltage loops, reducing hysteresis losses and capacitor current ripple. However, this mode of operation is dependent on whether or not the capacitor voltage is above the reference value. 23

41 Chapter 2 Power electronics D2 S2 D1 VDC Phase C1 S1 Figure 2-6: Modified C dump Two further modifications have recently been added to the C dump converter [26], The first circuit requires one additional diode but improves the C dump circuit by reducing the ratings of the semiconductor switch, dump capacitor and inductor. The dc source is not in the demagnetisation path and the energy in the off-going phase is dumped directly into the capacitor, allowing smaller capacitor and inductor to be used. Zero voltage loop is also possible through the energy recovery switch, reducing hysteresis losses and improving converter efficiency. The second modification is similar to the modified C dump in that the inductor is eliminated by preventing the capacitor from discharging into the dc source. Instead the energy is being used in the tum-on of the next phase. This is achieved by adding a blocking diode in series with the dc source [26] Bifilar topology The complexities of a split rail, or C dump converters can be avoided by splitting the motor winding. Bifilar converters can be used in which two windings and a single supply can be used rather than one winding and two supplies [19, 27, 28], The two 24

42 Chapter 2 Power electronics windings allow the magnetising action and demagnetising actions of the converter to be separated so that a single supply can be used for both magnetisation and demagnetisation The winding used for magnetisation is connected to a switch while the winding used for demagnetisation is connected to a diode, as shown in Figure 2-7. The bifilar winding utilises the magnetic coupling between two closely coupled coils to transfer current between the coils. vdca sw Figure 2-7: Bifilar converter Turning on the switch builds up current in the switch winding. When the switch is turned off, the magnetic flux generated by the current in the switch winding couples to the diode winding. Current flows in the diode winding back to the dc source. The demagnetisation path includes dc source so that current fall times are short, as with split rail topologies, but without the complication of the split rail. There are three main drawbacks to the bifilar topology. Firstly, only one winding is carrying current at any one time, so for a given stator slot, the copper utilisation is reduced resulting in the increased effective winding resistance. Secondly, the voltage rating of the switching device and freewheel diodes must be at least twice the rating of motor windings. Thirdly, the leakage inductance between the paired windings results in large voltage spikes when the switch is turned off, so a safe path for the energy stored in the leakage inductance must be provided. 25

43 Chapter 2 Power electronics The bifilar converter uses the magnetic recovery property of closely coupled coils to keep the component count of other elements in the system low, thereby meeting the minimum switch requirement for a unipolar converter with one switch per phase ratio. Furthermore, it requires ground-referenced switches which result in a simple and low cost solution. Power converters that uses only ground-referenced switches are attractive because the simple driving requirements of ground-referenced switches not only reduces the number of components, but also reduces the complexity of the converter Asymmetric half-bridge By far the most versatile topology for unipolar converters is the asymmetric halfbridge topology [28-30]. The major advantage with this circuit is that all the available supply voltage can be used to control the current in the phase windings. The converter is capable of applying the full supply voltage across the winding in either polarity and therefore has full current pulse programming capability to turn the current in each phase both on and off. As each phase winding is connected to its own asymmetric half-bridge there is no restriction on the number of phase windings. The converter offers complete independence in phase current regulation and full regenerative capability. However, the converter suffers from high switch per phase ratio and is therefore expensive. The converter is best suited to motors with few phase windings. One switch connects one end of the winding to the positive supply. This switch is termed a floating switch since it does not have its source or emitter connected to the power supply ground. The drive circuit for the floating switch is more expensive and complicated than the ground-referenced switch since it has to be either isolated or level shifted. The second switching device is referenced to ground and connects the other end of the winding to the lower supply. 26

44 Chapter 2 Power electronics it Dz S3 VDC- C1 11 o: 02 Figure 2-8: Asymmetric half-bridge converter for single-phase and two-phase windings The asymmetric half-bridge uses three main modes of operation, namely positive, zero and negative voltage loops to achieve full capability to program the current pulse Shared switch asymmetric half-bridge In an attempt to reduce the number of active devices per phase, continued research have generated converter topologies with reduced number of switches, but performance is often compromised, such as control simplicity and regenerative capability. One such example is the shared switch asymmetric half-bridge [31], where switches and diodes are shared between more than one phases, as shown in Figure 2-9. This is particularly advantageous for high phase numbers, but the cost saving has an impact even for two-phase drives. However the converter loses the ability to control the phase current independently. The pair of motor windings sharing a diode and a switch cannot have positive voltage loop in one winding and negative voltage loop in the other winding simultaneously. 27

45 Chapter 2 Power electronics Figure 2-9: Shared switch asymmetric half-bridge converter 2.3 Bipolar converters Full-bridge and half-bridge converters The half-bridge converter, shown in Figure 2-10, is an extension of single phase asymmetric half-bridge. Twice as many switches and diodes are used compared to an asymmetric half-bridge, which makes complete control possible. This arrangement is capable of positive and negative voltage loops in both positive and negative current directions in the phase winding. The half-bridge converter is therefore a bipolar converter as opposed to the asymmetric half-bridge which is only a unipolar converter. 28

46 Chapter 2 Power electronics C1 S2 D2 C2 Figure 2-10: Half-bridge converter Figure 2-11 shows the bipolar operation of the half-bridge converter with a split supply. Positive and negative voltage can be applied to the phase windings in both current directions. 02 D2 02 D1 D1 D1 Figure 2-11: Bipolar operation of half-bridge converter The half-bridge converter is the basic building block for most bipolar converters. Two half-bridges can be connected to form a single-phase full-bridge converter, as shown in Figure 2-12, to achieve a bipolar drive as in the half-bridge converter but using a single rail supply instead of a double rail supply.

47 Chapter 2 Power electronics S2 D2 S4 D4 D1 D3 Figure 2-12: Full-bridge converter As with the shared switch asymmetric half-bridge circuit in the unipolar drives, the shared switch arrangement can be extended to bipolar drives, requiring six switches to control two phases [32]. Figure 2-13 shows such an arrangement. S2 D2 S4 D4 S6 D6 D1 S3 D3 S5 D5 Figure 2-13: Shared switch full-bridge converter [32] Three-phase inverter The principle of single-phase full-bridge inverter may be applied to any number of phases. Using half-bridge module as a building block, a standard three-phase full- bridge inverter can be constructed, as show in Figure

48 Chapter 2 Power electronics S2 D2 S4 D4 S6 D7 VDC D1 S3 D3 S5 D5 3 phase load Figure 2-14: Three-phase full-bridge inverter The switching devices can then be controlled to apply positive, zero and negative voltage loops in both current directions to the three-phase windings Bipolar bifilar converter The basic unipolar bifilar converter can be upgraded to a bipolar topology by adding another switch and diode as shown in Figure

49 Chapter 2 Power electronics D1 S 2 D2 (a) Figure 2-15: Upgraded bifilar converter with bipolar capability In the unipolar bifilar converter, one winding is used for magnetisation and the other for demagnetisation. For bipolar version, both windings are used for magnetisation and demagnetisation, allowing current to flow in both positive and negative directions. The improved topology allows bipolar operation while retaining the simplicity of a bifilar converter requiring only ground-reference switches. 2.4 Power semiconductors The semiconductor switches in power converters are operating in a switching mode. When operating in switch mode, the switches are in one of the three states: fully on - rated at maximum circuits current and relatively small forward voltage drop, fully off - rated at maximum circuit voltage and virtually zero current; or in the transition from one to the other. The time spent in transition between these modes is where both voltage and current are non-zero and must therefore be minimised in the interests of efficiency and thermal operating limits. The ideal semiconductor switch would have the following characteristics: 32

50 Chapter 2 Power electronics 1. Zero resistance or forward voltage drop in on state 2. Infinite resistance in off state 3. Switch with infinite speed 4. Would not require any input power to make it switch Such an ideal device does not exist in reality, a more realistic characteristics of existing solid-state switch technologies would be: 1. Low conduction loss 2. Fast switching speed 3. Simple gate drive requirements Only gate controlled devices MOSFET and IGBT are considered in this thesis, due to their fast switching speed and the simplicity of their gate drive circuits as they only require pulses of gate current to effect switching action. Bipolar transistors require base current to effect the switching. The base current must be maintained during the time the device is required to conduct which makes the control of these devices difficult when high powers are involved. The Metal-Oxide-Semiconductor Field-Effect Transistor (MOSFET) is a voltagecontrolled switch. The current flowing through the MOSFET is controlled by an electric field rather then by a (base) current as in the bipolar transistor. This gives the MOSFET a very high input resistance and thus requires very small input currents. The insulated gate bipolar transistor (IGBT) combines the positive attributes of bipolar transistors and MOSFETs. Like the bipolar transistors, IGBTs have lower conduction losses in the on-state, especially in devices with larger blocking voltages. Similar to the MOSFETs, IGBTs have high impedance gates and only require pulses of gate current to effect switching action. However the IGBTs have longer switching times, especially at turn off while MOSFETs can be turned on and off much faster. The MOSFETs and IGBTs are simpler to drive because they are transconductance devices and can remain fully on by keeping the gate voltage above a certain threshold. Their control electrode is isolated from the current conducting silicon; a voltage must 33

51 Chapter 2 Power electronics be applied between the gate and source terminals to produce a flow of current in the drain. As the gate is isolated electrically from the source and therefore, in theory, no current flows into the gate when a dc voltage is applied to it. As a result, MOSFET and IGBT technologies use much simpler and more efficient drive circuits with significant benefits compared to bipolar devices. 2.5 Gate driver design Power MOSFETs and IGBTs are turned on by charging gate capacitance and turned off by discharging gate capacitance. For rapid device switching, this gate voltage must be charged rapidly against the capacitance of the gate. The gate drivers have to be capable of sourcing and sinking sufficient current to provide fast insertion and extraction of the controlling charge. The high side (floating devices) are devices which do not have their sources or emitters connected to the power supply ground but to the low side devices as shown in Figure device device - 0 Figure 2-16: High side device Depending on the state of the low side devices, the source of the high side device swings between the two rails. The high side devices are therefore not grounded but connected to a floating point; hence the term floating devices. 34

52 Chapter 2 Power electronics The floating devices are complicated to drive, as the gate drive signal must have the following characteristics [33]: To keep the device in conduction the gate voltage must be 10 to 15V higher than the source voltage. Being a high side switch, such gate voltage would have to be higher than the rail voltage, which is frequently the highest voltage available in the system The gate voltage must be controllable from the logic, which is normally referenced to ground. Thus, the control signal has to be level-shifted to the emitter/source of the high side power device, which, in most applications, swings between the two rails Isolated gate drivers Floating gate drive supply with opto-isolator The high side devices require level shifting of the control signals as well as an isolated power supply. Since driving the gates require a significant amount of energy transfer, transformers are usually used to provide non ground-referenced power in an isolated gate driver while opto-isolators are used as a means o f signal transfer. Figure 2-17 shows an implementation of such a system, using an isolated floating power and an opto-isolator for the gate control signal. The cost of such a system is quite high as opto-isolators tend to be relatively expensive and the cost impact of the isolated supply is not insignificant. However this type of driver is capable of applying continuous control voltage to the gate of the power devices. 35

53 Chapter 2 Power electronics Floating Power Supply Gate driver G ate control signal Opto-isolator Low side device Figure 2-17: Opto-isolator with floating gate drive supply As each high side switch requires one isolated power supply, one high frequency switch mode power supply which has several isolated secondary windings can be used to provide multiple non ground-referenced supplies in order to alleviate such a cost impact. The isolation between all the secondary windings then becomes an important design criteria. Pulse transformer A pulse transformer can be used to provide electrical isolation and floating drive signals. Transformer coupling of low level signals to power switches offers several advantages such as impedance matching, dc isolation and either step up or step down capability [34]. They can have the additional advantage o f providing a negative bias. 36

54 Chapter 2 Power electronics G ate control signal Pulse Conditioner Pulse Transformer device - 0 Figure 2-18: Pulse transformer gate driver The pulse transformer gate driver would be implemented as shown in Figure As transformer can deliver only AC signals since the core flux must be reset each half cycle, the maintenance of a continuous gate voltage requires repetitive firing of gate pulse. Fortunately most modem day power converters are controlled using microcontroller and the generation of multiple gate pulse in microcontroller is quite straight forward. Otherwise, a gate pulse generator block which converts long gate pulses into multiple short pulses could be used On the secondary of the gate pulse transformer, a pulse conditioner has to be used to convert the short pulses back the original levels of the gate drive pulse. The pulse transformer also provides a simple way to generate negative bias for the gate at turn-off and during the off state of the switch to improve the turn-off speed and the dv/dt immunity of the switching devices. The availability of the negative biasing enhances the reliability of the gate drive as it increases the gate drive noise immunity by preventing the device from being spuriously turn on by the noise generated in the circuit. 37

55 Chapter 2 Power electronics Non isolated gate drivers The input commands for the high side channel have to be level-shifted from the logic level to the source of the high side device, which can be as high as the rail voltage. Level shifting transistor circuits are now to be found in many new driver circuits such as the International Rectifier s IR2110 and its derivatives [35, 36]. However, the power for the high side has still to be delivered by other means. Bootstrap A non-isolated gate driver requires a supply voltage of 15V more than the dc link voltage. One popular approach to obtain a non-isolated supply voltage of 15V more than the dc link voltage is to use a bootstrapping technique, where a capacitor and diode form a charge-pumping circuit as shown in Figure 2-19 [35, 36]. Gate drive High side device Level shifter Low side device Figure 2-19: Bootstrap circuit Although this method is simple and highly cost effective, duty cycle and on-times are both constrained by the need to refresh the bootstrap capacitor. The gate charge for the high side switches is provided by the bootstrap capacitor which is charged by the 15V supply through the bootstrap diode during the time when the device is off (assuming that the source of the high side device swings to ground during that time, as it does in most applications). 38

56 Chapter 2 Power electronics Charge pump Another means used to generate an over-rail voltage is charge pumping and is sometimes used as a secondary energy source by some driver circuits. This method is more costly and is done by providing an auxiliary low power charge pumping system which oscillates continually [33], 2.6 Summary Although there are many different converter topologies that can be used to build a motor drive, the choice of the best combination of converter topology and motor design can only be made when the exact performance requirements of the system are known, as well as the different priorities including cost, efficiency and controllability. Motor drives for domestic appliance applications tend to be more influenced by the need to be low cost [2]. Bifilar converters are very attractive, since the topology not only achieves unipolar and bipolar current control using the minimum number of power devices in each category, but also does so using only ground-referenced power devices. These two features are very desirable in realising a low cost solution: The fewer power switching devices, the lower the cost, not only in terms of device cost, but also in terms of the gate drives, protection, heat sinks and snubbing requirement associated with the switching devices. Power converters that use only ground-referenced switches are attractive because the simple driving requirements of not needing level-shifting and isolated power supplies. This reduces the complexity and cost o f the converter. The bifilar converter is therefore of great potential to achieve a truly low cost drive and will be investigated in detail in Chapter 3. 39

57 Chapter 3 Bifilar converters Chapter 3 Bifilar converter 3.1 Operation and construction The bifilar converter uses the transformer action between the two closely coupled motor winding to achieve the magnetisation and demagnetisation functions of the power converter. As a result, the converter and the motor windings cannot be considered as separate items, but rather as a single unit. A true bifilar winding consists of two windings with the wire of each winding alongside the other, matching turn for turn and then placed into the stator slot. Alternatively, the windings can be wound separately and then placed into the stator slot, this method is called double wound. Bifilar winding would however provide the best magnetic coupling and therefore minimise the leakage inductance, which is highly desirable but the isolation between windings is reduced. Figure 3-la shows the bipolar bifilar converter and Figure 3-lb shows the bifilar winding being represented by the transformer equivalent circuit of magnetising and leakage inductances. Lm Lm jmotor Winding Motor Winding VDC-T V D C.- t- D2 D1 S2 S 2 D2 Figure 3-1: Bifilar converter and its transformer equivalent circuit 40

58 Chapter 3 Bifilar converters In its simplest form, bifilar converter has only one switch and one diode per phase. The main winding is connected in series with a switch while a diode is connected in series to the secondary winding. The dot convention is used to indicate the relative polarity between the two windings. When the voltage at the dotted terminal on one winding is positive, the induced voltage at the dotted terminal on the other winding is also positive. The power supply builds up the current in the motor winding when the switch is on. The magnetic flux generated by the current in the main winding couples to the secondary winding, inducing a positive voltage at the dotted terminal which reverse biases the diode in the secondary winding. When the switch is off during commutation, current in the main winding decays. The decay of the magnetic field in the main winding induces a negative voltage at the dotted terminal of the secondary winding. This forward biases the diode in the secondary winding and allows current to flow back into the power supply. The secondary winding is therefore used for demagnetisation to recover the commutation energy from the off-going phase back to the power supply. The bifilar converter is therefore one of the simplest and lowest cost converters, employing bifilar phase windings and ground-reference power switch with associated freewheel diode. The drive is greatly simplified as it only requires a standard pulse width modulation waveform and its complement. Furthermore, the drive allows open loop control without level-shifting or current feedback. This is in contrast with many other reduced component single switch topologies which suffer in terms of performance or control complexities [37]. The performance between monofilar and bifilar wound switched reluctance motor was investigated [27]. It was concluded that monofilar wound motors outperform their bifilar counterpart. However, the bifilar drive was considered optimal because it minimised drive cost. Furthermore, the choice of best combination of converter topology and motor design can only be made when the target application of the system is known. The use of non-standard motor windings is also not an obstacle, and the simplified power electronics and control is a positive advantage [38, 39]. 41

59 Chapter 3 Bifilar converters Upgraded bifilar converters with bipolar capability have been proposed for low cost three-phase induction motors because the ground-reference power switches would only require the simpler and cheaper low side gate drivers [40]. Using simple transformer theory, the secondary winding in Figure 3-lb can be referred to the primary, resulting in the circuit shown in Figure 3-2. S2 D2 Lm LL D1 Figure 3-2: Bipolar bifilar converter with secondary reflected through the transformer It can be seen by inspection that this arrangement is very similar to the half-bridge circuit shown in Chapter 2, but with the leakage inductance between the switches. The existence of this inductance limits the inherent shoot through current if the two switches in the same switching leg are turned on simultaneously. This fault tolerant feature is highly desirable for bipolar drives and has resulted in bifilar windings being proposed and investigated for induction motors to reduce the cost and improve the reliability of PWM inverter drives [41, 42]. 42

60 Chapter 3 Bifilar converters 3.2 Practical considerations There are several drawbacks to the bifilar converter. Firstly, only one winding is carrying current at any one time, reducing copper utilisation. So for a given stator slot, the effective winding resistance is increased as the additional secondary windings reduce the copper area available to the main winding. Secondly, the transformer action between the two windings leads to voltage across a winding being reflected across the opposite winding. This means that when one switch is on, the other is required to block the sum of the supply voltage and the reflected voltage which results in at least twice the supply voltage being seen by the other device. The utilisation of the switches is therefore reduced due to the higher voltage rating required. Thirdly, the imperfect coupling between the two windings results in large voltage spikes when the switch is turned off, so a safe path for the energy stored in the leakage inductance must be provided. An increase in the cost of the motor for a given output power at a given efficiency must be balanced against the reduced cost of power electronics. This compromise is central to the use of a bifilar converter drive. 3.3 Winding design Reducing the leakage inductance means intermingling the windings. Two extremes of winding arrangement exist: a true split double winding; a bifilar winding. A split double winding is made by winding the coils as normal but with the wire thin enough to allow room for two separate sets of coils in each slot. This facilitates insulation between the windings but increases the leakage inductance. 43

61 Chapter 3 Bifilar converters If both coils are wound simultaneously from two strands of wire, it is said to be bifilar wound. The two coils are intimately mixed and have very good magnetic coupling, and hence low leakage inductance. The choice between true split winding and bifilar winding depends on the requirements of the target applications. If isolation between the two windings is of utmost importance, as in the case of dual voltage drives presented in Chapter 4, then a split wound configuration is preferred. If the purpose of the bifilar winding is to drive motors that require bipolar excitation, while insulation between the windings and control circuits is still important, insulation between the two coupled coils can be relaxed. In this case, bifilar wound coils are highly desirable in order to keep the leakage inductance to a minimum. 3.4 Snubber requirements Whether a bifilar winding is split double wound or bifilar wound, the magnetic coupling between bifilar windings is never perfect as the open slots in the stator and the end winding allow flux paths which do not cross both windings. This leads to the existence of leakage inductance and leakage energy associated with it. During the demagnetisation of a bifilar wound phase winding, the energy stored in the leakage inductance generates a large back-emf, which if not dealt with properly, will lead to detrimental over voltages across the switching devices. Any proposed converter therefore has to deal with the leakage energy by allowing a safe path for the energy stored in the leakage inductance. An RCD turn-off snubber can be applied to the bifilar windings with two diodes feeding a common resistor and capacitor, as shown in Figure 3-3 [43]. The conventional RCD snubber is commonly used because it is simple and uses relatively cheap passive components. The dissipation in the RCD snubber is however relatively high [44, 45], possibly amounting to 5 to 10% of the rated motor power in a drive with a split double wound windings. 44

62 Chapter 3 Bifilar converters < ~ i - > T -< <!! -< S ' v i ^ R, >, c k < ^ Ll «D1 D2 S1 S 2 Figure 3-3: Bifilar converter with RCD snubber 3.5 Operation of an RCD snubber The leakage inductance associated with a bifilar phase winding creates a voltage overshoot seen by the switching devices when they are switched off. Since this voltage is always at least twice the supply voltage, any excess must be carefully controlled. The energy El associated with the leakage inductance at the point of turnoff is given by the equation EL = '/2 LLi2 (J), Eq. 3-1 where Ll = leakage inductance of the bifilar winding, i = current in the winding at switch off. If the switching device is a MOSFET this energy causes the voltage across the device to rise until avalanche breakdown occurs, dissipating the energy in the device [39], This is acceptable up to the thermal capability of the MOSFET. However, in higher power drives, IGBTs would be used which cannot tolerate a voltage overshoot in excess of their rated voltage. The energy must therefore be absorbed in a snubber capacitor, C, as given by 45

63 Chapter 3 Bifilar converters Vz C (Vmax2 Vmin2) = '/2 LLi2 (J), Eq. 3-2 where Vmin = initial voltage across the snubber capacitor, and Vmax = voltage across the snubber capacitor after receiving leakage energy. In practice Vmax will exceed the value given in Eq. 3-2 because the finite switching times of the devices causing a small additional energy which would otherwise have coupled to the other coil to flow into the snubber capacitor. The choice of circuit components must ensure that the snubber capacitor voltage does not drop below the dc link voltage, V d c, since this causes a needless recharging of the snubber capacitor through the coupled bifilar winding. Hence V mjn should be at least equal to V d c. The peak voltage seen by the switches is V d c + V max and so the component choice for the snubber must ensure that this total switch voltage remains within the rating of the devices. Figure 3-4 shows the principal operating modes of the conventional RCD snubber. The senses of the voltages shown in (a), V d c, V a, V b, V c, V La and V Lb, are applied consistently throughout Figure 3-4. Some intermediate modes are not shown. 46

64 Chapter 3 Bifilar converters VA VC C VDC D2 S2 (a) S1 on, current builds up in coil A, snubber capacitor discharges into resistor C D2 (b) S1 turn off, current in coil A flows into snubber, current in coil B builds up in reverse direction C D2 (c) curent in coil A reaches zero, current continue to flow in coil B Figure 3-4: Principal operating modes of a bifilar wound single-phase inverter with a RCD snubber 47

65 Chapter 3 Bifilar converters The voltage drops across the semiconductor devices are assumed to be negligible throughout the analysis of the circuits in this chapter. In (a), SI is turned on and coil A of the bifilar winding conducts in path (1) and the phase winding flux linkage increases. The voltage in loop (1) is given by Vdc = Va + Vla Eq. 3-3 At the same time any residual voltage on the snubber capacitor discharges into the resistor, dissipating energy in the process. Vc(t) = i2 R, Eq. 3-4 Vc(t) = vmax- ± 'fj! ft E(5-3-5 cc r 1 where ti = time when the capacitor voltage was at Vmax. The capacitor voltage decays exponentially during mode (a). Since coils A and B are coupled the voltage Vb = Va which is approximately the dc link voltage. The capacitor voltage in mode (a) is therefore prevented from decaying below VDc by the coupled voltage in coil B (shown by the dashed path (3) in fig. 2(a)). The voltage at P2, the voltage seen by S2, is clamped by the dc link voltage and the voltage across coil B, thereby giving a total voltage of no less than 2 Vdc- In (b), SI is turned off and current in coil A freewheels into the snubber capacitor. The voltage at PI will rise as the snubber capacitor charges while the voltage at P2 falls due to the voltage reflected across the bifilar winding. When the voltage at P2 has reached zero, the anti-parallel diode of S2 is forward biased and the energy associated with coil A can transfer to coil B. The bulk of the winding energy is returned to the dc link, reducing the phase current to zero. At the same time current in coil A continues to flow into the snubber, charging the capacitor to Vmax. The voltage 48

66 Chapter 3 Bifilar converters at PI, the voltage seen by SI, is the dc link voltage and the voltage across coil A, giving a total voltage of V d c + Vmax. The voltages seen by the switches SI and S2, are therefore a maximum of VDC + Vmax during tum-off but clamped to a voltage no less than 2 Vdc during a positive voltage loop. The governing equations during this period are given below: Vc = -VA - VLA Eq. 3-6 Vdc= -Vb + Vlb Eq. 3-7 In (c), leakage energy associated with coil A has been reset, current stops flowing in coil A and the snubber capacitor starts to discharge into the resistor. The bulk of the winding energy that couples across the bifilar windings continues to force current through coil B back to the dc link. Figure 3-5 is the experimental plot at turn off point of switch SI and shows the rise and fall of voltages at PI and P2 followed by the current transfer across the bifilar windings from coil A to coil B. T3K jazoa 17 *cqs 7-t EE pi 4 1 "ch l 2700 M l ch AG 20.Cl)iS C h3 V 250 V Ch4 250 V T race 1 - current in coil B (2 A /d iv ), trace 2 - current in co il A (2 A /d iv ), trace 3 - current in coil A (2 5 0 V /d iv ), trace 4 - current in coil A (2 5 0 V /d iv ) Figure 3-5: Experimental plots of RCD snubber at turn off of the switching device 49

67 Chapter 3 Bifilar converters A similar process occurs when coil B is magnetised and demagnetised. The value of resistor, R, is determined by the time constant of the RCD snubber such that the snubber capacitor discharges from Vmax to Vmin at the motor commutation frequency where maximum output power is delivered. The RCD snubber is designed to operate at the maximum motor commutation frequency where, in a typical application of a fan loaded system, maximum power is delivered. Running the motor at lower speed means that the snubber capacitor has a longer time to discharge and could therefore reach V d c, resulting in additional losses as the bifilar windings would recharge the snubber voltage to V d c - The problem is further compounded if the motor is running at lower torque and speed, the lower current at lower output power would result in less leakage energy to be captured in the snubber, which would therefore charge to a lower Vmax and have a longer time to discharge. It is therefore difficult to optimise the design of this circuit for a wide range of motor speeds. Furthermore, if PWM is required at low speed, the design is further compromised. This is because during PWM, many more commutations are required per unit time than in single pulse mode. As current transfer is involved with each commutation, the RCD snubber is required to capture and dissipate the leakage energy at a much higher rate, resulting in the increase of the dissipation rate of the snubber. 3.6 Summary RCD snubber is commonly used in bifilar converters because it is simple and uses cheap passive components. However with the increase use of switched mode power converters and increase power levels handled by these converters, it soon became 50

68 Chapter 3 Bifilar converters clear that technology had to be developed to recover the energy stored in the inductive and capacitive snubbers and return it to the converter supply system [46]. The added complexity to the converter has unfortunately been a serious deterrent for the industry to accept this up to the present, so most power converters at present use either no snubbers (just clamps) or dissipative snubbers. But the availability of the low cost microprocessors is beginning to change this trend [24]. The choice of the snubber options would depend on the priorities between cost and efficiency for each application. RCD snubbers would still be preferable if low cost is the most important factor, but if system efficiency is of higher priority, C dump circuits can be used to recover the energy stored in the snubber capacitors [23, 25, 26]. Alternatively, converters with inherent freewheel path, whereby the leakage energy is recovered back into the power source, can be used. These converters normally use diodes to provide a safe path for the leakage inductance energy to be returned to the supply, so no snubber is required. 51

69 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Chapter 4 Bifilar converter for dual voltage applications - Part Introduction The bifilar converters have their advantages and disadvantages; these were discussed in detail in Chapter 3. The compromise between cost and performance is central to the use of bifilar converters. Torrey has concluded that monofilar drives outperform their bifilar counterpart but bifilar drives were still preferred for its cost despite its underperformance [27], The compromise between cost and performance is unique to each target application and some application might benefit from using a bifilar converter. The improved bifilar converter with the addition of one switch and one diode allows: 1. Unipolar currents to be derived from two separate power supplies, where each winding of the coupled windings is connected to two separate power supplies. One set of windings can be connected to the battery supply and used to drive the motor from battery supply. The other set of windings can be connected to the mains supply. 2. The bifilar converter to be bipolar and can control the current flow in both directions, this improved bipolar capability enables bifilar converter to drive motors that require bipolar excitation such as bifilar wound ac or brushless dc motors whilst still retaining its simple electronics and control. The low cost, simple bipolar flux derivation and the capability for dual voltage application will ensure the attractiveness of bifilar converters for many applications. Chapters 4 and 5 present the bifilar converters developed for the operation of dual voltage applications. 52

70 Chapter 4 Bifilar converters for dual voltage applications - Part Dual voltage application A dual voltage powered drive is a drive that can be driven from a dual power source of differing voltages. The two power sources, usually one is mains and the other a battery, can both be used to power the motor. Present domestic appliances are usually either battery powered or mains powered. A few, often low power, exceptions exist which run from both (e.g. electric shavers). Battery powered motor driven systems are increasing in numbers and now range from cordless tools to electric scooters and cars. Battery powered motor driven systems are attractive due to the portability of such systems but are however often restricted by the capacity of the battery packs. These battery powered drives in the market are mostly supplied with a separate battery charger. Such a charger would normally include a transformer operating from an ac supply to step down the voltage, and therefore be bulky and heavy as transformer is a relatively large electromagnetic device. Recent chargers utilise switched mode power supply technology and are therefore smaller but are however more expensive. The use of a separate charger limits the portability of such a system as the system has to be constantly accompanied by a separate battery-recharging device. If extended operating duration is required, multiple battery packs (one charging while the second is in use) or a long recharge time are required. The user would normally be required to remove the battery from the system for the battery to be recharged by the external charger. The portability and convenience of the system would be improved if the battery charging capability were integrated into the system. The battery-powered device could be recharged by just connecting it to the mains without requiring it to be removed from the system to be charged by an external charger [47, 48]. There are many applications that will even benefit from a genuine dual voltage operation, where the drive could be operated from both mains and battery. The device would have the portability of a battery-powered drive, but if extended operation is required, the user can operate the drive from mains, ensuring uninterrupted use. 53

71 Chapter 4 Bifilar converters for dual voltage applications - Part Existing dual voltage powered drives Most of the existing dual voltage drives are essentially battery-powered drives with a front-end to rectify and step down the mains voltage to match the battery voltage. This rectified and stepped down voltage will then be used to supply the necessary power required to drive the battery powered motor and charge the battery. The front-end used could be a mains frequency transformer to step down the voltage and a diode bridge rectifier to rectify it into dc. This solution is reasonably inexpensive but is limited to low power applications since the transformer will get heavy and bulky for powers in excess of several tens of watts. Other solutions include using a diode bridge rectifier to rectify mains into dc before using a dc chopper to step down the voltage. In these solutions, the apparent dual voltage drive is essentially a battery-powered drive with a front-end mains rectifier and voltage transformer. These two components of the existing dual voltage drive are essentially separate components without any integration between the two. All the power required by motoring and charging has to flow through the front-end rectifier and transformer. For higher power levels, this would involve using significantly sized transformer core or expensive dc:dc converter components. Most of these circuits rectify and step the mains voltage down to battery voltage to charge and supply power to the battery powered drive. These circuits are not intended for mains ac supply motoring and the operating duration of these drives are therefore limited by the battery capacity Switched reluctance dual voltage powered drive To be able to run from mains and battery, a genuine dual voltage drive will essentially consists of two separate power converters, one for mains and one for battery. Using 54

72 Chapter 4 Bifilar converters for dual voltage applications - Part 1 converter topologies with the minimum number of switches is therefore of utmost importance to minimise the cost of the converters. The switched reluctance motor, with its unipolar excitation requirement and hence simpler converter compared to a bipolar drive, has the most promising potential for satisfying a low cost solution for a dual voltage drive. The use of coupled switched reluctance motor windings within a battery charging circuit has previously been proposed [49, 50]. This was beneficial as the motor winding was used as an energy transfer device, functioning as a transformer and replacing the transformer which is the single most expensive component in the existing dual voltage drive. A number of such conventional solutions with their corresponding advantages and disadvantages have been reviewed [49], The bifilar converter is suitable for a dual voltage powered switched reluctance drive because the flux required for torque production can be derived from two separate voltage sources, each connected to opposite sides of the bifilar winding. Current in either winding will produce the same polarity of torque in the switched reluctance motor, thereby allowing the motor to be powered from rectified mains or battery voltage. To achieve a dual voltage drive, the motor is oversized to accommodate both sets of windings which are rated to deliver the power required without the other one being present. The additional cost of motors and electronics can be compensated by the added functionality and capability o f the resultant drive. The bifilar winding also acts as an isolation transformer, electrical isolation is provided between the battery terminals and the mains ac supply, ensuring the safety of the system. A tums-ratio can be used to match the voltage difference between the two voltage sources and the existing power electronics and controls that are used for motoring would perform the additional function of an electronic ac/dc converter to achieve battery charging from the mains ac supply. The dual voltage drives can be configured in various topologies depending on cost and application requirements. With one of the coupled windings connected to the mains and the other connected to the battery, MMF that is derived from the mains windings can either be used to transfer power to the battery, producing a battery 55

73 Chapter 4 Bifilar converters for dual voltage applications - Part 1 powered drive with integral charging capability, and/or to generate motoring torque, producing a dual voltage powered drive. 4.3 The dual voltage lawnmower application A target application that would benefit from genuine dual voltage application would be a lawnmower. A lawnmower can be operated for extended duration from mains supply but would have the benefit of being truly portable when operated from the battery. The portability enables the lawnmower to be used where mains leads do not reach without additional extension leads. The lawnmower application has therefore been selected for further investigation. A project had been set up in conjunction with Black & Decker and the following objectives are set out for the lawnmower drive system: 1. Operation as a cordless lawnmower; 2. Operation as a mains powered lawnmower with no batteries present; 3. Operation as a battery charger when mains and batteries are present (both while rotating and not rotating); 4. Complete electrical isolation of the battery terminals from the ac supply. The target specification is summarised in Table

74 Chapter 4 Bifilar converters for dual voltage applications - Part 1 13" blade Typical cutting conditions Speed r/min Torque 0.50 Nm Power W Full load cutting conditions Speed r/min Torque 1.35 Nm Power W 13" blade Starting torque 2.6 Nm Acceleration time 0.50 sec Blade Inertia kg m2 no load speed r/min Acc.n (rad/s/s) rad/s/s Accn torque 2.83 Nm Estimated power requirements Output power W Target efficiency 0.70 % Input power W Open circuit volts V Battery Resistance 0.08 Q Battery current A Terminal volts V Table 4-1: Target specification for motor design 4.4 Bifilar converters for dual voltage applications Converters for bifilar windings have to deal with the leakage energy by allowing a safe path for the energy stored in the leakage inductance associated with the bifilar windings. This chapter presents converters where the safe paths are provided by capacitors used to capture the leakage energy while converters with inherent freewheel path are presented in Chapter 5. The energy transfer between the bifilar windings is central to the operation of the dual voltage drive. There are two modes of energy transfer across the coupled windings, forward and flyback conversions. Forward conversion occurs when current flows in the secondary winding at the start of the motor stroke while flyback conversion occurs when current flows in the secondary winding during phase commutation. 57

75 Chapter 4 Bifilar converters for dual voltage applications - Part 1 It should be noted that when current is flowing in both the primary and secondary windings during the forward conversion mode, the two MMFs are in opposition and any torque production in the motor is caused by the magnetising current in the primary winding. Forward conversion should therefore not be used as the main mode of energy transfer between the coupled windings and the bulk of energy transfer between the coupled windings should occur during the flyback mode. For converters that use capacitors to capture the leakage energy, the bifilar windings can be connected such that the two routes of energy transfer: into the dc link capacitors and the capture capacitors, occur through forward or flyback conversion modes, as shown in Figure 4-1: capture capacitor 1 capture capacitor 2 capture capacitor 1 capture capacitor 2 z z.d#jl li li l}: dc link capacitr r 1 link capacitor 2: dc link capacitt r 1 link capacitor 2 _ (a) Forward conversion into the dc link capacitor, flyback conversion into the capture capacitor (b) Flyback conversion into the dc link capacitor, forward conversion into the capture capacitor Figure 4-1: Connection sense of bifilar windings As the dc link capacitor voltages are fixed by the voltage of the power sources, the transfer of energy into the dc link capacitor is determined by the tums-ratio of the bifilar windings. On the other hand, the voltage across the capture capacitors is a design issue, and therefore programmable depending on the method the leakage energy captured in the capture capacitor is dealt with. The ability to set the voltage across the capture capacitor therefore determines and controls the transfer of energy 58

76 Chapter 4 Bifilar converters for dual voltage applications - Part 1 between the coupled windings. The configuration in Figure 4-1(a), with flyback conversion into the capture capacitors, is therefore preferred. In a dual voltage drive, the bifilar windings act as an isolation transformer between the two voltage sources. The proposed converter will therefore have to maintain the full isolation between the two voltage sources. As such, each side of the bifilar windings would have to be treated separately as a converter subcircuit. The converter subcircuits that use capacitors to capture the leakage energy can be further categorised by the way the captured energy is dealt with: 1. Dissipative - RCD snubber 2. Energy-efficient - C dump circuits RCD Snubber The conventional RCD snubber, presented in Chapter 3 and shown again in Figure 4-2, is simple and uses cheap passive components and would be used for the mains powered converter. The dissipation in the RCD snubber is however relatively high but efficiency is sacrificed to achieve a low cost drive. During phase commutation, the leakage energy associated with the bifilar winding is first being captured in a capacitor before being dissipated in a dump resistor. Energy dissipation rather than recovery across the mains windings is a compromise between performance and cost. For mains powered motoring, the lower drive efficiency is outweighed by the simplicity and low cost of an RCD snubber. 59

77 Chapter 4 Bifilar converters for dual voltage applications - Part 1 dump resistor- capture capacitc D 2 V D C dc link capaci o D 1 Figure 4-2: RCD snubber subcircuit C dump circuits Drive efficiency during battery motoring is of paramount importance in order to achieve prolonged operation when running from battery in the absence of mains. Converters that are used for battery side have to be capable of recovering the energy stored in the capture capacitors and returning it to the battery. Circuits based on C dump converters are therefore used for the battery converter. C dump converters are essentially circuits with additional dc:dc energy recovery circuits which consist of a switch, a freewheel diode and an inductor. Energy is first stored in a capacitor, this energy could be the battery phase commutation energy, or the coupled energy transferred from the mains side. A dc chopper is then used to recover the energy from the capacitor to the battery. There will be two variations of the C dump converter presented in this chapter, the difference of which is the way the transfer o f energy captured in the capacitor to the battery is effected: 1. Buck-boost converter 2. Buck converter The circuit diagrams o f the two converter configurations are shown in Figure 4-3: 60

78 Chapter 4 Bifilar converters for dual voltage applications - Part 1 capture capacitor S 2 S 2 capture capacitor D 4 inductor D 2 D 2 D 3 D 3 V1 dc link capaci V1 dc link capaci inductor D1 (a) Buck-boost configuration (b) Buck configuration Figure 4-3: C dump subcircuits The RCD snubber and C dump subcircuits are then combined to form a dual voltage converter, as shown in Figure 4-4. The bifilar windings are connected in the sense shown in Figure 4-1(a). dump resistor^ capture capacitor 1 capture capacitor 2 '&D3_ Coupled phase 1 j ; fc oupled phase 1 V2 dc link capacitt r 1 dc link capacitor 2 p. ' S S1 D1 D2 7S2 Figure 4-4: Dual voltage converter 61

79 Chapter 4 Bifilar converters for dual voltage applications - Part Operation stages of a dual voltage drive with snubbers In subsequent analysis the converter that initiates an increasing MMF from zero in a motor phase winding will be referred as the primary converter. The motor winding to which it is attached will be referred as primary winding. The other winding to which the primary is coupled will be referred to as secondary winding. The converter to which the secondary is attached to is the secondary converter. The switches in the primary converter are closed to initiate a current pulse in the primary winding, whether the mains winding or the battery winding is the primary winding will vary according to the drive operation Stage one The primary switch is closed and current builds up in the primary. Current flows in the secondary winding in a forward conversion mode as a result of close coupling into the secondary dc link via the anti-parallel diodes of the secondary switches. The secondary dc link will charge, increasing the secondary dc link voltage. When the secondary voltage rise above a certain magnitude, which is dictated by the choice of tums-ratio between the two coupled windings, current in the secondary decreases until eventually none flows. This happens when the secondary voltage is larger than voltage reflected across the secondary winding, reverse biasing the anti-parallel diode of the secondary switch. Vdcp + Vsp + Vrp < N (V d c s + Vos + V rs) Eq. 4-1 where V d c p = Primary DC link voltage Vsp = Voltage drop across the primary switch Vrp = Voltage drop across primary winding V d c s = Secondary DC link voltage V d s = Voltage drop across the secondary diode V r s = Voltage drop across secondary winding N = Tums-ratio of the coupled winding 62

80 Chapter 4 Bifilar converters for dual voltage applications - Part Stage two The primary switch is opened and commutation occurs. Some current will continue to flow in the primary until the leakage flux has been reset but the majority of current will flow into the primary capacitor or transfer across to the secondary capacitor depending on the voltages on each snubber capacitor. The second stage is where mains and battery powered motoring are different. When commutating during mains motoring, the secondary capacitor voltage is continually being discharged via the dc chopper to recover the energy into the battery while the primary capacitor is slowly being discharged through the resistor. This results in the voltage across secondary capacitor when referred through the tums-ratio being lower than the voltage across the primary capacitor, resulting in the secondary being the preferred path and most current transferring to the secondary. N (V ccp + Vdp + Vrp) < Vccs + Vos + Vrs Eq. 4-2 where V c c p = Primary capture capacitor voltage V d p = Voltage drop across the primary freewheel diode Vccs = Secondary capture capacitor voltage V ds = Voltage drop across the secondary freewheel diode On the other hand when commutating during battery motoring, current will initially flow in the secondary winding charging the secondary capacitor. The voltage across the secondary capacitor, which discharges into the dump resistor, will rise as it is charged. The current in the secondary will decrease until eventually none flows. At the same time, the voltage across the primary capacitor is continually being discharged to recover the commutation energy to the battery. The high secondary voltage when referred to the primary will ensure that most current continues to flows in the primary. N {V ccp + Vdp + V r p ) > Vccs + Vds + V rs Eq

81 Chapter 4 Bifilar converters for dual voltage applications - Part Stage three Current in both the primary and secondary has fallen to zero, resetting the winding flux. Depending on the choice on tums-ratio of the voltage across both primary and secondary capacitors, most current may transfer to the secondary with leakage current only flowing in the primary, or current only flows in a negative voltage loop into the capture capacitor in the primary. 4.6 Mains RCD snubber, battery C dump with buck-boost configuration Figure 4-5 shows the dual voltage converter with mains RCD snubber and battery C dump with buck-boost configuration [23]. ST 4 = CL CH RH A D4 Mains A DL BATTER' SL1 SL2 SH2 SH1 Figure 4-5: Dual voltage converter with mains RCD and battery C dump buck-boost configuration 64

82 Chapter 4 Bifilar converters for dual voltage applications - Part Discussion of modes of operation M ains powered motoring The operating modes of the converter in mains powered motoring is shown in Figure 4-6. cb t S b T A D 1 D3 2 A D1 D3 A ZSL A L l} l C1 5L1 SI hq t V hq l (a) Positive voltage loop (b) Negative voltage loop cb f ±: CH < RH ib i t : CH < RH A D1 D3 A A D1 D3 A M BATTERY r t C1 2 s l C2l BATTERY i t : C1 Al C2^= SH1 SH1 (c) Energy recovery - transferring energy from C l to Ll (d) Energy recovery - L l freewheeling back to battery Figure 4-6: Mains powered operating modes (motoring and charging) (only one coupled phase is shown) 65

83 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Mains motoring is achieved by the phase magnetisation of the phases by switching Shi and Sh2 according to the rotor position sensors, as shown in Figure 4-6(a). A voltage is induced in the coupled battery winding which would tend to charge the battery through forward conversion. When the battery is uncharged or battery voltage is low, current will flow into the battery. The tums-ratio will be chosen such that this current will decrease as the battery voltage rises and eventually none flows, as determined by Eq At low motor speeds it is usual to limit the current in the motor phase windings by the use of pulse width modulation (PWM) control. This can be achieved by chopping Shi or Sh2 as appropriate. During commutation, energy can either be transferred to the battery side or continue to flow in the primary side depending on the states of the capture capacitor voltages as determined by Eq. 4-2, as shown in Figure 4-6(b). The energy recovery switch, Sj, controls the voltage across Cl such that the energy stored in Cl is continuously being recovered to the battery. If voltage across Cl were lower than voltage across Ch when referred through the tums-ratio, then most of the current would transfer to the battery side. This energy would then be used to charge the battery via the dc chopper Battery powered motoring The operating modes of the converter in battery motoring is shown in Figure 4-7. Battery motoring is achieved by the phase magnetisation of the battery phases by switching SLi and SL2 according to the rotor position sensors, as shown in Figure 4-7(a). A voltage is induced in the mains winding which charges the mains dc link capacitor, according to equation Eq However, once the mains dc link capacitor is charged, negligible current will flow in the mains winding. 66

84 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Mains Mains BATTERY BATTERY' (a) Battery Phase magnetisation (b) Phase demagnetisation Mains Mains BATTERY" (c) Energy recovery - transferring energy from Cl to Ll (d) Energy recovery - Ll freewheeling back to battery Figure 4-7: Battery motoring operating modes (only one coupled phase is shown) During phase magnetisation, current builds up in each of the respective phase windings, drawing energy from the battery. As the rotor of the switched reluctance motor approaches the aligned position the respective phase winding switch is turned off. During the subsequent demagnetisation, shown in Figure 4-7(b), the field energy recovered from the motor is initially transferred to Cl, before subsequently being recovered back to the battery via the energy recovery circuit. The energy recovery switch, St, controls the voltage across Cl such that voltage across Cl are lower than voltage across Ch when referred through the tums-ratio, resulting in the battery winding being the preferred current path and current continues to flow in the battery side into Cl- Some current will initially flow in the mains winding to charge the mains 67

85 Chapter 4 Bifilar converters for dual voltage applications - Part 1 snubber capacitor but quickly decreases to zero when the voltage of the mains snubber capacitor rises above the voltage across Cl when referred through the tums-ratio. The transfers of current between the coupled winding occurs according to Eq PWM control is adopted at low speed to regulate the current in the motor phase windings and this is achieved by chopping Sli or Sl2 as appropriate. Each time Sli or Sl2 is turned off energy will be transferred to Cl- Again the energy recovery switch, St, is controlled in such a way that the battery winding is the preferred current path to avoid energy being transferred to and dissipated in the mains side Phase demagnetisation and energy recovery circuit The energy recovery circuit consists of C l, S t, L l, D l and the battery. During demagnetisation of a phase, when either S l i or S l 2 is switched o ff, the energy stored in the magnetic field associated with the phase winding is transferred to Cl. The transfer of the energy stored in Cl to the battery can be achieved in one of two ways depending on the operation pattern of St. Firstly, the switch St can be turned on for a time equivalent to one quarter of the resonant period of the resonant circuit formed between Cl and Ll. The voltage on Cl will fall to zero and all the energy is transferred to the inductor, L l, in one block. The energy is then passed to the battery while St is off as in Figure 4-7(d). Whilst such a scheme is simple to control, requiring no feedback of the voltage on Cl or the current in L l, it does lead to a requirement for the capacitor and the inductor being large enough to store all the energy in one block. The second control method involves the continuous running of the energy recovery switch as a high frequency power converter. This allows the inductor to be significantly smaller as the energy is returned to the battery over a much longer period of time. A typical control scheme would regulate the energy recovery switch to regulate the voltage on CL. The additional voltage feedback which is required to make such a system work effectively is justified with the savings in inductor and switch rating. 68

86 Chapter 4 Bifilar converters for dual voltage applications - Part Battery charging circuit The battery can be charged without the motor needing to rotate. Energy can be transferred to the battery in two ways. 1. If the battery voltage is low, current will flow directly into the battery, due to forward converter action when the high voltage switch, SHi or Sh2, is on. This is shown in Figure 4-6a. The tums-ratio between the bifilar windings must be chosen to limit this forward conduction path, as described in equation Eq At the end of each forward conduction cycle the energy stored in the magnetic field is recovered through flyback mode into the capacitor, CL, thus avoiding the build up of flux in the motor. 2. When the battery voltage rises, with the appropriate tums-ratio, the higher battery voltage will oppose current flowing directly into it and forward converter action ceases. The battery is then charged through the flyback conversion mode. While switch S h i or S h 2 is on (keeping S l i or S l 2 off), current increases in the mains winding, thus establishing the flux in the core, as shown in Figure 4-6a. When Shi or Sh2 is switched o ff, the decay in magnetic field in the primary winding induces a negative voltage at the dotted terminal of the secondary winding. This forward biases the diode in the secondary winding and allows current to flow into the capture capacitor. Stored magnetic energy is thus transferred to the capacitor C l while the S h i or Sh2 is off, as shown in Figure 4-6b. The energy transferred to Cl can then be recovered through the energy recovery circuit back to the battery by switching energy recovery switch St on, initially transferring the commutation energy to L l, as shown in Figure 4-6c. When S t is then switched o ff, as shown in Figure 4-6d, current in Ll continues to freewheel through diode Dl thereby charging the battery. The battery voltage at which the crossover between these two modes occurs is controlled by the tums-ratio of the coupled windings. 69

87 Chapter 4 Bifilar converters for dual voltage applications - Part Mains RCD snubber, battery C dump with buck configuration Figure 4-8 shows the dual voltage converter with mains RCD snubber and battery C dump with buck configuration [26]. The dc chopper now operates with buck principle to recover the energy back to the battery. A zero voltage loop is now possible. ST CH RH A DL A D1 D2 A A D4 Mains BATTERY _ C1 ± C2 z t SL1 SL2 SH2 SH1 Figure 4-8: Dual voltage converter with mains RCD and battery C dump buck configuration 4.9 Discussion of modes of operation Mains powered motoring The operating modes of the converter in mains powered motoring are shown in Figure

88 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Mains 'BATTERY BATTERY (a) Positive voltage loop (b) Zero voltage loop A DL1 (c) Negative voltage loop Figure 4-9: Mains powered operating modes (motoring and charging) (only one coupled phase is shown) The mains powered operation for buck configuration is similar to the one for buckboost configuration. Mains motoring is achieved by switching S h i and S h 2 according to the rotor position sensors. The main difference is the existence of a zero voltage loop in the buck configuration which aids the PWM current control during low speed. When either S h i or S h 2 is switched off, S t is switched on at the same instance. A zero voltage loop is established across the battery side and would be the preferred current path in the bifilar windings, as shown in Figure 4-9b. The majority of current therefore transfers to the battery side and freewheels through Sy. Only the leakage current that is not coupled continues to flow in the mains side. The zero voltage loop capability alleviates the switching frequency and enables a better current regulation during PWM.

89 Chapter 4 Bifilar converters for dual voltage applications - Part 1 During commutation, the current transfer between mains and battery side is being governed by Eq. 4-3, as shown in Figure 4-9c Battery powered motoring Battery motoring is achieved by the phase magnetisation of the battery phases by switching S l i and S l 2 according to the rotor position sensors as in the buck-boost configuration. The additional zero voltage capability available for the buck-boost configuration is beneficial for PWM current control at low speed. When Sli and Sl2 are switched off, switching Sj on at the same time achieves a zero voltage loop. The current in the winding freewheels through S t, as is shown in Figure 4-10(b). Battery phase demagnetisation is achieved by turning all the switches off, as shown in Figure 4-10(c). The phase demagnetisation is similar to the buck-boost configuration in that the current flows in the respective windings depending on the tums-ratio and the state of the snubber capacitors, according to Eq The energy recovery switch, Sj, is controlled such that the battery phase would be the preferred current path to avoid any current unnecessarily being transferred to and dissipated in the mains snubber. The energy stored in the capacitor CL can then be recovered via the inductor when St is switched on, as shown in Figure 4-1 la. Thus by controlling S t, the voltage across Cl and the charging rate into the battery can be controlled. 72

90 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Mains BATTERY BATTERY (a) Positive voltage loop (b) Zero voltage loop A DL1 (c) Negative voltage loop Figure 4-10: Battery powered operating modes (only one coupled phase is shown) Phase demagnetisation and energy recovery circuit The energy recovery circuit consists of C l, S t, L l, D l i, D l 2 and the battery. During demagnetisation of a phase, when all the battery switches are switched o ff, the energy stored in the magnetic field associated with the phase winding is transferred to C l via the diodes D1 and D l 2- Once captured in C l, the energy is then recovered back to the battery in the following modes. The energy is Cl is first transferred to Ll via the battery, Dj and Dli when St is switched on, as shown in Figure 4-11(a). Figure 4-11(b) shows that when the switched

91 Chapter 4 Bifilar converters for dual voltage applications - Part 1 is subsequently switched off, the current built up in Ll continues to flow into the battery. M ains Mains BATTERY BATTERY (a) Energy recovery - transferring energy from CL to Ll (b) Energy recovery - Ll freewheeling back to battery Figure 4-11: Energy recovery modes Battery charging circuit The charging modes in the buck configuration is similar to the buck-boost configuration in that energy can be transferred to the battery in forward and flyback conversion modes. The battery can be charged without the motor needing to rotate. The tums-ratio between the bifilar windings is chosen such that forward conversion during the mains magnetisation process trickle charges the battery. The majority of the charging is achieved in flyback mode. When the mains winding is demagnetised, the flux decay in the main winding induces a voltage which allows a current to flow in the closely coupled battery winding. This energy is first stored in Cl before being recovered via the buck dc chopper to the battery in the same way as the demagnetisation process of the battery phase. 74

92 Chapter 4 Bifilar converters for dual voltage applications - Part Implementation Motor construction In a lawnmower application, the drive must be able to provide considerable starting torque and two-phase switched reluctance motor is the best choice in terms of performance and cost. Single-phase would suffer from the inability to satisfy the starting torque requirement while three-phase switched reluctance motor would have incurred more expensive power electronics. Two-phase switched reluctance motors are generally constructed with twice as many stator poles as rotor poles (eg. 4-2 or 8-4 configuration). The chosen motor topology is a two-phase, eight-pole stator with a four-pole rotor. With the use of an eight-pole stator, the system can exploit the parallel and series connection of individual windings to achieve a tums-ratio between the bifilar windings. This minimises the difference in number of turns to be wound in each of the battery and mains coils to be coupled, achieving a better coupling and minimising the leakage inductance between the bifilar windings. A motor with stator outside diameter of 89 mm with 60 mm stack length is used. This dimension is constrained by the lawnmower deck to which the motor will be mounted. The rotor is asymmetric in design to facilitate self-starting. The rotor is graded to extend the regions of rising inductance in one phase into that of the next phase to ensure the continual availability of torque to achieve starting in the system [51]. 75

93 Chapter 4 Bifilar converters for dual voltage applications - Part 1 / / ' X R 0,02b rri R m RO Figure 4-12: Stator and rotor laminations W inding design To minimise the leakage inductance, the battery and mains windings should ideally be wound together on each slot with the same number of turns and use the combinations of different series parallel connections between the coils to achieve different total number of turns per winding. For instance a two-phase eight-stator pole switched reluctance motor would have four coils per motor phase winding. If four coils of the battery winding are connected in parallel while four coils of the mains winding are connected in series, a tums-ratio if 4:1 is achieved. If however a tums-ratio of greater than four is required, then battery and mains coils of different number of turns, which would inevitably results in higher leakage, would have to be adopted. As the transformer action between the two closely coupled motor windings is central to the operation of the bifilar converter, the converter and the motor windings cannot be considered as separate items, but rather as a single unit. Therefore the design of the windings has to take into account the converter configuration being used. 76

94 Chapter 4 Bifilar converters for dual voltage applications - Part 1 The main constraint in determining the tums-ratio of the coupled windings would be the forward conversion during mains motoring. The tums-ratio is chosen such that forward conversion during mains motoring will trickle charge a battery, but not overcharge it. The internal resistance of the battery yields different battery voltages during battery charging and discharging. To charge the battery, a higher battery terminal voltage is required to overcome the internal resistance of the battery but during discharge, the internal resistance of the battery causes a voltage drop and reduces the battery terminal voltage. Thus, during mains motoring, for a battery charging voltage of 30V and a rectified mains dc link voltage of 340V, a voltage ratio of 11:33 is obtained. A bifilar winding with a tums-ratio of 11:33 would be required to match the mains and battery voltages. The voltage drops across the devices are neglected since they are small relative to the mains dc link voltage. A tums-ratio of 12 is therefore selected. As the use of parallel and series connection in an eight-pole stator yields a tums-ratio of 4:1 before the use of different numbers of turns in the slots. With a target tums-ratio of 12:1, each coupled coils will have a tums-ratio of 3:1. With a tums-ratio of 12:1, the forward conversion during mains motoring would be 340/12 = 28.3V. This will ensure that forward conversion would only trickle charge a discharged battery up to 28.3V. During battery motoring, for a typical battery discharging voltage of 22V, the forward conversion would charge the mains dc link voltage to 22x12 = 264V. This voltage is well below the mains dc link voltage of 340V. For a coupled winding with tums-ratio of N :l, the relation between the number of turns of each of the windings is given by: mm Eq. 4-4 rrilv where iuhv is the number of turns in the mains winding and mlv is the number of turns in the battery winding 77

95 Chapter 4 Bifilar converters for dual voltage applications - Part 1 With the tums-ratio set at 12:1, the choice of number of turns for the motor windings, m H v and i u l v, was estimated by spreadsheet using the finite element results by considering the change of flux associated with a slot winding as the rotor moves between the positions of pole unalignment and alignment at the rated speed. The optimum number of turns is that which gives a rotational EMF equal to the available supply voltage at the rated speed. More turns than this will produce a larger rotational EMF, preventing full power from being achieved at rated speed, whilst less turns would increase the current handling requirements of the drive, raising the cost. Once the number of turns is determined, the wire diameter to be used depends on manufacturing constraints in achieving a coil-packing factor on the available slot area. Based on previous winding experience and the additional complexity in winding bifilar coils, a more conservative coil packing factors of 0.12 and were estimated for mains and battery winding respectively. The specification for each pole is: Low voltage coil 40 turns 0.90 mm diameter High Voltage coil 120 turns 0.40 mm diameter Table 4-2: Winding specification On each of the eight poles there will be a mixture of low voltage and high voltage coils. The low voltage coils of each phase (four poles) should be connected in parallel to create a N-S-N-S four-pole pattern while the same four-pole pattern will be created by four high voltage coils connected in series. This connection pattern is repeated for the second phase. Both phases contain identical coupled windings to allow two-phase starting from either mains or battery supplies. 78

96 Chapter 4 Bifilar converters for dual voltage applications - Part Com parison between C dump subcircuits The two dual voltage converters, as presented in sections 4.6 and 4.8, were built for the lawnmower application in order for comparison between the two to be made. As the efficiency in battery powered motoring is of paramount importance to ensure the longest cutting capacity with each battery pack. The comparison between the two converters had been concentrated in battery powered motoring mode to determine the effectiveness of the energy recovery circuit in producing an efficient drive to achieve the maximum cutting capacity per battery pack. Both converters are operated from a 22V supply with a load of 0.5Nm at 3500 r/min, the typical cutting condition of the lawnmower. The experimental results of both the circuits are presented below: Buck-Boost Buck P osition sensor signal 5 V /d iv B attery current 2 0 A /d iv Tck HUM SouEsTT 120 Acqs 14 Acqs Cl Freq Hz Low signal am p litu d e B attery v o lta g e lo V /div lo.omv Ml ooms h t I 17 May 1999 P osition sensor signal 5 V /d iv Battery phase current 2 0 A /d iv 36 Acqs Cl Freq Hz Low sig n al a m p litu d e Battery phase v o ltage lo V /div OB- - 79

97 Chapter 4 Bifilar converters for dual voltage applications - Part 1 43 Acqs P osition sensor signal 5 V /d iv C l Freq Hz Low signal am p litu d e Cl Freq Hz Low signal am p litu d e Snubber inductor current 10 A /d iv Snubber capacitor v o lta g e lo V /d iv ES i-s V 17 May 1999 EX Figure 4-13: Experimental waveforms for buck and buck-boost configurations Results Buck-boost Buck Output Power / W Input Power / W Power into motor / W Motor Efficiency / % Overall Efficiency / % Table 4-3: Experimental results for buck and buck-boost configurations From the experimental results, it was observed that the current in the snubber inductor, L l, is lower in the buck configuration but the voltage across the snubber capacitor, C l, is higher in the buck configuration. A trade-off therefore exist between a smaller inductor but higher voltage ratings for the switches in the buck configuration and a larger inductor but lower voltage ratings for the switches in the buck-boost configuration. The buck configuration was selected for further investigation for various reasons: 1. It uses a smaller and hence cheaper inductor. 2. It achieves a continuous battery charging current 80

98 Chapter 4 Bifilar converters for dual voltage applications - Part 1 3. It has zero voltage loop capability which reduces the current ripple during PWM chopping, thus reducing noise and losses associated with the current ripples. 4. With the improved capability and falling prices of modem semiconductors, the higher voltage ratings required for the switches in the buck configuration poses no significant cost disadvantage Power electronic converter The complete power electronic circuit, with mains RCD snubber and battery C dump with buck configuration, is shown in Figure 4-14 together with the component list in Table 4-4. As will be shown in section , the maximum power deliverable during battery motoring was limited by the rating of the MOSFETs. Two MOSFETs were therefore connected in parallel to deliver the maximum powered specified in Table 4-1. RH A DL1 A D1 D2 A A D4 Mains 24V SL1.1 SL1.2 SL2.1 SL1.2 SH2 SH1 Figure 4-14: Circuit diagram o f the power electronic converter 81

99 Chapter 4 Bifilar converters for dual voltage applications - Part 1 No Component Qty Description 1 SH I-2 2 IRGPH30KD 2 SL1.1-SL2.2 4 IRF ST 1 IRF Cl 2 1 OOOjlxF 63V Electrolytic 5 C pF 500V Electrolytic 6 CL jliF 40V Electrolytic 7 CH 1 3.3pF 630V Polypropylene 8 RH 1 3* 10k 17W resistor (in parallel) 9 D l-2 1 BYV42E 10 D3-4 1 RGRH D LI-2 2 BYV79E 12 LL 1 576pH 3C80 core approx 60 turns of 0.914cm Table 4-4: Component list for the power electronic converter Control circuits The complete control circuit developed for the lawnmower drive is shown in Figure The control of the complete system, namely motoring in all modes, charging, braking and monitoring of speed and voltage, is implemented in an 8-bit microcontroller, Microchip s PIC 16F84/10P. The full list of all operating modes is shown in Table

100 Chapter 4 Bifilar converters for dual voltage applications - Part 1 V o r a w i\ n t i t l r! I G A DFEIfiNED HY j TDNR?R7iRA? rr.n o a ; v o l t a g e : c o n t r o l f o r l a w n m o w e r (LON TIROL. C 1 ROLJ ( T REV. ISOuE 2 DATE DEC 1399 DRAWN DY j I. C D irr ITII 3iir r t n o. j i nr i Figure 4-15: Schematic layout of the control circuit diagram 83

101 Chapter 4 Bifilar converters for dual voltage applications - Part 1 When motor is not in motion Mains Present Battery Present Battery fully Handle pulled Operating Modes charged V V V V Mains motoring, battery voltage being monitored at 16 khz, if battery voltage rise above 27V, battery MOSFET is turned on to discharge battery, thereby regulating battery voltage to 27V V V V X Charge battery until 27V, then revert to standby mode (monitoring handle being pulled) V V X S Mains motoring, mains winding commutation energy trickle charging battery V V X X Battery Charging V X N/a Mains motoring, commutation energy charge dc link. Battery MOSFET is turned on when dc link voltage rises above 27V, thereby regulating dc link voltage to 27V. Note: DC link is being protected from over voltage during mains acceleration by a transzorb connected across it. V X N/a X Attempt to charge dc link, detecting full voltage and revert to standby mode (monitoring handle) X V Irrelevant V Battery motoring X S Irrelevant X No power to PIC! No drainage from battery When motor is already running and handle is released, system utilises battery to brake the mower. Braking is not possible if battery is not connected. Table 4-5: Operating modes o f the lawnmower

102 Chapter 4 Bifilar converters for dual voltage applications - Part 1 The inputs to the controller are: On/Off Mains present Battery full Portb,l (active high) Portb,2 (active high) Portb,0 (active high) This signal is dropped down from +15/12V supply voltage and fed through a filter into the microcontroller input This signal is derived from a voltage divider from +15V regulated supply from mains Battery voltage and a reference voltage are fed into a voltage comparator LM 311 and the output is fed into the microcontroller input Opto sensor Portb,4 Schmitt triggered output opto HOA2001 is used for the opto sensor. The sensor output is further filtered across a filter into microcontroller input Table 4-6: Inputs to the microcontroller The outputs from the controller are: HS1 on signal HS2 on signal H S1 off signal Portb,6 Portb,7 Porta,0 Bipolar outputs are required for the transformer isolated IGBTs as they have to be negatively biased when not in use to avoid misfiring HS2 off signal Porta, 1 LSI Porta,3 These signals are fed directly into gate drivers IR2113 and LS2 Porta,2 INT202 H-sw Porta,4 Self-sustain Portb,3 microcontroller supply self-sustain signal Table 4-7: Outputs from the microcontroller 85

103 Chapter 4 Bifilar converters for dual voltage applications - Part Gate drive circuits Battery gate drive As isolation is not required between the top switch and the bottom switch, International Rectifier s IR2110 gate driver, with independent high and low side referenced output channels, was used to drive the MOSFETs with the supply to the top switch driver being generated by a bootstrap capacitor method. The high side channel is used to control the energy recovery switch while the low side channel is used to control one of the low side switches. The second low side switch is being driven by a low side only driver, Power Integration s INT-202. In applications involving batteries, the output voltage is present before the input power is applied to the converter. In these cases, the source of the switch and the negative node of the bootstrap capacitor are sitting at the output voltage and the bootstrap diode is reverse biased at start-up. It is therefore not possible to deliver the initial charge to the bootstrap capacitor. Since there is enough voltage differential between the input and output voltages, from 24V battery to the 15V logic supply, a simple circuit comprise of a resistor, a diode and a zener diode can solve the start-up problem as shown in Figure RSTART DSTART VB DZ OUT VS BATTERY C Figure 4-16: Bootstrap start-up circuit for battery applications 0 86

104 Chapter 4 Bifilar converters for dual voltage applications - Part 1 In this start-up circuit, Dstart serves as a second bootstrap diode used for charging the capacitor at power up. The bootstrap capacitor will then be charged to the zener voltage of Dz, which is chosen to be 15V in the circuit. The charge current of the bootstrap capacitor is limited by the start-up resistor. For best efficiency, Rstart is selected to limit the current to a low value since the second bootstrap path takes over once the circuit is operational. Mains gate drive As isolation is required between the logic control system and the high voltage switches, the system adopts an isolated gate drive implemented using pulse transformer gate driver as described in Chapter 2. The IGBTs are switched on or kept off by maintaining the base voltage at +15V or -15V. The -15V negative biasing is important during battery operation to prevent the IGBT misfiring caused by any dv/dt impressed on the collector emitter of the IGBT coupling back to its base via the bifilar windings. Bipolar drive signals are therefore required to drive the pulse transformers that swing between positive and negative voltages. A bipolar drive is implemented using the Motorola gate drivers MC34151 to achieve a push pull switching circuit driving the primary of the pulse transformer. As transformers can deliver only AC signals, gate pulses have to be refreshed to maintain adequate IGBT gate voltage, which can be easily implemented using the programmable micro controller. The pulses are refreshed at 16 khz by pulsing a +15V to keep the IGBT on or a -15V to switch it off. Short pulses of lps are used to drive the pulse transformer to avoid the transformer saturation. There are two output pins from the microcontroller for each IGBT; a pulse on one gives -15V on the transformer whereas a pulse on the other gives +15 V to the primary of the transformer. Due to leakage effects, gate voltage cannot be maintained indefinitely from just one pulse. Continual switching of the input is required to maintain a good noise margin. 87

105 Chapter 4 Bifilar converters for dual voltage applications - Part 1 The pulse conditioning circuit at the output is then used to hold the gate of the IGBT at +15V or -15 V between the refresh pulse, thus keeping the IGBT on or off. 3V6 ZTX751 2 x UF R gate transform! 3V6 ZTX651 15V 15V 7 \ 68k source Figure 4-17: Gate pulse transformer conditioning circuit The circuit diagram of output conditioning stage is shown in Figure The circuit uses two active transistors, ZTX 751 and ZTX 651, to hold the base of the IGBTs at +15V or -15 V in order to keep the IGBT on or off as required. Zener clamps are provided from each IGBT gate to emitter to avoid damage due to excess gate voltage. When mains is not present and the pulse transformer is not running, a 68k ohm gatesource pull down resistor is used to prevent any dv/dt impressed on the collector emitter of the IGBT coupling back to its gate via the bifilar windings Software design Start-up PWM routine At start-up, an initial PWM duty cycle is adopted to limit the current and hence the snubber voltage. The presence of leakage energy, which has to be dissipated in the mains snubber, has limited the PWM frequency adopted in mains motoring. As a result, mains PWM chopping is implemented at a lower frequency of 2 khz compared to 16 khz for battery motoring. The PWM duty cycle is progressively increased as the speed increases to provide the acceleration torque required while still keeping the current and snubber voltage within safe operating limits. The profile of the PWM ramp for mains and battery motoring

106 Chapter 4 Bifilar converters for dual voltage applications - Part 1 are shown in Figure The start-up routine also includes a 10 ms delay to the first four position sensor pulses to damp oscillations between the position sensor transitions. The inclusion of the delay has proved to be extremely successful in starting the motor. An initial blanking pulse is applied to the corresponding phase at each position sensor transition during the PWM routine. This enables the current to build up rapidly at the beginning of each pulse to a preset level without the penalty of rapid rise in snubber voltage. The current is subsequently regulated by PWM chopping at 16 khz and 2 khz for battery and mains motoring respectively. Duty Cycle for Battery Motoring Duty Cycle for M ains Motoring 2500 Figure 4-18: PWM duty cycles Single pulse routine At speeds above 2500 rpm, the PWM duty cycle reaches 100% and beyond this, speed control is achieved by varying the pulse width of the pulse applied within each 45 degrees rotor angle. A hysteresis of 200 rpm is also implemented to avoid oscillations between single pulse and PWM modes and the system will only re-enter PWM mode when the speed falls below 2300 rpm. The present algorithm also limits the no load speed to just below 4200 rpm by cutting back the pulse width progressively as the speed increases. 89

107 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Load algorithm The load of a lawnmower is of a fan load type and the PWM ramp up is adjusted to provide the required acceleration torque. At present, the motor accelerates from stand still to single pulse mode in under 1.8s. The load algorithm is designed to deliver a target of 0.5 Nm at 3500 rpm. As the load increases further the target is to deliver 1.35 Nm at 2500 rpm. This is the maximum power deliverable and beyond that, the motor will re-enter PWM routine and follow the PWM ramp down. This controllable ramp down to a stop will ensure that at all instances, the current and snubber voltage is maintained within controllable limits. The load algorithm is shown in Figure S' Speed (rpm) Figure 4-19: Load algorithm Braking routine When the user switches off, the system utilises the battery to brake the motor. Plugging is used and additional power drawn from the battery is used to brake the motor. This is achieved by switching on the phase with decreasing inductance to produce negative torque to brake the motor. 90

108 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Charging routine The system charges the battery at a near ultrasonic frequency of 16 khz. Energy is stored in the mains winding by switching the high voltage IGBTs on. This energy is then transferred across the bifilar winding into Cl when the IGBT is switched off. St is then turned on to transfer the stored energy from Cl to Ll and the battery and when St is subsequently turned off, current continues to flow from Ll into the battery. The use of RCD snubber to dissipate the leakage energy associated with the bifilar winding influences the choice of frequency and duty cycle adopted for charging the battery. At present, the use of a near ultrasonic charging frequency of 16 khz is achieved at the expense of a lower duty cycle adopted. The battery is therefore only charged at a rate of approximately 0.5 A. A higher charging rate could only be achieved by reducing the leakage inductance or adopting a lower switching frequency. The battery charging would then be audible. It has been suggested that periods of audible charging could be used as a tone to indicate charging mode. Power supplies In order to fulfill the requirements of a genuine dual voltage drive, the control logic supply has to be derived from both mains and battery as follows: Mains Battery Description Not present Present Control voltage is supplied from the +12V centre tap of the battery, through a transistor and to the system. The transistor is turn on when user switch on, and is maintained on by the microcontroller. When the user switches off, the system enters braking mode, after another delay of 3s, the microcontroller switches the transistor off, thereby cutting the logic supply and preventing battery drainage. 91

109 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Present Not Present Control voltage is supplied directly through a mains step-down transformer, rectifier and +15V regulator. The system initiates charging or motoring depending on the state of the switch. If the user switch is off and battery is fully charge, the system reverts to standby mode. Present Present Control voltage is drawn from the +15V from mains instead of the +12V from the battery. The system initiates charging or motoring depending on the state of the switch. If the user switch is off and battery is fully charge, the system reverts to standby mode. Table 4-8: Power supply requirements The circuit configuration adopted to fulfill the requirements is depicted in Figure The control voltage from the battery is connected via a transistor switch to the lawnmower supply. The transistor will only be turned on when the user turns on or there is a self-sustain signal from microcontroller. The self-sustain signal from the microcontroller is set to high whenever the user switches on. There is a delay applied to this signal to enable it to stay high for a further 3 seconds after the user switches off. This signal enables the system to be powered for another 3 seconds to achieve braking. In all other situation, the battery supply will be cut-off and prevented from being drained. The mains, on the other hand, is directly connected the lawnmower supply to enable charging even when the user switch is off. 92

110 Chapter 4 Bifilar converters for dual voltage applications - Part 1 User on/off +12V from battery +15V from mains transistor PIC self-sustain signal supply to lawnmower controls Figure 4-20: Power supply configuration Voltage regulation routine As a genuine dual voltage drive, the drive has to be operational even when the battery is removed. During mains powered motoring in the absence of the battery, all energy that is transferred across to the battery side has to be stored in the dc link, causing the dc link voltage to rise rapidly. As the control logic voltage is derived from the battery side, the rising voltage will result in failure. The system has to therefore monitor the dc link voltage at ultrasonic frequency and switch on the battery switches whenever the voltage across the dc link rises above the threshold value. In this case, the voltage across the battery side, whether it is a fully charged dc link or a fully charged battery, is regulated to prevent it from overvoltage. By switching the battery switches on, the voltage of the battery side is clamped to the voltage on the mains side divided by the tums-ratio. If the battery voltage rises above this voltage, current is drawn from the battery side and vice versa. The same algorithm is also applied to regulate the voltage of a fully charged battery to avoid overcharging it. 93

111 Chapter 4 Bifilar converters for dual voltage applications - Part Experim ental results C om parison of m easured flux-linkage characteristics against design The static tests were performed to compare the experimentally measured characteristics o f the motor with the ones predicted by FEA during the design process. Figure 4-21 shows that the measured motor characteristics agrees closely with the characteristics simulated by FEA <+>-MMF Curves for B&D 8/4 Motor i U / Y -> J, / : 1 > r i : 1 : ' 7 i Unaligned (FEA) Aligned (FEA) Unaligned (Static test) Aligned (Static test) / r / i h /» I / * MMF (Ampere Turns) Figure 4-21: Comparison o f measured flux-linkage characteristics against design D eterm ination of flux-linkage curves over a range of ro to r angles Flux-linkage curves were measured over a range o f rotor positions over one rotor pole-pitch, i.e. over two successive aligned positions. The motor has a graded airgap, and hence the rising and falling edges o f the inductance are asymmetric. Figure 4-22(a) and (b) shows the static flux linkage curves plotted every 5 o f rotor orientation during (a) the falling inductance and (b) the rising inductance region. The effect o f the graded airgap influencing the symmetry o f the inductance profile is clearly depicted when the flux-linkage characteristics are plotted against the rotor orientation over the rotor pole-pitch as shown in Figure

112 Chapter 4 Bifilar converters for dual voltage applications - Part Flux Linkage vs Current for HV winding of B&D dual voltage motor 0.50 Flux linkage ve current (for rising inductance region) for HV winding of B&D dual voltage motor Flux Linkage (Vs) 0 > I Current (Ampere) Current (Ampere) Figure 4-22: Mains winding flux-linkage characteristics for (a) rising inductance and (b) falling inductance direction of rotation Flux Imkage vs Rotor An&c si Different Current Levels Rotor Angle (radians) aligned unaligned aligned Figure 4-23: Static magnetisation curves over one rotor pole-pitch o f the mains winding 95

113 Chapter 4 Bifilar converters for dual voltage applications - Part D ynam om eter test results Test results from the dynamometer using battery excitation have been performed are shown in Figure At the time o f these tests the peak torque was limited by the rating o f the MOSFETs. This was subsequently improved by using two in parallel. Torque vs Speed Torque vs Input Power W I > Torque (Nm) Torque (Nm) 96

114 I Chapter 4 Bifilar converters for dual voltage applications - Part 1 Torque vs Output Power Torque vs Efficiency s' Output Power (W) / s / s g 30 >s o c 25 Ui Torque (Nm) Torque (Nm) Figure 4-24: Experimental test results from dynamometer 97

115 Chapter 4 Bifilar converters for dual voltage applications - Part 1 T = 0.25 Nm, Speed = 3872 rpm T = 0.5 Nm, Speed = 3470 rpm T = 0.75 Nm, Speed = 3093 rpm 18 Acqs 2 Acqs P osition sensor signal 5V /div C1 Freq Hz Low sig n al a m p litu d e Cl Freq Hz Low sig n al a m p litu d e Low sig n al a m p litu d e Battery current 20A /div B attery voltage 25V /div I.OmV M p s Ch1 I 1 J u n 1999 (5h2I fl.ftmv M' 50'flpi H IH V 5 99 Acqs P osition sensor signal 5V /div Cl Freq Hz Low sig n al am p litu d e Cl Freq Hz Low sig n al a m p litu d e Cl Freq Hz Low sig n al a m p litu d e Battery phase current C4 M in 2 0 A /d iv Battery phase voltage 2 5 V /d iv 1 j u n 1999 j u n J u n

116 Chapter 4 Bifilar converters for dual voltage applications - Part 1 lekuuiiii lu uld A 277 Acqs TeK R un: looks/s HI Res 11 Acqs P osition sensor signal 5V /div C1 Freq 2 S 8.12 Hz Low sig n al a m p litu d e Cl Freq Hz Low sig n al a m p litu d e Low sig n al a m p litu d e Snubber inductor current 5A /div C4 Min Snubber capacitor voltage 25V /d iv 1 j u n 1999 SUB V 1 j u n 1999 Figure 4-25: Waveforms of dynamometer experimental results 99

117 Chapter 4 Bifilar converters for dual voltage applications - Part 1 In order to investigate the reasons for the lower than expected efficiency the motor has been operated from a 24V battery at different output torque levels and the results are given below. Test Measured Torque Speed Input Eff Notes output power (Nm) (rpm) Power (%) (W) (W) A B C Typical cutting condition D Lower speed but higher torque Table 4-9: Experimental results at different output torque levels An analysis has been done at the typical cutting speed (Test C) to determine the split of the losses and the results of this are shown in Figure Power Flow Analysis other loss copper loss 30% output pow er 53% Figure 4-26: Power flow analysis 100

118 Chapter 4 Bifilar converters for dual voltage applications - Part 1 The largest section of the losses is due to copper but the iron losses are still very significant. Further tests were undertaken to confirm if the iron loss was related to hysteresis or to eddy currents. Assessing the source of the iron losses In the subsequent experiment, the motor was run from the low voltage phases while the high voltage phases were used as sensing coils. The voltage on the sense coil was measured and used to calculate the flux-linkage in the high voltage coils. This is then compared to the low voltage flux-linkage curves. The results are shown in Figure Experiment Point C LV HV Sense coil = 0.01 ~ Tim e (m s) Figure 4-27: Comparison of flux linking the bifilar windings It can be seen from Figure 4-27 that the flux-linkage calculated by the integration of the sense coil voltage is almost the same as the flux-linkage calculated by integration of the source coil voltage (taking account of the resistance drop). This indicates that eddy current losses are not excessive. 101

119 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Determination of Hysteresis Losses The motor was then parked at the aligned position of one of the phases. The phase was then pulsed to mirror the same operating condition as the experimental results shown in Figure 4-25, i.e. with the same voltage and opto sensor pulsing frequency and thus aiming to reproduce the same flux-level as in the running condition. The other low voltage phase was left unconnected in this test. The objective of the static test is to remove the dynamic losses incurred in the earlier experiments and enable the iron losses to be determined. The results of the static experiments are shown in Figure Since there are no mechanical losses incurred in the static experiments, the losses classified under the category Others must be iron losses. 250 Copper Loss Others A C D Experiments Figure 4-28: Breakdown of power flow in static iron loss test Since there was no current flowing in the high voltage sense coil, the flux-linkage in the high voltage coil was obtained from the integral of the voltage. The flux-linkage curves are shown in Figure The curves show that the magnetic material exhibit considerable hysteresis characteristics. The losses resulting from hysteresis can then be calculated by determining the area under the hysteresis curve and these values agree closely with the other losses identified in the static test shown in Figure

120 Chapter 4 Bifilar converters for dual voltage applications - Part * 0.03 Experiment Point A g* 0.02 B9 <JQ <Cfl x 0.03 r- = 0.02 sere < Experiment Point C Experiment Point D 0.03 r <8 < Current (A ) Figure 4-29: Hysteresis losses in the motor It can therefore be concluded that the steel laminations used to construct the motor exhibit a significant hysteresis loss which resulted in a higher than expected iron loss. It is suggested that better grade steel could be very beneficial in this application. 103

121 Chapter 4 Bifilar converters for dual voltage applications - Part Lawnmower test results The motor has been fitted to a 13 lawnmower deck for final testing of all the electronics. Acceleration performance The speed of the motor was measured at an interval of every 100 ms to determine its acceleration profile for battery and mains from standstill to 2500 rpm and the results are shown in Figure 4-30 and Figure 4-31 respectively. Battery acceleration o g time (ms) Figure 4-30: Battery acceleration 104

122 Chapter 4 Bifilar converters for dual voltage applications - Part 1 mains acceleration E XJ 0)0) a (A time (ms) Figure 4-31: Mains acceleration Experimental waveforms across most operating modes Experimental waveforms across most operating modes were also collected and are shown in Figure Waveforms showing acceleration profile Mains with battery present O n /o ff sw itch lov/div Gate source voltage to mains IGBT 20V /div T ek Run: 2 S.0 k S /s ^Sa Acceleration from standstill to single pulse = 1.968s Gate source voltage to battery M OSFET 20V /div V oltage across battery terminal 20V /div 400 ms/div C h i' "16.6 V Cfi2 1.00'V M 4adrris"C h1 a n 1.00 V Ch V.Battery phase is turned on to regulate the battery voltage as the battery voltage rises 26jan2ooo above 27V 11:11:29 105

123 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Mains without battery O n /o ff sw itch lov/div Gate source voltage to mains IGBT 20V /div Tele R u n : k s / s ^Sa m p le A : s : s C4 M ax V C3 M ax 92 0 tn V Acceleration from standstill to single pulse = 1.720s Gate source voltage to battery M O SFET 20V /div V oltage across battery terminal 20V /div V KT^UBms 'CHI * 4 'v 2 6 J a n :1 2 :5 6 Battery phase is turned on during acceleration to regulate LV dc link voltage to 27V 400 m s/div Battery only O n /o ff sw itch lo V/div Gate source voltage to mains IGBT 20V /div TeK Run: 2S.0kS /s Sam ple A : s : s C4 M ax V C3 M ax sc omv Acceleration from standstill to single pulse = 1.592s Gate source voltage to battery M OSFET 20V /div V oltage across battery terminal 20V /div n nrv V r w v V M '4 0 0 m s C h i S 4 V 2 6 J a n :1 4 :1 8 Vgs(HV) is negatively biased to avoid mains misfiring 400 m s/div 106

124 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Waveforms during PWM mode Mains motoring Position sensor signal 5V /div T ek H riia Single Seq k S /s A : m s : s M ains phase winding current 2A /div M ains snubber voltage 250V /div M ains phase winding voltage 200V /div C4 M ean V 2.00 m s/div Waveforms during transition from PWM to single pulse mode Mains motoring Position sensor signal 5V /div Tek otnra single Seq 25.0kS/s M ains phase w inding current 2A /div M ains snubber voltage 250V /div M ains phase winding voltage 200V /div C4 M ean V 2.00 m s/div 12 JUl :5 3 :

125 Chapter 4 Bifilar converters for dual voltage applications - Part 1 Battery motoring Position sensor signal 5V /div TeK a r m S in g le S ea k S /S Battery phase w inding voltage 20V /div C1 Freq Hz Low sig n al a m p litu d e M ains snubber voltage 250V /div C4 Max V Battery phase w inding voltage 20A /div 2.00 m s/div 12 J u l :1 4 :2 9 Figure 4-32: Experimental waveforms for motor drive testing in the lawnmower deck Waveforms during single pulse mode at full speed with no load The following graphs show the waveforms collected at the motor s full speed with no load in all three different operating modes, namely mains motoring with battery present, mains motoring without battery and battery motoring. Mains motoring with battery present O n /off sw itch lov/div M ains phase current 5A /div T e k H B i a k S /s C2 F req Hz Low s ig n a l a m p litu d e Pin = 326W Speed = 3863 rpm Power per phase 2kW /div M ains phase voltage 400V /div 1.00 m s/div Chi tarrv Ch2 "S.OO ACr Ml.66ms CRT ana 2.00 v M a th l W m s 108

126 Chapter 4 Bifilar converters for dual voltage applications - Part 1 O n/off sw itch lov/div M ains dc link current 5A /div M ains pow er lkw /div M ains dc link voltage 200V /div Tek 3E73 so.oks /s m s m s C2 F req 4S9.9S5C1 HZ Low s ig n a l am plitude C3 M ean V M l M ean W C2 M ean 1.21 A 1.00 m s/div S'ffff AST' M I'.iJSms' "CRTAI 27 J a n :5 8 :0 9 Mains motoring without battery O n /off sw itch lov/div M ains phase current 5A /div T e k 2 H 5 S k S /s C2 F req Hz Low sig n a l a m p litu d e Pin = 296W Speed = 3876 rpm Power per phase 2kW /div M ains phase voltage 400V /div C2 M ean m A 1.00 m s/div w l o o m s 2 7 J a n :5 2 :0 7 O n /off sw itch lov/div Tek a n a so.oks /s M ains dc link current 5A /div M ains pow er lkw /div M ains dc link voltage 200V /div C2 F req HZ Low s ig n a l am p litu d e C3 M ean V M l M ean W C2 M ean A 1.00 m s/div 27 j a n :5 5 :

127 Chapter 4 Bifllar converters for dual voltage applications - Part 1 Battery motoring O n /o ff sw itch lov/div Battery phase current 20A /div Tek a s a s o.o k s /s C2 F req HZ Low s ig n a l am plitude C3 M ean V Pin = 229.6W Speed = 4005 rpm Pow er per phase 400W /div M l M ean 5.74m V V Battery phase voltage 20V /div C2 M ean m V 1.00 m s/div H II i n i (i M TSoms C f i r \ C h V M a th l 10.OmVV r o o m s 4.2 V 2 7 J a n :0 7 :4 2 O n /off sw itch lov/div Tek n r a g s o.o k s /s A : m s : u s Battery phase current 20A /div C2 F req H2 Low s ig n a l a m p litu d e C3 M ean 32m V Pow er per phase 400W /div M l M ean 1.20 W Battery phase voltage 20V /div C2 M ean A 1.00 m s/div V at h! W m s re.oafl-'rtajtims 27 J a n :1 3 :4 4 Figure 4-33: Waveforms during single pulse mode at full speed with no load 4.12 Battery powered drive with integral charging capability As mentioned before, the dual voltage drive can be configured as dual voltage powered drive or battery powered drive with integral charging capability depending on cost and application requirements. The dual voltage powered capability, with its much added functionality, incurs a penalty on the overall system cost. If only integral charging capability is required, it can be added to a low cost battery powered drive with minimal cost. The same converter topology would be used but the mains winding is used for charging and not mains powered motoring. It will only carry battery charging current and is therefore rated at battery charging power. Furthermore, the reason for having 110

128 Chapter 4 Bifilar converters for dual voltage applications - Part 1 two mains winding phases is to satisfy the required starting torque. For a battery powered drive with integral battery charging capability, mains motoring is not required and therefore only one bifilar coupled phase is required. This not only maximises the copper area for the principal battery windings, but also reduce the number of switches and associated drivers required for the resultant converter. Also, the motor was oversized in dual voltage drive to accommodate both windings rated for motoring needs, but if a smaller mains winding rated for battery charging power is required, a smaller motor could possibly be used, further reducing the cost of the drive Conclusions A full working prototype of the lawnmower has been produced which is capable of: Operation as a cordless lawnmower; Operation as a mains powered lawnmower with no batteries present; Operation as a battery charger when mains and batteries are present (both while rotating and not rotating; Complete electrical isolation of the battery terminals from the ac supply. The use of a C dump converter at the battery side has allowed phase demagnetisation energy to be recovered, thus maintaining the efficiency of the drive when powered from the battery. Furthermore, the energy-efficient C dump converter has a singleswitch forward voltage drop which is essential in low voltage battery powered applications. The system utilises the coupled winding in both forward and flyback modes to achieve battery charging. Furthermore, the energy recovery circuit also serves multiple purposes, namely recovering the phase commutation energy and also charging the battery. These multiple functions of the energy recovery circuit saves components which would otherwise require separate circuits. Another advantage of the drive is that it requires only one high side (floating) switch which simplifies the 111

129 Chapter 4 Bifilar converters for dual voltage applications - Part 1 converter design and ultimately its cost. All other power switches are referenced to ground. Finally, the use of a coupled winding for battery charging, as previously described [49, 50], effectively isolates the mains from the battery and thus maintains the safety of the drive during charging. The elimination of a costly and bulky supply frequency transformer from the battery charger circuit leads to a cheaper and more portable battery powered drive which contains its own integral battery charger. However there are a number of limitations on the existing solution. The converter circuit requires snubbers, which suffer from many constraints, such as the limit on the PWM frequency on the mains side and the efficiency penalties for having a dissipative snubber. Converters with inherent freewheel capability will be presented in Chapter 5. The converter provides a path for the leakage energy to be recovered to the source and eliminates the need for snubbers, thereby eliminating the restrictions the snubber had on the converters presented in this chapter. 112

130 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Chapter 5 Bifilar converter for dual voltage applications - Part Introduction The imperfect coupling between bifilar windings leads to the inevitable existence of leakage energy associated with the bifilar windings. The leakage energy has to be dealt with by the power electronic converters by allowing a safe path for the energy stored in the leakage inductance to be recovered or dissipated. Chapter 4 presented converters whereby capacitors are first used to capture the leakage energy, before being dumped in a resistor or recovered using an auxiliary dc/dc converter. The weak links in the circuits presented in Chapter 4 were the snubber and energy recovery system necessitated by the system leakage flux. At low speeds, PWM chopping is usually adopted to keep the currents within acceptable limits. At high speed chopping is no longer required as the current is naturally limited by the motor back-emf and single pulse switching is adopted. The snubber and energy recovery system would therefore be required to operate at a motoring frequency in the region of hundreds of hertz, Hz for the motor presented in Chapter 4, when single pulse mode is adopted at high speed. While at low speed when PWM chopping is used to limit the winding current, the snubber and energy recovery frequency would rise dramatically, in the region of kilohertz, possibly to as high as 20kHz if ultra-sonic chopping were required. The high leakage inductance as a result of using bifilar windings is being exacerbated by the demanding starting torque specification requirement for the lawnmower, resulting in snubbing and energy recovery requirements that would have involved large capacitances and inductances. Another problem with dissipative snubbers is that they are not energy efficient; the dissipated heat is not only wasted but also needs to be extracted from the system. This may require larger heatsinks and additional fans for cooling. Furthermore, the size of 113

131 Chapter 5 Bifilar converters for dual voltage applications - Part 2 the snubber, energy recovery components and heatsinks may be too bulky to be packaged in the portable appliance. Alternatively, converters with an inherent freewheel path, whereby the leakage energy is recovered back into the power source, can be used. The freewheeling capability provides a path for the leakage energy to be recovered and eliminates the need for snubbers and energy recovery components in the converter. This chapter presents energy-efficient power converters with inherent freewheel paths whereby the leakage energy is recovered back to the power source. 5.2 P ow er electron ic converter The shared switch asymmetric half-bridge was seen as offering the best potential for cost reduction in the battery side of the two-phase switched reluctance motor in the dual voltage drive while a voltage doubler split dc converter with small capacitors was proposed for the mains side. The complete power converter circuit for a two-phase dual voltage drive providing snubberless operation is shown in Figure 5-1: Single-switch per phase converters are cost-effective but there is often a compromise in the available voltage for energisation or de-energisation of the windings [10, 23, 44, 52]. The split dc converter successfully achieves single-switch per phase but requires two dc link capacitors and halves the available supply voltage. However it does not sacrifice control flexibility or efficiency [37]. 114

132 Chapter 5 Bifilar converters for dual voltage applications - Part 2 DH3 DH2 SH1 Mains HV Ph B HV Ph A DH4 C2 SH2 DH1 SL3 DL2 2 2 DL1 Batte r y i-- LV Ph B LVPh A SL2 SL1 Figure 5-1: Complete circuit diagram for the power electronic converter No Component Qty Description 1 SH I-2 2 IRGPH50KD 2 SL1-2 2 IRF SL3 1 IRF Cl uF 1000V Polypropylene 5 C pF 63 V Electrolytic 6 Dni, DH2 2 RGRH D l i, DL2, D CTQ100 Table 5-1: Component list for the power electronic converter 115

133 Chapter 5 Bifilar converters for dual voltage applications - Part 2 The voltage doubler split dc two-phase converter proposed for the mains side combines the split dc supply converter with a voltage doubler circuit to create a simple and low cost topology for the mains converter. The two capacitors used for voltage doubler split dc two-phase converter are of low capacitance, film type, rather than electrolytic. Traditionally, switched reluctance drives have always required significantly sized dc link capacitance. This is because ac input must be rectified and filtered to ensure that the appropriate dc link voltage is maintained, relatively free from mains frequency ripple and supply voltage fluctuation. Cost requirements meant that electrolytic capacitor is usually used, which is usually the weak link, in terms of life expectancy of the final appliance system. Electrolytic capacitors are large, expensive and have a short expected lifetime in comparison to the semiconductors. The use of film type capacitors therefore directly addresses the problem associated with the switched reluctance drive; the excessive ripple currents flowing in the dc link create problems for electrolytic capacitors leading to higher cost and potential long term reliability issues [53]. One of the disadvantages in conventional split dc converters fed from a full-bridge rectifier is the difficulty in maintaining the centre point voltage under all operating conditions. Correct operation of the circuit requires that each phase to be switched in sequence to allow balanced charging and discharging of the split dc capacitors. The split dc capacitors have therefore to be sized accordingly to supply current to each phase winding down to very low speed. In the proposed voltage doubler split dc converter, as either capacitor can be recharged by the mains supply frequency, one motor phase winding could be energised for a significant period to provide very low speed torque. The continual recharge of the capacitors by alternate halves of the mains cycle enables smaller capacitors to be used. The use of smaller capacitors maximises the voltage utilisation of the ac supply and the rating of the power switches. This is because the midpoint voltage is no longer particularly stiff as the capacitor voltage rises rapidly when charged by mains or winding commutation energy, resulting in the voltage available to energise and de- 116

134 Chapter 5 Bifilar converters for dual voltage applications - Part 2 energise the phase windings to be typically significantly greater than half the total dc link voltage rather than half the voltage as in normal split dc circuit [43]. 5.3 Configuration of the bifilar windings There are various combinations that the bifilar winding can be connected. The bifilar windings can be connected such that forward conversion on the secondary is via the diodes of the converter or anti-parallel diodes of the switches. Firstly, the connection of the bifilar windings are such that currents always flow into the dotted terminal of the primary winding during positive voltage loops, as shown in Figure 5-1. This would ensure that if current is flowing in both primary and secondary windings, their MMF are not in opposition and that positive voltage loops across both windings would always develop a torque producing current. Secondly, both the mains and battery windings are connected such that forward conversion is through the anti-parallel diodes of the switches and not the discrete diodes. When the mains windings are energised in a Positive Voltage Loop (PVL), the increasing flux in the mains windings induce a positive voltage at the dotted terminal of the coupled battery windings, as shown in Figure 5-2. If the battery windings are connected such that forward conversion is through the discrete diodes and not the anti-parallel diodes of the switches, as shown in Figure 5-2a, switching on the battery switches in synchronism with the position sensor during mains PVL would result in a large circulating current flowing through the windings, as shown in Figure 5-2b. 117

135 Chapter 5 Bifilar converters for dual voltage applications - Part 2 S2 ik D1 S2 VDC' T C1 Vinduced Vinduced S3 D2 02 S3 (a) M ains P V L, V induced forces forward conversion paths through discrete diod es (b) M ains P V L, V mduced a llo w s inappropriate circulation paths w hen sw itch es are turned on Figure 5-2: Forward conversion paths through discrete diodes during mains PVL The battery windings are therefore connected as shown in Figure 5-3, such that forward conversion flows through the anti-parallel diodes of the switches. Turning on the battery switches in synchronism during mains powered motoring does not result in large circulating currents. S2 S2 VDC- VDC- Vinduced Vinduced D2 S3 D2 S3 (a) M ains P V L, V induced forces forward conversion paths through anti-parallel d iod e o f sw itch es (b) M ains P V L, current i w ill flo w in the preferred bifilar path w h en sw itch es are turned on Figure 5-3: Forward conversion paths through anti-parallel diode o f switches during mains PVL This method of connection is preferred because it prevents any circulating current when switches across both side are switched on and off in synchronism with the 118

136 Chapter 5 Bifilar converters for dual voltage applications - Part 2 position sensor signals. Current will flow in the preferred path depending on the mains dc link voltage, the battery voltage and the tums-ratio o f the coupled windings. This self-balancing feature of the bifilar winding is utilised to prevent the battery from overcharging during mains operation. This is achieved by switching on the correct battery phase as dictated by the position sensor during mains motoring to discharge the fully charged battery. Current will flow from the battery in a torque producing sense, supplementing mains motoring and preventing the battery from overcharging. 5.4 Operation stages of a dual voltage drive with freewheel capability The consistent definitions of primary windings, primary converters, secondary windings and secondary converters, as presented in Chapter 4, are adopted through out this chapter Stage one Closing the primary switch(es) builds up current in the primary winding. Current flows in the secondary winding as a result of close coupling via the anti-parallel diodes of the switches. The secondary current stops flowing when the voltage across the secondary dc link rises above a threshold. The magnitude of the threshold is governed by the tums-ratio between the coupled windings Stage two Opening the primary switch(es) commutates the current in the primary winding. Current will transfer to the secondary or continue to flow in the primary depending on whether it is battery or mains powered motoring. Some current will definitely continue to flow in the primary until the leakage flux has been reset. 119

137 Chapter 5 Bifilar converters for dual voltage applications - Part 2 When commutating during battery powered motoring, current will initially transfer to the mains winding, flowing into one of the mains split dc link capacitors. As the mains dc link capacitance is small, the voltage across it quickly rises above the voltage across the battery dc link when referred through the tums-ratio, resulting in the battery winding being the preferred current path. The majority of current therefore continues to flow in the battery winding, returning the winding commutation energy to the battery and its dc link capacitor. When commutating during mains powered motoring, current will initially flow in the mains side charging the mains dc link. The voltage across the small mains dc link capacitance will increase rapidly and result in the majority of the current transferring across the coupled winding to flow in the battery winding, charging the battery and its dc link Stage three Current in both the primary and secondary has fallen to zero, resetting the winding flux. Current will always flow in the preferred path depending on the tums-ratio and the voltage across both the primary and secondary capacitors. 5.5 Discussion of modes of operation Mains powered motoring Mains motoring is achieved by the magnetisation and demagnetisation of phases by switching Shi and Sh2 according to the rotor position sensors. At low speed it is usual to limit the motor phase currents with PWM control. A unique PWM algorithm is developed for the voltage doubler split dc link with small capacitance whereby the PWM duty cycle is both dependent upon the motor speeds and also modulated against the mains cycle. Although the mains converter has no zero voltage loop capability, the availability of zero voltage loop across the battery side allows a semi-soft chopping, 120

138 Chapter 5 Bifilar converters for dual voltage applications - Part 2 whereby the majority of current transfers and freewheels in the battery windings while only current associated with the leakage inductance continues to flow in the mains winding. The use of the split dc link requires each phase to be switched in sequence to allow balanced charging and discharging of the split dc capacitors. However using PWM at low speed results in imbalanced voltages between the split dc links, as chopping one phase continuously would overcharge the opposite capacitor. This is because during negative voltage loop the phase energy extracted from the off-going phase charges the opposite capacitor. Complementary switching, whereby both IGBTs are switched in complement to each other, is therefore adopted during initial start-up to avoid voltage building up in the opposite capacitor. At higher speed, complementary switching is important to maintain a more balanced capacitor voltage between mains positive and negative cycles by continually extracting phase energy to charge the opposite capacitors when they are not charged by mains, thereby minimising the effects of supply frequency modulation. The PWM algorithm with complementary switching will be explained in detail later in the chapter. Although single pulse is adopted at high speed, the IGBTs are still being modulated at mains peak in an attempt to minimise the ripple effect the voltage doubler has on the small dc link capacitances. The control strategy is best elaborated in conjunction with the operating modes of the motor. Figure 5-4(a) - (e) shows the modes for the motor phase A. 121

139 Chapter 5 Bifilar converters for dual voltage applications - Part 2 DH3 M ains " SH1 T C1 DH2 : : C1 DH2 M ains HV Ph B' HV Ph"B DH3 SH1 HV P h A HV Ph A DH4 C2 SH2 SH2 DH4 DH1 DH1 B atte ry!-- Battc L V P h B L V P h A LV Ph B LV Ph A (a) P hase A p o sitiv e v o lta g e loop during m ains p o sitiv e c y cle (m ains charging C l), forward conversion trickle charges the battery (b) Phase A p o sitiv e volta g e loop during m ains negative c y cle (m ains charging C 2), forward conversion trickle charges the battery M ains H V P ffb HVPh A' DH4 SH 2 DH1 (c) P hase A zero v o lta g e lo o p, o n ly leakage current flo w s in m ains w in d in g w h ile m ajority o f current transfers to the secondary w in d in g 122

140 Chapter 5 Bifilar converters for dual voltage applications - Part 2 DH3 Mains (gh SH1 t C1 DH2 C1 DH2 HVPh'B DH3 Mains G> HV Ph B SH1 H V P h A" HV Ph A' DH4 ±: C2 SH2 C2 SH 2 DH1 DH4 DH1 (d) P hase A n egative v o ltage loop during m ains p o sitiv e c y cle (m ains charging C l), current transfers to the coupled w in d in g to flo w into the battery (e) Phase A n egative voltage lo o p during m ains negative c y cle (m ains charging C 2), current transfers to the coupled w ind ing to flo w into the battery Figure 5-4: Principal operating modes of voltage doubler split dc converter In mode (a), phase A is in positive voltage loop when mains is in positive cycle, resulting in the voltage ripple across the small dc capacitor Cl being applied to phase A. Therefore the speed dependent duty cycle of S h i has to be further modulated against mains cycle to eliminate the effects of supply frequency modulation. At mains zero crossing, the speed dependent duty cycle is used. The speed dependent duty cycle is slowly being reduced as mains voltage rises, reaching a minimum at mains peak. During the positive voltage loops, a voltage is induced in the coupled winding which would tend to charge the battery. The modulation of PWM against mains cycle ensures that the forward conversion mode does not overcharge the battery. In mode (b), phase A is in positive voltage loop when mains is in negative cycle and is not charging C l. Here the duty cycle of S h i is only dependent on motor speed. Cl is being discharged by phase A while capacitor C2 is being charged by mains. Depending of the state of the battery charge and the voltage across the mains dc link capacitance, some current might flow in the battery winding through forward conversion mode. 123

141 Chapter 5 Bifilar converters for dual voltage applications - Part 2 In mode (c), when S h i is switched off, S l 3 is turned on at the same instance. A zero voltage loop is established across the battery side and would be the preferred current path in the bifilar windings, as shown in Figure 5-4c. The majority of current therefore transfers to the battery side and freewheels through Su- Only the leakage current that are not coupled continues to flow in the mains side. In mode (d) and (e), all switches are turned off, the negative voltage loop of phase A occurs during mains positive and negative cycles respectively. In (d), the energy extracted from phase A charges capacitor C2 while capacitor Cl is recharged by mains. In (e), the commutation energy from phase A charges capacitor C2, and capacitor C2 would be further recharged by mains if the mains voltage were able to forward bias diode D h 4 - A s the mains dc link capacitance is small, the phase commutation energy will quickly charge it, resulting in the majority of the phase energy transferring to the coupled battery winding to flow into the battery and its dc link capacitance. Similar operating modes occur for the magnetisation and demagnetisation of phase B Battery charging In Figure 5-5(a), Shi is turned on and current builds up in HV phase A, some current will flow in the battery windings as a result o f forward conversion. In Figure 5-5(b), S h i is switched off and S h 2 is turned on simultaneously. The decaying magnetic field in HV phase A induces a negative voltage at the dotted terminal of LV Phase A. A current will flow in LV phase A through flyback mode of the converter. At the same time, the increasing magnetic field in HV Phase B induces a positive voltage at the dotted terminal of LV Phase B, resulting in a current flowing in LV Phase B through forward conversion mode. The combination of the flyback and forward conversion modes between phase A and phase B results in a current that is flowing in series across both LV Phases into the battery. 124

142 Chapter 5 Bifilar converters for dual voltage applications - Part 2 M S hi turned on Shi turned o f f and Sh2 turned on sim ultaneously SH1 HV P h A ' H V P h A ' DH4 SH 2 C2 SH2 DH4 DH1 DH1 SL3 DL1 DL2 l ^ SL3 A L DL1 Battei yi- t C LV Ph B LV Ph A LV Ph B LV Ph A SL2 SL1 SL2 SL1 (c) S H2 turned on (d) S H2 turned o f f and S Hi turned on sim ultaneously Figure 5-5: Operating modes of the charging algorithm Both switches are off and the windings are reset. In Figure 5-5(c), Sh2 is switched on and current builds up in HV phase B while some current flows in LV phase B through forward conversion mode. 125

143 Chapter 5 Bifilar converters for dual voltage applications - Part 2 In Figure 5-5(d), Sh2 is switched off while SHi is turned on, this operation mode is similar to the one shown in Figure 5-5(b), with the resultant current flows in series across both LV phases into the battery as a result of the combination of the flyback and forward conversion modes between phase A and phase B. Both switches are off and the windings are reset. The control strategy adopted is to maximise the utilisation of both windings to charge the battery. This is achieved by ensuring that the battery charging current would flow through both LV phases in series, hence the need of operating modes shown in Figure 5-5(a) and (d) to prime the HV phases before forcing a current flowing in series through both phases into the battery in operating modes shown in Figure 5-5(b) and (e). The switching control strategy is devised such that battery charging is achieved irrespective of the rotor position. Both HV Phases are primed in turn to ensure that current flow through the LV phases in the direction depicted in operating modes shown in Figure 5-5(b) and (e) repetitively. Operating modes shown in Figure 5-5(b) or (d) will dominate the charging current depending of the rotor position Battery powered motoring Switches S li, S l 2 and S u are controlled in response to the rotor position. For magnetisation of phase A, S li and S l 3 are turned on to impress the battery voltage on phase A, building current in it. For the magnetisation of phase B, Sl2 and SL3 are turned on instead. During the magnetisation of the battery phases, forward conversion forces current to flow into the coupled mains windings into the mains dc link via the anti-parallel diodes of the switches. Once the mains dc link is charged, negligible current will flow in the mains windings. A zero voltage loop is available on the battery side, allowing soft chopping to be employed to control the phase current at low speed. To achieve the zero voltage loops in the corresponding phases, SL3 is turned off while the bottom switch associated with 126

144 Chapter 5 Bifilar converters for dual voltage applications - Part 2 each phase is kept on, current freewheeling in a zero voltage loops via D o and one of the turned on switches. All the battery switches are turned off during phase demagnetisation. Current initially transfers across to the mains winding to charge the mains dc link capacitance, but the voltage of the small dc link capacitance quickly rises above the voltage across the battery and its dc link when referred through the tums-ratio. The majority of current therefore continue to flow in the battery windings, returning the phase energy back to the source. 5.6 Im plem entation Control circuits In order to achieve full isolation between the battery and mains side, the control of the two converters were separated. Two separate microcontrollers, one on each side of the isolation barrier, were used to control all the operating modes o f the lawnmower. The battery microcontroller controls the battery motoring and acts as the master microcontroller and sends encoded instructional signals to the HV microcontroller. The HV microcontroller, acting as a slave microcontroller, decodes the instmctions received from LV microcontroller and controls the mains motoring. Table 5-2 described the communication and synchronisation between the HV and LV microcontroller through the three optocouplers. A brief description of the operating modes of the LV and HV microcontrollers are given in Table 5-3: 127

145 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Optocoupler Direction Function Optocoupler 1 Battery to mains Instruction channel signals are encoded (the signals conveyed are mains positive, mains negative, charging and switch off signals) Optocoupler 2 Battery to mains Data channel position sensor signal is conveyed during motoring, battery state is conveyed during charging Optocoupler 3 Mains to battery Mains zero crossing signal to modulate mains PWM with respect to mains cycle to prevent battery from being drained when mains is not present Table 5-2: Description of the optocoupler functions User switch Mains Battery Operating mode No power Opto3 cuts logic power to battery side to prevent draining the battery HV microcontroller powers up and initiate charging, Opto3 switches on LV logic and wakes LV microcontroller, which send battery charging signal across, confirming charging state. No battery is detected and system shuts down again HV microcontroller powers up and initiate charging, transferring energy to LV side, powering up the LV microcontroller, which would monitor the dc link voltage and send appropriate signal across HV side to start, stop 128

146 Chapter 5 Bifilar converters for dual voltage applications - Part 2 or float charging No power Closing the switch powers up LV control, which would sense user switch on, mains absence and start battery motoring and send mains off signal to HV microcontroller. Note : HV microcontroller would be powered up by the transferred energy across the bifilar winding during battery motoring HV microcontroller would charge LV dc link, and power the LV microcontroller up. LV microcontroller would sense mains presence and user switch is on and send instruction through Optol to initiate mains motoring. LV microcontroller would monitor LV dc link voltage and switch on MOSFETs to maintain the dc link voltage Mains presence would trigger LV microcontroller via Opto3, which would switch MOSFETs off and send instruction through Optol to initiate mains motoring. Battery would be charged by the commutation energy and the LV microcontroller would monitor battery voltage and switch on MOSFETs to prevent the battery from overcharging. Table 5-3: Operating modes o f the lawnmower system 129

147 Chapter 5 Bifilar converters for dual voltage applications - Part 2 The purpose of each of the 3 optocouplers in each operating mode are given in the following table: User Mains Battery Optol (Instruction channel) Opto2 (Data channel) Opto3 (switch and logic signal) Description Switch off Opto3 prevent battery drainage Charge instruction Dc link state Switch on/ mains present Charge Battery Switch on/ instruction state mains present Opto3 maintains supply to LV microcontroller, which monitors the dc link voltage and relay it via Opto2 Opto3 maintains supply to LV microcontroller, which monitors the battery voltage and relay it via Opto Mains off Switch off/ LV microcontroller powered instruction mains not up by user switch, HV present microcontroller is powered up but kept off by Optol turn o ff instruction Mains Position Switch on/ Mains charges LV dc link, Motoring sensor mains which supply power to LV present microcontroller Mains Position Switch on/ Motoring sensor mains present Table 5-4: Description of the optocouplers in each operating modes 130

148 Chapter 5 Bifilar converters for dual voltage applications - Part 2 The communication and synchronisation between LV and HV microcontrollers are summarised below: Operating mode LV microcontroller HV microcontroller Battery motoring and Braking Mains motoring Battery charging LV microcontroller sends turn off signal of 60 ps every 10ms to HV microcontroller LV microcontroller sends 520ps to indicate mains positive zero crossing LV microcontroller sends 290ps to indicate mains negative zero crossing LV microcontroller sends charging signal of 170ps every 10ms HV microcontroller decodes off signal and keeps IGBTs off Upon decoding mains zero crossing signals, HV microcontroller counts each timer period to determine the mains cycle and adjust the PWM duty cycle accordingly Table 5-5: Description of communication and synchronisation between LV and HV microcontrollers 520uS j Mains positive zero crossing 290uS Mains negative zero crossing 170uS Charging 60uS 0... u r -! Switch Off Figure 5-6: Timing diagram for instruction channel codes 131

149 Chapter 5 Bifilar converters for dual voltage applications - Part 2! C ix r W " 1 z J ~ - S Hi U - I T a p ~ H J 3 t <> <x -c=h x n r* _a _ a j 0*. ZTX7i O H s; l - «r < T & -fchh O-, - f e ""O- " O- <0>- I X -o 77; C* * ISp>; W N* UcT ^ P! 0 "0 I-HHh,., 0 N i l ZTX771 - ib- ss.«-+l,v rife X T,?=»: "" ---- NC 1.0 r'x x = u u lyjicrotgax^ Figure 5-7: Schematic layout of the control circuit diagram 132

150 Chapter 5 Bifilar converters for dual voltage applications - Part 2 The inputs to the LV microcontroller are: Mains zero crossing Portb,0 (active high) A rising edge received on optocoupler 3 indicates beginning o f mains positive cycle On/Off Portb,l (active high) This signal is dropped down from +24V supply voltage and fed through a RC filter into the microcontroller input Voltage comparator Portb,2 (active high) Battery voltage and a reference voltage are fed into a voltage comparator LM 311 and the output is fed into the microcontroller input Mains present Portb,3 (active high) The mains zero crossing signals are used to charge a lpf to provide a mains present signal Opto sensor Portb,4 Schmitt triggered output opto HOA2001 is used for the opto position sensor. The sensor output is further filtered across a RC filter into microcontroller input Table 5-6: LV microcontroller inputs The outputs from the LV microcontroller are: S li Porta,0 These signals are fed directly into gate drivers IR2113 and S l2 Porta, 1 INT202 S l3 Porta,4 Instruction Porta,2 Encoded signals are transmitted on this channel channel Data Channel Porta,3 Position sensor signal is conveyed during motoring, battery state is conveyed during charging Self-sustain Portb,5 Microcontroller supply self-sustain signal. This enables the microcontroller to sustain its power to provide control instructions during braking when the user switch is off. Table 5-7: LV microcontroller outputs 133

151 Chapter 5 Bifilar converters for dual voltage applications - Part 2 The inputs to the HV microcontroller are: Instruction Portb,0 Encoded signals are received on this channel channel Data Channel Portb,5 position sensor signal is conveyed during motoring, battery state is conveyed during charging Table 5-8: HV microcontroller inputs The outputs from the HV microcontroller are: Shi Porta,0 These signals are fed directly into gate drivers 1R2213 Sh2 Porta, 1 Table 5-9: HV microcontroller outputs Battery charging system A 4 stage charging system has been implemented: 1. Rapid charging at 2.2A until battery voltage first reaches 29.4V 2. Medium charging at 0.5A until battery voltage reaches 29.4V again 3. Low charging at 0.2A until the voltage reaches 29.4V for a third time 4. Float charging at 28.6V The system relies on the rate of rise of the voltage across the LV dc link to differentiate between battery being present and not present. When the battery is not present, the voltage across the dc link rises rapidly upon charging and triggers the system into stage 4 of float charging without going through stage 2 and 3. The voltage sensing is achieved using a LM 311 comparator and is transmitted to the HV microcontroller using the data channel optocoupler

152 Chapter 5 Bifllar converters for dual voltage applications - Part 2 At start-up, the HV microcontroller waits for 60ms before checking the data channel (battery state), if the battery is fully charged within the first 60ms, then the battery is not present. The system is therefore charging a dc link capacitor which voltage rises rapidly upon charging. If the battery is not fully charged within 60ms, then rapid charging is engaged to charge the battery. The system would step to the next stage of charging when it detects a high on the data channel (battery state). When progressing through a charging stage, the system waits for approximately 2 seconds before next checking the data channel. Both motor phases are used to deliver maximum battery charging capability. The gate voltage waveforms for each of the stages are shown below with the exact times given in Table lek kleee b.ooms/s 14? Acqs ] Rapid current charging C4 +W idth 17.00us C4 -W id th S Gate soure voltage o f (2V /div) C l + w id th 6.20JJS C l -W id th 22.00JJS G ate soure voltage o f Sh2 (2 V /d iv ) Ch V M 10.Ojas Ch4 I 2.68 V 28 Apr

153 Chapter 5 Bifilar converters for dual voltage applications - Part 2 le k id U U J b u U W s "Scqs E T ] M edium current charging C4 +Width S.OOjiS C4 -W idth 26.80ns G ate soure voltage o f Shi (2 V /d iv ) Cl +Width 2.60ns Cl -W idth 29.20ns G ate soure voltage o f Sh2 (2 V /d iv ) a m '2.00 v" Ch V -T M 10.0ns Ch4 J 2.68 V 28 Apr :16: M5A... I T 5 Acqs L ow charging current C4 +Width 2.80ns C4 -w id th 34.20ns G ate soure voltage o f Shi (2 V /d iv ) C1 +Width 2.80ns C1 -W idth 34.20ns G ate soure voltage o f Sh2 (2 V /d iv ) aanxoov Ch V M 10.0ns Ch4 I 2.68 V 28 Apr :18:10 T e k a s n j s. o o m s / s [ T 8 Acqs F loat v o ltage charging C4 +Width i.s o n s Low resolution C4 -W idth 33.20ns Unstable histogram Cl +Width 1.80ns Low resolution Cl -W idth 33.20ns Unstable histogram G ate soure voltage o f Sh! (2 V /d iv ) G ate soure voltage o f Sh2 (2 V /d iv ) [ V" Ch V M 10.0ns Ch4 / 2.68 V 28 Apr :19:37 Figure 5-8: Switching algorithms for battery charging 136

154 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Shi on Shi on Shi off SHi on Shi off Shi off Period Sh2 on Sh2 off Sh2 off SH2 on SH2 on Sh2 off Rapid (2.2A) 6.2 ps 10.8ps 11.2 ps 6.2 ps 12.0 ps 21.6 ps 68 ps Medium (0.5A) 2.6 ps 2.4 ps 26.8 ps 2.6 ps 2.4 ps 31.2 ps 68 ps Low (0.2A) 2.8 ps 0 ps 34.2 ps 2.8 ps 0 ps 28.2 ps 68 ps Float (28.6V) 1.8 ps 0 ps 33.2 ps 1.8 ps 0 ps 31.2 ps 68 ps Table 5-10: Switching algorithm for battery charging system The new lawnmower charger was compared to existing Black and Decker plug-top chargers by charging fully discharged batteries. Battery Charging system Notes 12 VB&D 12V plug-top charger Tests ran in parallel but 24 V SR Switched reluctance were stopped and restarted lawnmower charger over several days 24 V B&D 24V plug-top charger Test ran continuously Table 5-11: Comparison of 3 battery charging system The waveforms at each of the charging stages in the switched reluctance three stage charger are shown in Figure

155 Chapter 5 Bifilar converters for dual voltage applications - Part 2 S w itched R eluctance M o to r C h a rg in g 24V B attery a 20 o 15 V oltage Current Time (Hrs) B lack & D ecker 12V C h a rger a> o> 2 o > V oltage Current Time (Hrs) B lack & D ecker 24V C harger 0.9 5) V oltage Current Time (Hrs) Figure 5-9: Charging current and voltage profile of 3 battery charging system 138

156 Chapter 5 Bifilar converters for dual voltage applications - Part 2 l e k tfm H M 5 A '8 3 Acqs Rapid charging C 3 M e a n V G ate soure voltage o f Shi (5V /div) C4 Max 6.0 A C4 M ea n Battery voltage (2 0 V /d iv ) Battery current (5 A /di v) C h i ' 5 :66 V Ch2 ' S. Off V afljg 1.00 V C h AO 3 0 J a n :5 5 :3 9 le k H M ^ U.5 0 M 5 A Acqs Medium charging C 3 M ean V Gate soure voltage o f Shi (5V /div) 2-»- C4 M ax 3.8 A C4 M ea n m A Battery voltage (2 0 V /d iv ) Battery current (5 A /d iv ) C h t C h 3 S.'O'fl V V V M n s C h 4 > m A AO I er'h lihii 2. SOM S /s Acqs : T Low charging C3 M ea n V G ate soure voltage o f S h i (5 V /d iv ) C4 M ax 1.8 A C4 M ean 190m A B attery voltage (2 0 V /d iv ) B attery current (5 A /d iv ) C h A O 3.2 V Figure 5-10: Experimental waveforms using 8/4 motor to charge the battery 139

157 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Battery discharge to determine Amp Hour rating In order to determine the effectiveness of the battery charging using the three charging systems the batteries were discharged across resistor banks to determine the capacity of the batteries. Battery Discharge V Battery 24V Motor charged 24V B&D charged Time (Hrs) Figure 5-11: Battery discharge profile for 3 charging system The area under the current - time curve is then integrated to arrive at the capacity of the battery (Ah). From the integration, the capacities of the batteries are: 12V battery charged by Black & Decker 12V battery charger : Ah 24V battery charged by switched reluctance motor charger : Ah 24V battery charged by Black & Decker 24V battery charger : Ah Since the initial condition of the batteries is not known, the slight difference in the battery capacities might be attributed to the battery condition. However, it has been shown that all the three charging system investigated have achieved adequate battery charging. Battery charging using the switched reluctance motor reaches the float charge stage in less than 20 hours. The equivalent 24 V plug-top transformer charger takes over 60 hours to reach the same level of charge. 140

158 Chapter 5 Bifllar converters for dual voltage applications - Part 2 The lawnmower system has been shown to be effective in battery charging. The system also offers considerable flexibility to minimise the total charge time by an optimum choice of the charging current in each stage Software design Start-up algorithm An algorithm is developed such that if turning on the phase that corresponds to the rotor position does not successfully turn the rotor after a prescribed time, it is assumed that the rotor is in the aligned or near aligned position where the rotor sits in equilibrium. At this position, the opposite phase is in the unaligned position and pulsing the opposite phase should pull the rotor from the equilibrium position. Start-up implementation During battery motoring, LV microcontroller will detect the changes in state of the position sensor and will determine whether the motor is starting correctly from standstill. If the position sensor transition does not occur in the prescribed time the LV microcontroller will switch on the opposite phase. During mains motoring, if the position sensor transition does not occur in the prescribed time the LV microcontroller will override the position sensor signal to HV microcontroller requesting a change in phase energisation after a set time without the HV microcontroller software having to be modified. HV microcontroller will only act on instructions received and will not even be aware of the phase swapping. The LV microcontroller has been programmed to: swap mains phase by changing the position sensor data channel after a set time 141

159 Chapter 5 Bifllar converters for dual voltage applications - Part 2 turn on corresponding LV switch when the phases are swapped (This is to provide ZVL during chopping) stop the starting procedure after swapping opposite phase for 4 times until the user switch is next turned off and on again. PWM routine The PWM duty cycle is progressively increased as the speed increases to provide the acceleration torque required while still keeping the current and dc link voltages within safe operating limits. A short blanking pulse in applied at the beginning of each current pulse to rapidly build up the phase current while the winding inductance is still low. The profile of the PWM ramp for mains and battery motoring are shown in Figure During mains motoring, a system whereby high voltage motoring duty cycle is modulated by the mains 50 Hz cycle is required to prevent overvoltages across the relatively small 20pF split dc link capacitors. At each position sensor transition, a speed dependent duty cycle is calculated, as shown in Figure 5-12 above. The microcontroller then tracks the mains cycle at each timer interrupts and modulates the duty cycle against the mains cycle. At mains zero crossing, the speed dependent duty cycle is used, but the speed dependent duty cycle is slowly being reduced as mains voltage rises, reaching a minimum at mains peak. 142

160 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Duty cycle for battery motoring I ~ o o Q , 1 _t L l t i i» i : ->V V f * S p e ed (rpm) Duty cycle for mains motoring Speed (rpm) Figure 5-12: PWM duty cycles Furthermore, complementary switching, whereby the switches of the two-phase windings are driven in complement to each other, is adopted during PWM. This method has proven to be effective as complementary switching transfers energy from one capacitor to the other capacitor, and thus allowing one capacitor to supply the phase current even when mains is charging the other capacitor. However the complementary switching duty cycle is limited to 50% to avoid currents building up in the non-motoring phase. The implementation of this strategy has resulted in a smoother and faster acceleration of 1.248s from standstill to 3000 revs/min, which is 143

161 Chapter 5 Bifilar converters for dual voltage applications - Part 2 comparable to the battery acceleration. The improved acceleration profile is shown in Figure m a i Single Seq 50.0k5/s H V split dc total voltage (500V /div) A : Sm s s C4 Max 1.60 V H V rectifier diode current (40A /div) C2 Max 560 V C4 RMS 332mV Position sensor signal (5V /div) C l Freq H2 Low sig n al am plitude 3 i V 3 ju n :37:37 Time to 3000rpm = 1,248s Figure 5-13: Acceleration profile using main motoring Single pulse routine In battery motoring, at speeds above 2700 rpm, the PWM duty cycle reaches 100% and beyond this, speed control is achieved by varying the pulse width of the single pulse applied within each 45 degrees rotor angle. A hysteresis of 200 rpm is also implemented to avoid oscillations between single pulse and PWM modes and the system will only re-enter PWM mode when the speed falls below 2500 rpm. In mains motoring, single pulse mode is entered at 1800 rpm. But even in single pulse mode, the IGBTs are still chopped at mains peak to balance the mains voltage ripple resulting from the voltage doubler circuit. The chopping duty cycle at mains peak is being increased at speeds above 2700rpm. In both modes, the switching algorithm limits the no load speed to just below 4000 rpm by cutting back the pulse width progressively as the speed increase. A summary of the switching algorithm during mains motoring is shown in Table

162 Chapter 5 Bifllar converters for dual voltage applications - Part 2 Opto High Opto Low Mains +ve Mains -ve Mains +ve Mains -ve Top capacitor charged by mains, Bottom capacitor charged by leakage Bottom capacitor Top capacitor charged by mains, charged by mains, Top capacitor Bottom capacitor charged by leakage charged by leakage Bottom capacitor charged by mains, Top capacitor charged by leakage Initial startup (>525rpm) SH1 PWM start-up duty cycle modulated by mains Complement of SH1 (fixed at 50% duty cycle) PWM start-up duty cycle Complement of SH1 (fixed at 50% duty cycle) Complement of SH2 (fixed at 50% duty cycle) PWM start-up duty SH2 cycle SL1 on to provide ZVL on to provide ZVL off off Complement of SH2 (fixed at 50% duty cycle) PWM start-up duty cycle modulated by mains SL2 off off on to provide ZVL on tojdrovide ZVL SH1 PWM speed dependent duty cycle modulated by mains Low Speed ( rpm) SH2 off off PWM speed dependent duty cycle off off PWM speed dependent duty cycle SL1 on to provide ZVL on to provide ZVL off off PWM speed dependent duty cycle modulated by mains SL2 off off on to provide ZVL on tojdrovide ZVL Single pulse - speed dependent pulsewidth PWM at low duty cycle at mains peak Single pulse - Speed dependent pulse-width SH1 off off Single pulse - speed dependent pulse-width Single pulse - Speed PWM at low duty dependent pulse- cycle at mains width peak SL1 off off off off Medium Speed (2000rpm rpm) SH2 off off SL2 off off off off SH1 Single pulse - speed dependent pulsewidth PWM at high duty cycle at mains peak Single pulse - Speed dependent pulse-width off off High Speed (>2700rpm) SH2 off off Single pulse - Speed dependent pulsewidth SL1 off off off off SL2 off off off off Table 5-12: Mains switching algorithm Single pulse - speed dependent pulse-width PWM at high duty cycle at mains peak 145

163 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Load algorithms The load algorithm is designed to deliver a target of 0.5 Nm at 3500 rpm. As the load increases further the target is to deliver 1.1 Nm at 2500 rpm. This is the maximum power deliverable and beyond that, the motor will re-enter PWM routine and follow the PWM ramp down. This controllable ramp down to a stop will ensure that at all instances, the current and snubber voltage is maintained within controllable limits. The load algorithms for battery and mains powered motoring are shown in Figure Battery motoring single pulse width ^ 30 = Speed (rpm) Mains motoring Single Pulse Width J 50 > 40 o > 30 Q Speed (rpm) Figure 5-14: Load algorithms 146

164 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Stall condition The initial PWM duty cycle is set such that in the event of the motor stalling, the PWM starting duty cycle on mains and battery are both low enough to prevent the current and dc link voltage from rising out of control. A start-up duty cycle of 38% was selected for battery motoring. The initial duty cycle for mains was set at 30%, complementary switching was also adopted to avoid the dc link capacitor voltage from rising out of control. The complementary switching in mains motoring is limited to a maximum of 50% to avoid too much current building up in the nonmotoring phase. Power down and braking When the user switches off, the system utilises the battery to brake the motor. Braking is therefore not possible when battery is not present. Plugging was used and additional power was drawn from the battery to brake the motor. This is achieved by energizing the phase with decreasing inductance to produce negative torque to brake the motor. The existing algorithm brakes the motor to about 250 revs/min, after which it waits for another 5s before initiating battery charging or cuts off the supply depending on whether mains and/ or battery is present. Voltage regulation scheme During charging, the state of the battery charge is conveyed through the data channel optocoupler to the HV microcontroller which determines the charging rate to avoid overcharging the battery. A voltage regulation scheme is also required during mains motoring as the motor commutation energy transfers energy to the battery. To avoid overcharging the 147

165 Chapter 5 Bifllar converters for dual voltage applications - Part 2 battery, Sl3 and one of the switches associated with each phase are turned on according to the position sensor signal when the battery reaches 29.4V. The switches are turned on with a fixed duty cycle at a frequency 18kHz to discharge the battery, thus preventing it from overcharging. 5.7 Experimental results Dynamometer test results The experimental results under all motoring modes, namely mains powered motoring with battery present, mains powered motoring without battery present and battery powered motoring are shown in the Figure The system achieves the highest efficiency in the battery powered motoring mode where it is most important. In all experiments, the motor is loaded until the stall condition. 148

166 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Torque Speed * 2500 Mains no bat Mains with bat Bat only Torque (Nm) Efficiency Mains no bat Mains with bat Bat only Torque (Nm) 1.5 Input and Output Power 900 ^ 600 Z 500 0) Pin with bat Pout with bat Pin no bat Pout no bat Pin bat only Pout bat only Torque (Nm) Figure 5-15: Experimental results for 8/4 motor 149

167 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Battery motoring The waveform below shows the current in the battery and LV dc link during battery motoring. 2 A cqs L V dc link current (loo A/div) B B attery current (5 0 A /d iv ) C h4 S.OOV 0.8Nm Tek Run: 2.50M S /S Sample EE L V dc link current (lo O A /div) C4 M in Battery current (50A /d iv ) B IB s.oov M2.0Dms' Ch3 /" 'm V Figure 5-16: Waveforms showing battery motoring for 8/4 motor 150

168 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Mains motoring with battery present No load T e k h m h l.oon 4 5 A cqs H V total split dc volta g e (5 0 0 V /d iv ) L V dc link voltage (2 0 V /d iv ) C4 M in L V dc link current (lo O A/div) H V Phase A current (lo A /d iv) 0.8Nm T e k R u n : M S /S S a m p le H V total split dc voltage (5 0 0 V /d iv ) L V dc link voltage (2 0 V /d iv ) L V dc link current (lo O A /div) H V Phase A current (lo A /d iv) Figure 5-17: Waveforms showing mains motoring with battery present for 8/4 motor Waveform above shows during single pulse mode IGBTs are still PWM controlled at mains peak. PWM control enables a wider current pulse to be applied to the motor winding as opposed to a narrow single pulse and this has improved both the mains motoring efficiency and speed Lawnm ower test results The motor has been fitted to a 13 lawnmower deck for final testing of all the electronics. The waveforms showing transitions between some o f the operating modes are shown 151

169 Chapter 5 Bifilar converters for dual voltage applications - Part 2 in Figure Triggering Mains User Events: turn on turnoff Mains User Mians turn on turn on turn off Tek Ru n: lo.o k s k s Sam ple f-t H V dc lin k (500V /div) C2 Max 840 V L V v o lta g e (20V /div) C1 Max 1.60 V C3 Max 25.6 A L V current (20A /div) C3 Mean 700mA H V P h a se A current (lo A /d iv ) 500 V M 3 Apr V 1:37:20 Operating Battery Mains Battery Mains Battery modes: motoring motoring charging motoring motoring Triggering Events: Mams turn off User turn off turn on Mains turn on T ek 3 i r a i 0 Acqs C2 Max 700 V H V dc link (500V /div) HI C3 c i Max 1.64 V Max 24.0 A C3 Mean 1.00 A L V voltage (20V /div) LV current (20A /div) H V Phase A current (10A /div) 500 V M 1.00 s C h 2 V 400 V 3 Ap r 2003 Ch3 10: 11:37:45 Operating Mains Battery Battery Mains modes: motoring motoring motoring motoring Figure 5-18: Waveforms showing different operating modes 152

170 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Mains switched on with no battery present The waveform in Figure 5-19 shows mains charging and maintaining the LV dc link voltage. The system would stay in standby mode monitoring the user switch. Triggering Events: Mains turn on H V dc link voltage (5 0 0 V /d iv ) L V dc link v o lta g e (2 0 V /d iv ) H V phase A current (5 A /d iv ) Feb Operating modes: Ra^id charge Float charge Figure 5-19: Waveforms showing mains switch on with no battery present 5.8 Converter comparison In the converter presented in the Chapter 4, the presence of leakage energy associated with the bifilar winding was dissipated in the mains side in the RCD snubber. In order the charge at a high frequency of 16kHz, a low duty cycle charging was implemented to minimise the amount of leakage energy at which the snubber is required to capture and dissipate. The battery was therefore only charged at a rate of approximately 0.5 A at a near ultrasonic frequency. The converter in Chapter 5 presents no such problem as the freewheel paths present in the converter allows the leakage energy to be recovered. Charging was therefore implemented at 18kHz and a charging rate as high as 2.2A was achieved. 153

171 Chapter 5 Bifilar converters for dual voltage applications - Part 2 A comparison is made between the converters with snubbers presented in Chapter 4 and converters with inherent freewheel capability presented in Chapter 5 and is shown in Table Converter LV topology HV Major number of Number configuration topology additional switches of power components used diodes required used Converter with C dump buck RC Snubber 5 5 snubber - buck configuration Snubber capacitor and (1 floating + configuration inductor 4 ground) Converter with C dump buck- RC Snubber 5 6 snubber - buck- boost Snubber capacitor and (1 floating + boost configuration inductor 4 ground) configuration Converter with Shared switch Split dc Split dc 5 5 inherent asymmetric voltage capacitors (2 floating + freewheel half-bridge doubler 3 ground) capability Table 5-13: Converter comparison - components Cost-effective single-switch per phase converter has been proposed for the high voltage side while (N+l)-switch converter has been proposed for the low voltage side. So a total of five semiconductor switching devices has been used for the two-phase dual voltage converters presented in Chapter 4 and 5. It should be noted that the MOSFETs which are connected in parallel in Chapter 4 could be replaced by a single MOSFET with lower on state resistance. These MOSFETs with lower on state resistance are used in the converter presented in Chapter 5. However, the major additional components required for the snubbers in the converter proposed in Chapter 4 are eliminated in the converter proposed in Chapter 5. This is achieved with the introduction of freewheel paths for the energy associated with the leakage inductance. The only major additional components required in the converter proposed in Chapter 5 are the film type capacitors for the HV split dc link capacitors. 154

172 Chapter 5 Bifilar converters for dual voltage applications - Part 2 A comparison in the battery powered motoring mode was made and the performance of the converter presented in Chapter 5 has improved over the converter presented in Chapter 4, both in terms of output power capability and in terms of efficiency. The comparison results are shown in Table 5-14 and Figure Converter with inherent freewheel capability (Chapter 5) Converter with snubbers capacitors (Chapter 4) Charging system 4 stage charging of 2.2A, 0.5A, 0.2A and a float Single stage charging of 0.5A charge Maximum output power W W (battery motoring) Peak Efficiency 52.2% 46.6% Peak torque (battery motoring) l.onm 0.75Nm Table 5-14: Converter comparison - performance and functionality 155

173 Chapter 5 Bifilar converters for dual voltage applications - Part 2 Torque speed cu rve s fo r d iffe re n t pow er co n ve te rs _ * 2500 g 2000 g w Chapter4 converter Chapter 5 converter Torque (Nm) Input and o u tp u t p o w er fo r d iffe re n t p o w er c o n v e rte r ~ 500 r Input pow er - C h4 Output power - C h4 Input pow er - C h5 Output power - C h Torque (Nm) E fficie n cy fo r d iffe re n t pow e r c o n v e rte r LU Eff - C h4 Eff - C h Torque (Nm) 1.2 Figure 5-20: Comparison between converters presented in Chapter 4 and 5 156

174 Chapter 5 Bifilar converters for dual voltage applications - Part Conclusions Chapter 5 has presented a power converter design for the dual voltage application which is cost-effective and delivers increased performance over the converter developed in the Chapter 4. The converter combines a novel voltage doubler split dc converter for the mains converter with the cost-effective shared switch asymmetric half-bridge for the battery converter. The circuit has low passive component count and uses three ground-referenced switches and two floating switch to achieve a simple and low cost converter. The voltage doubler circuit enables the use of smaller and more reliable film capacitors which has resulted in a reduction in component cost and size. The use of smaller capacitors has in turn maximised the voltage utilisation of the ac supply and the rating of the power switches. Novel control strategies have been developed for starting, acceleration and running the motor despite the substantial modulation of dc voltage. The most significant features of the new converter developed are: the motor is fully functional as a mains motor or as a battery motor or both (the battery does not have to be charged or even connected); there are no dissipative snubbers in the circuit; only five switching devices are required; full electrical isolation is maintained between the electronics of the high voltage and the battery; a new 3stage + float rapid charge system has been implemented including the automatic detection for the absence of the battery; motor power and efficiency is slightly under specification due to the use of the bifilar windings to accommodate mains and battery windings; costly and unreliable high voltage electrolytic capacitors are not required. 157

175 Chapter 6 Bifilar converters for bipolar motor drives Chapter 6 Bifilar converter for bipolar excited motor drives 6.1 Introduction The bifilar converter, besides its capability to derive unipolar currents from dual voltage sources, is arguably the simplest and cheapest bipolar topology converter for motor drives that require bipolar excitation [40, 54, 55]. This chapter presents the bifilar converters for motors that require bipolar current excitation. Examples of motors that require bipolar excitations include induction, brushless dc and flux switching motors. As described in previous chapters, the bipolar bifilar converter offers a cost-effective solution to supply the bipolar flux required for the motor excitation. However, unlike the bifilar winding in dual voltage applications, where full isolation is required between the two coupled coils, full isolation is not required between the two coupled coils in the bipolar excited motor. A common safe path can therefore be provided for the leakage energy associated with the bifilar windings, with both coupled coils feeding a common snubber. Providing a common safe path would be cost effective as the snubber circuit is shared between the coupled coils, keeping component count to a minimum. The power converters can be classified by the way they deal with the leakage energy associated with the bifilar windings: 1. dissipative conventional RCD snubber, novel RCD snubber 2. non-dissipative - novel LCD snubber 158

176 Chapter 6 Bifilar converters for bipolar motor drives 6.2 RCD snubber RCD snubbers are commonly used with a bifilar converter to reduce the voltage overshoot caused by the leakage inductance, thereby protecting the semiconductor devices. The RCD snubber is commonly used because it uses passive components, with two diodes feeding a common resistor and capacitor, and is therefore simple. A detailed analysis of the RCD snubber was presented in Chapter 3. However, the RCD snubber suffers several drawbacks: it was dissipating heat unnecessarily; the capacitors and resistors were bulky and difficult to package; the capacitors and resistors were costly, and the cost of the removal of the heat was not insignificant. 6.3 Novel RCD snubber To overcome the problem associated with the conventional RCD snubber, much work had been concentrated in optimising the RCD snubber, both in terms of snubber efficiency and component ratings, results of which have been the subject of publications [54, 55]. An innovative RCD snubber system, shown in Figure 6-1, is proposed for a bifilar wound single-phase ac inverter or for each leg of a multi-phase inverter. The power converter consists of two ground-referenced power switches, three diodes, two capacitors and two resistors per phase. This section will demonstrate that the proposed converter achieves not only a reduction in overall power dissipation in the power converter, but also a reduction in the cost and physical size of the passive components, allowing the use of this inverter circuit on motor drives up to several kw in rating. 159

177 Chapter 6 Bifilar converters for bipolar m otor drives ^ D el D c2 Rc1 Rc2 Cc1 Cc2 A di S1 S2 Figure 6-1: Single-phase inverter for bifilar winding with novel RCD snubber 6.4 Discussion of modes of operation The principal operating modes of the circuit are depicted in Figure 6-2. The senses of the voltages shown in (a) are applied consistently throughout Figure 6-2. Some intermediate modes are not shown. In (a), SI is switched on, thereby applying positive voltage to the motor. At the same time, any remaining charge on capacitor Cel will discharge through SI, D1 and Rcl. Path (1) shows the magnetisation of coil A while path (2) shows the discharge of capacitor Cc 1 through the resistor Rc 1. During the phase demagnetisation when S1 is turned off in (b), the current in coil A is decreased by transferring the current to the coupled coil B. There are four current paths in this mode: 160

178 Chapter 6 Bifilar converters for bipolar motor drives VA foil A Coil IB) v b C VDC Cc1 Rc1 Rc2 VCC1 VRC1 VRC2 VCC2 (a) S1 turns on, current builds up in coil A, Cc1, if charged, will discharge into Rc1 Cop A i :Dci i Coil C V (6 ) Rc1 Rc2 Cc1 i k, D1 (b) S1 turns off, leakage current in coil A flows into capacitor Cc1, majority of winding current transfers to the coupled coil B through paths (2), (3) and (4) Coil PC Coil C V (6) Rc1 Rc2 Cc1 Cc2 (c) Coil A reset, current continue to flow in coil B Figure 6-2: Principal operating modes o f a bifilar wound single-phase inverter with a novel RCD snubber 161

179 Chapter 6 Bifilar converters for bipolar motor drives In path (3), current associated with the leakage inductance which does not transfer to the coupled coil B, will continue to flow in coil A into the snubber capacitor C el. The majority of the winding current in coil A will transfer to the coupled coil B through paths (4), (5) and (6). Path (4) involves the winding energy in coil A circulating through Cel, Rcl, Rc2, Cc2 and coil B. This can only happen at a rate controlled by the winding inductances since the current in coil B cannot rise instantaneously. In path (5), the voltage across capacitor Cc2 will forward bias diode D l, discharging Cc2 charged by previous turn off of S2 through D l, Rc2 and coil B and partially returning the snubber energy to the motor and dc link. In path (6), as voltage at P2 approaches zero, the anti-parallel diode of S2 will be forward biased and the bulk of the phase winding energy is coupled across the bifilar windings and current is established in the coupled winding and the anti-parallel diode of S2 to continue to flow in the coupled winding. The winding energy is then returned to the dc link, reducing the phase current to zero. Paths (3) and (6) can be described by Eq. 6-1 and Eq. 6-2 respectively while the voltage equations in loop (4) and (6) are given by Eq. 6-3 and Eq. 6-4 respectively: V c - -V A - VLA V d c = -V b + V lb Eq. 6-1 Eq. 6-2 VA + Vla + Vcci + Vrci - Vrc2 + Vcc2 + Vlb - Vb V d c = V rc2 + V cc2 + V lb - V b Eq. 6-3 Eq

180 Chapter 6 Bifilar converters for bipolar motor drives In (c), leakage energy associated with coil A has been reset and current stops to flow in coil A. The bulk of the winding energy that couples across the bifilar windings continues to force current through coil B back to the dc link in path (6). Similar operating modes occur for the magnetisation and demagnetisation of coil B. As the novel RCD snubber is fully reset each cycle, it can be used during PWM at the start-up of the motor. However it is important to note that the operation of the new snubber is different in PWM and single pulse modes. In PWM mode, coils A would repeatedly be energised and de-energised in a unipolar switching pattern in order to regulated the current in it. In single pulse mode, coils A and B will be magnetised and demagnetised in a bipolar switching pattern. The snubber operation during the PWM mode consists of: leakage energy is stored in the capacitor associated with each switch when that switch turns off. This is shown in path (1) in Figure 6-2; the same switch now turns on again and the bulk of the snubber energy is dissipated as the capacitor discharges into the switch and the snubber resistor. This is shown in path (2) in Figure 6-2. In single pulse mode the following procedure occurs : leakage energy is stored in the capacitor associated with each switch when that switch turns off. This is shown in path (1) in Figure 6-2; The energy stored in the capacitor associated with each switch is returned during the turn off of the opposite phase. This is shown in paths (4) and (5) in Figure 6-2; Figure 6-3 shows the experimental plots at turn off and show the four paths that are flowing. Figure 6-3(a) shows that path (2) and (3), both of which flow through capacitor Cc2, start to flow at turn off while Figure 6-3(b) shows that path (4) only start to flow later when the voltage at P2 approaches zero. 163

181 Chapter 6 Bifilar converters for bipolar motor drives Tek ll& n! 50.0MS/S 29 Acqs [" T ] Icci (0.5 A /d iv ) Icc2 (0.5 A /d iv ) V c c, (2 5 0 V /d iv ) V c c 2 (2 5 0 V /d iv ) 12 Jun :18:37 ( a ) : E xperim ental plots at turn o f f for n o v el R C D snubber T race 1 - current in capacitor C e l (0.5 A /d iv ), trace 2 - current in capacitor C c2 (0.5 A /d iv ), trace 3 - v o lta g e across capacitor C e l (2 5 0 V /d iv ), trace 4 - v o ltage across capacitor C c2 (2 5 0 V /d iv ) T ekae JH 100MS/S 31 Acqs t -F ] Icci ( la /d iv ) IcoiiB (la /d iv ) V ccj (2 5 0 V /d iv ) V P2 (2 5 0 V /d iv ) 13 Jun :13:33 (b) : E xperim ental plots at turn o f f for n o v el R C D snubber T race 1 - current in capacitor C e l (1 A /d iv ), trace 2 - current in coil B (la /d iv ), trace 3 - v o lta g e across capacitor C e l (2 5 0 V /d iv ), trace 4 - v o lta g e at P2 (2 5 0 V /d iv ) Figure 6-3: Experimental plots at turn off for novel RCD snubber The values of the resistor Rcl is chosen such that the capacitor Cel is fully discharged before operating mode (c). This is to avoid the capacitor from discharging 164

182 Chapter 6 Bifilar converters for bipolar motor drives dissipatively in the switches as shown in path (2) of mode (a). The same principle is applied in choosing Rc2. Since the capacitors are fully discharged at the start of each turn off operation, the voltage across the capacitor after absorbing the leakage energy is given by, Vi Cc (Vmax2) = Vi. LLi2 (J) Eq. 6-5 where Ll = leakage inductance of the bifilar winding, i = current in the winding at switch off, Vmax = voltage across the snubber capacitor after receiving leakage energy. Eq. 6.5 gives a simple solution on which the value of the capacitors, Cel and Cc2, are determined in terms of the specified parameters. The peak switch voltage is Vmax + V dc- 6.5 Comparison between conventional and novel RCD snubber Power dissipation The leakage energy stored in the capacitor of the novel RCD snubber is partially dissipated in the resistor but some energy is also recovered, as described in path (2) and (3) at turn off in Figure 6-2(b). In path (2), current circulates from one phase into the opposite bifilar phase in a torque producing sense. In path (3), some of the energy stored in capacitor from the previous pulse is flowing through the resistor and the opposite phase winding back to the dc link. On the other hand, the leakage energy captured in the conventional RCD snubber is fully dissipated in the snubber resistor. 165

183 Chapter 6 Bifilar converters for bipolar motor drives Rate o f transfer At turn off in the novel RCD snubber, current path (2) and (3) flows in the opposite bifilar windings even before the voltage at P2 in mode (b) reaches zero. Faster transfer would mean that the capacitors Cel and Cc2 would be required to capture less energy as the energy is transferred across the bifilar phase winding more quickly. However in the conventional RCD snubber, energy only transfers across the bifilar winding when VPi reaches 2 V dc and VP2 approaches zero to forward bias antiparallel diode of S2 in mode (b) Size o f capacitor The faster transfer of the novel RCD snubber would mean that less energy would be captured in the capacitors. Furthermore, the new RCD snubber configuration utilises two smaller capacitors, Cel and Cc2, to capture the leakage energy of the motor windings. These two capacitors are arranged such that each is used separately to capture the leakage energy of only one of the coupled phase windings. As a result, the capacitors in the novel RCD snubber are charged and discharged at half the frequency of the conventional RCD snubber. Another advantage of the novel RCD snubber is the capacitors are charged from zero, enabling them to store more energy for a given capacitance and maximum voltage, as given in Eq This results in the requirement of a smaller capacitor to capture the leakage energy associated with the bifilar wound coils. 166

184 Chapter 6 Bifilar converters for bipolar motor drives Dependence of snubber performance on motor speed As the capacitor in the novel RCD snubber must be designed to fully discharge at maximum motor speed, running the motor below this speed would not be detrimental as the snubber is fully reset in each cycle. The path which clamps the minimum voltage in a bifilar converter with a conventional RCD snubber, as described on page 50 in Chapter 3, does not exist in the novel RCD snubber. Since it is difficult to optimise the design of the conventional RCD snubber for a wide range of motor speeds, more dissipation occurs in the conventional RCD snubber when the motor is not running at the snubber design point. Table 6-1 summarises the advantages of the new RCD snubber over the conventional RCD snubber: Novel RCD snubber Partially dissipating, partially recovering energy Faster transfer Smaller capacitor Not speed dependent Conventional RCD Snubber Wholly dissipative Slower transfer Larger capacitor More speed dependent Table 6-1: Characteristics of novel and conventional RCD snubbers 167

185 Chapter 6 Bifilar converters for bipolar motor drives 6.6 Experimental results The two snubbers, shown in Figure 6-4, have been built and tested on a motor driving a blower in order to verify the operating principles described. The experimental results of both snubber circuits are taken at identical conditions to enable direct comparisons to be made. Both snubbers are operated at the same supply voltage of 250Vdc and with the blower running at two different speeds, one at low speed (2500 rev/min) and the other at high speed (5500 rev/min), to characterize the snubbers. Comparisons are then made on the performance of the snubbers in terms of snubber losses and capacitor voltages. Del Dc2 ;=1uF Rc1=3k3 Rc2=3h Cc1=0.1uF Cc2=0.1uF D2 D2 S2 S2 (a) Novel RCD snubber Figure 6-4: RCD snubbers (b) Conventional RCD snubber Figure 6-5 shows the experimental waveforms of current and voltage of the snubber resistor(s) at low speed while Figure 6-6 shows the experimental waveforms at high speed. Figure 6-7 shows the voltage of the snubber capacitor(s) at high speed. 168

186 Chapter 6 Bifilar converters for bipolar motor drives Tek E m m looks/s 7 Acqs I T ] UE C2 Max 80mA C3 Max 260 V M 1 Mean 2.86 W ML C1 Freq Hz Low signal am plitude 5.00 V CTU 1OOmAO Ml.OOms Ch1 f 250 V M athl 10.0 W 1.00m s (a): Experim ental results for n o v el R C D snubber at r/m in T race 1 - encoder signal (5 V /d iv ), trace 2 - current in resistor R c l (O.lA /d iv ) trace 3 - v o lta g e across R c l (2 5 0 V /d iv ), trace M l - pow er dissipation in R c l (lo W /d iv) M axim u m v o lta g e across R c l = V, M ean p ow er across R c l = W Tek 5.00MS/S 3 Acqs T- - IB C2 Max 114mA C3 Max 310 V M 1 Mean W Cl Freq HZ Low signal am plitude Ml' S.oo V Ch2 So.omAO M i.o o m s Chi / 250 V M athl 1.00ms 10.0 w 12 Sep :20:24 (b): E xperim ental results for conventional R C D snubber at r/m in T race 1 - en coder signal (5 V /d iv ), trace 2 - current in snubber resistor (0.0 5 A /d iv ), trace 3 - v o lta g e across snubber resistor (2 5 0 V /d iv ), trace M l - p ow er dissipation in snubber resistor (lo W /d iv) M axim u m v o lta g e across snubber resistor = 3 10V, M ean p ow er across snubber resistor = W Figure 6-5: Experimental results at 2500 r/min 169

187 Chapter 6 Bifilar converters for bipolar motor drives Tek HUIM 10.0MS/S ^ 1 Acqs C2 Max 132mA C3 Max 420 V Ml Mean 4.60 W ML Cl Freq Hz Low signal amplitude Chi S.'M V BBS Ch3 250 V Mathl 25.0 W 16'OmAri M SOOjis Ch1 J 500ps 12 ju n :59:00 (a): E xperim ental results for n o v el R C D snubber at 5500 r/m in T race 1 - encoder signal (5 V /d iv ), trace 2 - current in resistor R c l (O.lA /d iv ) trace 3 - v o lta g e across R c l (2 5 0 V /d iv ), trace M l - pow er dissipated in R c l (2 5 W /d iv ) M axim u m v o lta g e across R c l = V, M ean pow er across R c l = W Tek HMBI 1O.OMS/s 2 Acqs [ T ] C2 Max 90mA C3 Max 290 V Ml Mean W Ml' Cl Freq Hz Low signal amplitude " C f m o r v SO.OmAO ' M 500'iis CHI f Ch3 250 V Mathl 10.0 W 500us 12 ju n :22:10 (b): E xperim ental results for conventional R C D snubber at r/m in T race 1 - en coder signal (5 V /d iv ), trace 2 - current in snubber resistor (0.0 5 A /d iv ), trace 3 - v o lta g e across snubber resistor (2 5 0 V /d iv ), trace M 1 - p ow er dissipated in snubber resistor (lo W /d iv) M a x im u m v o lta g e across snubber resistor = V, M ean p ow er across snubber resistor = W Figure 6-6: Experimental results at 5500 r/min 170

188 Chapter 6 Bifilar converters for bipolar motor drives TeKHBlIM 10.0MS/S ^ 33 Acqs C3 Max 465 V C4 Max 475 V C3 Min 5 V C4 Min 5 V Ch3 250 V M 500MS Ch3 \ 280 V 12:05:48 (a): E xperim ental results sh ow in g snubber capacitor voltages in n ovel R C D snubber T race 1 - gate sig n a ls to SI (5 V /d iv ), trace 2 - voltage across capacitor C e l (2 5 0 V /d iv ), trace 3 - voltage across capacitor C e l (2 5 0 V /d iv ) M ax v o lta g e across snubber capacitor = V, m in volta g e across snubber capacitor = 5 V TeKSErai 1o.oms/ s 24 Acqs C3 Max 535 V C3 Min 440 V Chi n je 5.00 V 250 V to soojis Chi X 13 Jun :08:18 (b) : E xperim ental results sh o w in g snubber capacitor v o lta g es in conven tional R C D snubber Trace 1 - gate signals to S I (5 V /d iv ), trace 2 - voltage across snubber capacitor (2 5 0 V /d iv ) M ax v o lta g e across snubber capacitor = V, m in v o ltage across snubber capacitor = V Figure 6-7: Experimental results showing snubber capacitor voltages at 5500 r/min 171

189 Chapter 6 Bifilar converters for bipolar motor drives Table 6-2 summarises the main differences observed from the experimental results. The losses in new RCD snubber is 42.0% of the losses in conventional RCD snubber at low speed and 50.3% at high speed. As described in previous section, the higher losses in conventional RCD snubber at lower speed is due to the additional losses incurred when the snubber is not running at its desired speed. Novel RCD snubber Cel = Cc2 = O.lpF Rcl = Rc2 = 3k3 Conventional RCD snubber C = lpf R = 3k3 Max capacitor voltage 475V [Figure 6-7(a)] 535V [Figure 6-7 (b)] Min capacitor voltage 5 V [Figure 6-7(a)] 440V [Figure 6-7 (b)] Total losses in snubber 5.72W [Figure 6-5(a)] 13.62W [Figure 6-5(a)] resistor(s) at low speed (2500 rev/min) Total losses in snubber 9.2W [Figure 6-6(a)] 18.3W [Figure 6-6(b)] resistor(s) at high speed (5500 rev/min) Table 6-2: Experimental results o f novel and conventional RCD snubbers 172

190 Chapter 6 Bifilar converters for bipolar motor drives Table 6-3 compares the cost and component count of the novel and conventional RCD snubber systems. The use of the new snubber has resulted in the use of physically smaller resistors and capacitors. Even though the component count is higher, the overall cost of the new snubber is about 45.4% of the cost of components used in the conventional snubber for the same test motor. Description Units required Total cost Conventional RCD snubber 0.47pF 1000V capacitor UF 5408 diode W 6k8 resistor Total Novel RCD snubber 0.1 pf 1000V capacitor UF 5408 diode W 3k3 resistor Total Savings (not including additional saving through smaller heatsink) 5.32 Note : Prices are taken from 1 off prices in UK and are therefore only intended as a guide. Table 6-3: Component rating and cost comparison between novel and conventional RCD snubbers Heat removal represents a significant cost in small motor drives and the reduction in power dissipation achieved with the novel RCD snubber reduces the cooling requirement. This will lead to a further cost reduction over that estimated in Table

191 Chapter 6 Bifilar converters for bipolar motor drives 6.7 Summary of novel RCD snubber This section has presented a novel RCD snubber for bifilar wound single-phase inverter driven motors. It is applicable to motors that require bipolar excitation up to several kw in power. The new snubber can be extended to the bifilar wound threephase induction motor proposed in [40] to achieve further reduction in drive cost. The novel topology not only preserves the simple and low cost control features of a conventional RCD snubber, but also improves it by partially recovering the snubber energy and reducing the costly and bulky components of a conventional RCD snubber. The overall circuit is also more cost effective than a full-bridge single-phase inverter. 6.8 Novel LCD snubber The novel RCD snubber presented in this chapter, despite its attractiveness of simplicity and low cost, is still dissipative. This not only reduces the efficiency of the system, but the dissipated heat also needs to be extracted and removed from the system. The cooling requirements represent a cost that could be further saved by recovering the leakage energy, which leads to a converter with non-dissipative snubber presented in this section. High frequency PWM is normally used in power converter to control the motor current at low speed. The power dissipation in RCD snubber increases with switching frequency and this reduces the efficiency of a high frequency switching converter. To remedy this drawback, a nondissipative snubber, composed of a capacitor and a inductor, is used. The proposed LCD (inductor, capacitor and diode) snubber has the advantage over the RCD snubber in that the power dissipation by snubber resistor is eliminated. 174

192 Chapter 6 Bifilar converters for bipolar motor drives 6.9 Description of the LCD snubber Figure 6-8 shows the novel LCD snubber proposed for a bifilar wound single-phase ac inverter or for each leg of a multi-phase inverter. The circuit uses an inductor Lc 1 to recover the energy captured in capacitors Cel or Cc2 during turn-off. This is a very simple, easily controlled and cheap topology which is based on an energy recovery snubber presented in [56, 57] but has been adapted for use in a bipolar bifilar converter. Del Dc2 Cc1 Dc3 Dc4 Cc2 Let IGBT1 IGBT2 Figure 6-8: Single-phase inverter for bifilar winding with novel LCD snubber 6.10 Discussion of modes of operation Figure 6-9 shows the principal operating modes of the LCD snubber. The positive and negative voltage loops of the LCD snubber are similar to the novel RCD snubber in that current is drawn and returned to the source voltage V dc- However, the difference between the two converter lies in the treatment of the leakage energy. 175

193 Chapter 6 Bifilar converters for bipolar motor drives ^ D el Dc2 Coil B Cc1 Dc3 Dc4 Cc2 C VDC veal Vcc2 P2 IGBT1 IGBT2 (a) S1 turns on, current builds up in coil A, Cc1, if charged, will discharge resonantly into the inductor ~ Del Dc2 IGBT1 IGBT2 (b) S1 turns off, leakage current in coil A flows into capacitor C c1, majority of winding current transfers to the coupled coil B through paths (4) and (5) Del Dc2 Cc1 Dc3 Dc4 C c2 C2 P1 : IGBT1 IGBT2 (c) Coil A reset, current continue to flow in coil B Figure 6-9: Principal operating modes o f a bifilar wound single-phase inverter with a novel LC snubber 176

194 Chapter 6 Bifilar converters for bipolar motor drives In (a), coil A is in a positive voltage loop when SI is turned on, as shown in path (1). At the same time, path (2) shows that any residual voltage on the snubber capacitor transfers resonantly to the snubber inductor and the snubber capacitor is inversely charged to be ready to absorb the bifilar leakage energy for the turn-off transition. (b) shows the demagnetisation process of coil A. When switch SI is turned off, the current in coil A is decreased by transferring the current to the coupled coil B. There are three current paths in this mode: In path (3), current associated with the leakage inductance which does not transfer to the coupled coil B, will continue to flow in coil A into the snubber capacitor Cel. As the voltage Vcel increases and Vce2 decreases, the energy captured in Cc2 during the previous turn-off is returned through the inductor to the power supply via Dc4 and coil B, as shown in path (4). The use of an inductor has the advantage over RCD snubbers in that the power dissipation in the snubber resistor is eliminated. Furthermore, some of the snubber energy is also returned to the motor as the current in coil B is flowing in a torque producing sense, the remaining snubber energy is returned to the source nondissipatively through an inductor. When the voltage at P2 approaches zero, the anti-parallel diode of S2 will be forward biased and the bulk of the phase winding energy would be returned to the power source in the conventional manner involved in all bifilar converters, as shown in path (5). Paths (3) and (5) can be described by Eq. 6-1 and Eq. 6-2 respectively while the voltage equation in loop (4) is given by Eq Vdc = Vlc + Vcc2 - Vb Eq

195 Chapter 6 Bifilar converters for bipolar motor drives In (c), leakage energy associated with coil A has been reset and current stops to flow in coil A. The bulk of the winding energy that couples across the bifilar windings continues to force current through coil B back to the dc link. Similar operating modes occur for the magnetisation and demagnetisation of coil B. As in the novel RCD snubber, the LCD snubber allows repetive switching of each switch and can therefore be used during PWM at the start-up of the motor. The unipolar switching pattern during PWM mode and bipolar switching pattern during single pulse mode result in different snubber operation. In PWM mode the snubber operation is: leakage energy is stored in the capacitor associated with each switch when that switch turns off. This is shown in path (1) in Figure 6-9; the same switch now turns on again and the energy in the snubber capacitor is transferred resonantly to the snubber inductor. This is shown in path (2) in Figure 6-9. In single pulse mode the following procedure occurs : leakage energy is stored in the capacitor associated with each switch when that switch turns off. This is shown in path (1) in Figure 6-9; the bulk of the energy stored in the capacitor from the first switch off is recovered when the second switch turns off. This is shown in path (4) in Figure 6-9. Figure 6-10 shows the experimental plots of the snubber capacitor, Cel, and snubber inductor, Lcl, in single pulse mode. Snubber capacitor Cel is charged positively when it absorbs the leakage current during phase demagnetisation coil A. The energy in the snubber capacitor is recovered resonantly through the inductor during the demagnetisation of the opposite coil, coil B. There is some recharging of the snubber capacitor C el during armature current dead-time but any remaining charge in the 178

196 Chapter 6 Bifilar converters for bipolar motor drives snubber capacitor is transferred to the inductor during subsequent phase magnetisation of coil A. T e k a f c i a 5.00M S/S [ T 30 A cqs Cl Freq Hz Low signal a m p litu d e Ic c l(1 0 A /d iv ) EE Coil A demagnetisation C3 Max V V c c l(2 5 0 V /d iv ) C4 Max 14.8 A Coil A magnetisation CTTT C h3 S. 00 V V H fe 10.0 AQ Coil B demagnetisation M 2 0 O ps C h i J 2.5 V i s j Ut :4 2 :4 2 (a) E xperim ental p lo ts sh o w in g snubber capacitor current and voltage T race 1 - P osition sensor 5V /d iv Trace 2 - C apacitor C e l current lo A /div T race 3 - V o lta g e across capacitor C e l V /d iv T e k M M W M S /S 18 A cqs [ T... C l Freq, k H z I c c l(1 0 A /d iv ) Coil B demagnetisation C4 Max 12.0 A C3 Max 172 V V c c l(2 5 0 V /d iv ) C3 M in V Coil B magnetisation (b) C h Afi M 2 OOps C h 1 I 2.5 V 16 j u n :0 2 :4 2 E xperim ental p lots sh o w in g snubber inductor current and v o ltage T race 1 - P osition sensor 5V /d iv T race 2 - Inductor L c l current 10A /d iv T race 3 - V o lta g e across Inductor L c l V /d iv Figure 6-10: Experimental plots showing the LCD snubber operation in single pulse mode 179

197 Chapter 6 Bifilar converters for bipolar motor drives 6.11 Implementation A novel LCD snubber was constructed to replace the conventional RCD snubber used in a power tool application. Although the existing RCD snubber in the power tool application represents 1% of the total power handled by the power converter, the size of the RCD snubber is not insignificant [39]. An LCD snubber has been built for the bifilar converter to allow it to be tested against the conventional RCD snubber Sizing the snubber components The size of the capacitor was based on an estimate of the amount of leakage energy which had to be absorbed and an acceptable overvoltage. The inductor was then chosen to give a resonant frequency of 100 khz between the inductor and the capacitor. This meant that the reset time of the snubber would be 5ps, allowing maximum flexibility in the choice of PWM switching frequency. For the leakage inductance of the power tool drive, the component values chosen for the LCD snubber were, Lcl = 25 ph (air cored), Cel = Cc2 = loonf. The components values used for the existing RCD snubber were R = 3k3 and C = lpf. 180

198 Chapter 6 Bifilar converters for bipolar motor drives 6.12 Experimental results Load test comparison between LCD and RCD snubbers The LCD and RCD snubbers were compared on the power tool drive and the experimental waveforms under different operating load points are shown in Figure LCD Snubber C h 1 : P o sitio n sen so r 5 V /d iv C h 2 : Arm ature w in d in g current 10A /d iv C h3 : IG B T 1 colecto r em itter v o lta g e 250V/<jliv RCD Snubber Ch 1 : P osition sensor 5V /d iv Ch2 : Arm ature w in d in g current 10A /d iv Ch3 : Snubber v o lta g e V /d h C h 4 : F ield w in d in g current loa/c 782 Acqs [ f it a : 832ns : m s C2 Max 19.2mV Ch4 : F ield w in d in g current 10; iv A :8 2 4 n s : 824ns C2 Min -19.2m v C2 Min -18.8m V C3 Max 750 V C3 Max 760 V C3 Min 480 V C4 Max 6.8 A C4 Max 6.8 A Chi S.00 V Ch2 io.omv to n s Chi f 2.5 V isjun 2000 Ch3 250 V S ib 10.0 Afi 12:06:43 chi S.w v aasj lo.omv v Ch3 250 V CR AO 200ns chi f 2.7 V 14 Jun :47:51 Torque = 0.0 Nm, Input power = 835.2W, Speed = rpm, Vce(max) = 760V Torque = 0.0 Nm, Input power = 887.0W, Speed = rpm, Max snubber capacitor voltage = 750V 181

199 Chapter 6 Bifilar converters for bipolar motor drives T e F K S /S A: ms 1.704m d C2 Max I9.6 m v TeF K 5 /S 1341 Acqs t A : m s C2 Min 19.2mV C2 Min 20.4mV C3 Max V C3 Max 840 V C3 Min V C4 Max 8.0 A C4 Max 8.0 A Chi S.'flffV Chi 1O mv 4M 200m s Ch1 Ch3 250 V H Q 10.0 AQ 2.5 V 15 jun :0 1 :1 1 _ lo.omv M 200M S Ch1 I 2.7 V 14 u n AQ 09:50;54 Torque = 0.2 Nm, Input power =1150.5W, Speed = rpm HekHMHI 250k5A 14 \7 Acqs a : ms C2 Max 20.4mv Torque = 0.2 Nm, Input power =1139.8W, Speed = rpm "TeraEJiTTTnwr 1408 Acqs - Hf A : 872M S : 1.740ms C2 Min 21.2mv C2 Min 20.4mv C3 Max V C3 Max 920 V C3 Min V C4 Max 9.2 A C4 Max A cm sins v Ch3 250 V "TiTffmV W 706'm s Chi f 2.? V isjun Afi 1 3 :0 5 :1 3 Chi 5.ooV io.om v m 2 o o m s Chi f 2.7 v 14 i Un 2000 Ch3 250 V C M 10.0 AQ Torque = 0.4 Nm, Input power =1542.9W, Speed = rpm Torque : 0.4 Nm, Input power =1542.8W, Speed = rpm Acqs 1255 Acqs r t: T J 904MS 5.396m s C2 Max 20.8mV C2 Min -20.8rnV 1-» 1 1 A 1 11:4 / : 1 ; 1 / \ / / \ 1 / \ I C3 Max 980 V \ /!. i \. v / I- /.. [ j A: ms : 2.700m s C2 Min mv C3 Max 840 V C3 Min 400 V C4 Max 11.2 A 1 v ( 1 C4 Max 11.2 A 3-> : 1 1 l.. ;. i i Ch3 250 V C h AQ Ch3 250 V T O m V M m s Chi f 2.7 V i 4 j u n 2000 Torque = 0.6 Nm, Input power =1912.0W, Speed = rpm Torque = 0.6 Nm, Input power =1924.2W, Speed = rpm 182

200 Chapter 6 Bifilar converters for bipolar motor drives l e k flf U H l k 5 / s 1479 Acqs T e lt 25ok57s A c q s A : u s -4us C2 Max 22.8m V a : ms 5.572m s C2 Min 22.4m v C2 M n 22.4mv C3 Max 860 V C3 Max 980 V C3 Min 380 V C4 Max 13.2 A C4 Max 13.2 A 10 OmV M m s Ch1 f 2.5 V I 5 u n 2000 W 10.0 AQ 13;12;3] 5 00 V 250 V 16.OmV to 200ms Ch 1 f 2.7 V 14 ju n 2000 CR AQ 10:03:43 Torque = 0.8 Nm, Input power =2283W, Speed = rpm l e k l f l M «J 5 0 k 5 / s 1447 Acqs Torque = 0.8 Nm, Input power =2294W, Speed = rpm l e k L im w 2 5 o k S 7 s 1588 Acas A: 9 6 0ns : m s A: 968ms 6.768m s C2 Max 23.2mV C2 Min 22.4m v C2 Min mv C3 Max 820 V C3 Max 880 V C3 Min 360 V C4 Max 15.2 A C4 Max 15.6 A Chi Ch V 250 V 10.OmV to p s Ch1 f 2.5 v IS J u n A f i 13:16:35 cfvi 5Tffi Ch3 250 V Id.ttmV to m s Ch 1 f 2.7 V 14 j u n AQ 1n-no-ns Torque = 1.0 Nm, Input power =2641W, Speed = rpm leki.imbi256ks/s Acqs" A: 1.008m m s C2 Max 21.2mV Torque = 1.0 Nm, Input power =2653W, Speed = rpm lakumbi 256k5/s Acqs t I! A: 1.020m s ms C2 Min 2 3.6m v C2 Min m v C3 Max 830 V C3 Max 760 V C3 Min 360 V C4 Max 17.2 A C4 Max 17.2 A C h 1 Ch V 250 V Fi4 io.o m v 10.0 AQ ivi 2oojis C hi f 2.5 V i s j u n :22:58 ciir s.iro Ch3 250 V C 10.OmV M ms Ch1 / 2.7 v 14 Ju n :42:00 Torque = 1.2 Nm, Input power =2985W, Speed = rpm Torque =1.2 Nm, Input power =2910W, Speed = rpm 183

201 Chapter 6 Bifilar converters for bipolar motor drives H T ek a u g i 2 5 0k5A 1081 Acqs TeT C flfiiai ^t>ttes7s ^ 1641 A c^s a : 1.088m s <S: m s A: 1.108m s : 5.544m s C2 Max m v C2 Min m v C2 Min m V C3 Max 930 V C3 Max 910 V C3 Min 330 V C4 Max 17.6 A C4 Max 18.0 A Chi 5 66 V 10.OmV M 20 Ops Chi f 2.5 V i s i u n 2000 Ch3 250 V CR4 10.0AC C h r Ch3 S.flO V 250 V IB lo.omv M 20C1(jls C hi / 3.1 V i 4 j u n AQ 10:50:50 Torque = 1.4 Nm, Input power =3105W, Speed = rpm Iek tmmill 1048 Acas Torque = 1.4 Nm, Input power =3022W, Speed = rpm ISk aisia 250kSA Ac^s A 1.180m s 3>: ms C2 Max 32.0mV A : 1.200ms 8.388m s C2 Min m v C2 Min -32.0m V C3 Max 1.040kV r C3 Max 1.040kv C3 Min 310 V C4 Max 18.4 A C4 Max 18.8 A rrn s.o o v Ch3 250 V 10.0 mv M 20 0 n s C hi / 2.5 V i 5 j u n Afi 14:00:48 C hi Ch V 250 V 16/OmV M 200ms CHT f T T V 14 iu n AQ n :0 1 ;1 1 Torque = 1.6 Nm, Input power =3230W, Speed = rpm Torque = 1.6 Nm, Input power =3193W, Speed rpm Figure 6-11: Load test comparison between LCD and RCD snubbers The comparison between the performance of the LCD and RCD snubbers are shown in Figure 6-12, Figure 6-13, Figure 6-14 and Figure It can be observed that the LCD snubber matched the performance of the conventional RCD snubber despite using much smaller components. The efficiency of the LCD snubber outperforms the RCD snubber at full pulse near the maximum power output. The efficiency of the LCD snubber is lower at lower power levels because there is some recharging of the snubber capacitor during armature current dead-time. 184

202 Chapter 6 Bifilar converters for bipolar motor drives y y LCD Snubber Q RCD Snubber Efficiency, % Speed, rpm Dotted lines. Indicate -Power out Torque, Nm Figure 6-12: Output torque against motor speed curve V LCD Snubber - RCD Snubber Torque, Nm Figure 6-13: Output torque against efficiency curve 185

203 Chapter 6 Bifilar converters for bipolar motor drives Power out, W Power in, V LCD Snubber Q RCD Snubber J I-...! Torque, Nm Figure 6-14: Output torque against input power curve - y LCD Snubber - RCD Snubber Torque, Nm Figure 6-15: Output torque against output power curve 186

204 Chapter 6 Bifilar converters for bipolar motor drives Dynamic performance of LCD Snubber Start-up performance Figure 6-16 shows the start-up performance using the LCD snubber with small capacitors. Two blanking pulses are used to kick start the motor while maintaining the snubber voltage below 1 kv. Tek Run: 25.0MS/S Sample Q0U3E C h i : V olta g e across C V /d iv 950 V Ch2 : Arm ature current 2 0 A /d iv (2 A /m V ) Ch3 : Snubber voltage 250 V /d iv Ch Afl 19 Jul :49:25 C h4 : F ield current 10A /d iv Figure 6-16: Snubber performance during start-up Acceleration performance Figure 6-17 shows the acceleration performance using the LCD snubber with small capacitors. The system was able to meet the acceleration specification of the power tool without causing overvoltage despite using a much smaller capacitance. 187

205 Chapter 6 Bifilar converters for bipolar motor drives T ek Run: l o o k s /s A : m 0 s C h i : Position sensor 5V /div C2 Min m V Ch2 : Armature Current 20A /div (2 A/m V ) Ch3 : Snubber voltage 250V /div 21)u l :1 5 :2 4 M a x sn u b b er v o lta g e = V Figure 6-17: Snubber performance during acceleration 6.13 Com parison between LCD and RCD snubbers Figure 6-18 shows a photograph comparing the size of the components in the novel RCD and LCD snubbers while Table 6-4 lists the cost of the components for the novel RCD and LCD snubbers New Snubber 0.1 pf 1000V capacitors UF V 3A diodes 25mH inductor RC Snubber 0.47pF 1000V capacitors 17W 3k3 ceramic resistors Figure 6-18: Photograph comparing the size of the components used in the RC and proposed snubber circuits 188