EEC132B Winter Final Project: To Be Handed in by End of Instruction: Monday March 19

Transcription

1 EEC132B Winter 2012 Final Project: To Be Handed in by End of Instruction: Monday March 19 Objective: Design of the passive circuitry associated with a balanced Schottky diode microstrip mixer. References: 1. H.P. Application Note #976: ``Broadband Microstrip Mixer Design -The Butterfly Mixer.'' 2. H.P. Application Note #963: ``Impedance Matching Techniques for Mixers and Detectors.'' 3. H.P. Application Note #154: ``S-Parameter Design.'' 4. H.A. Atwater, ``The Design of the Radial Line Stub: A Useful Microstrip Circuit Element,'' Microwave Journal, 28, 149 (1985). 5. W. Alan Davis, ``Microwave Semiconductor Circuit Design,'' Chapts. 3, 8 and S. Maas, ``Microwave Mixers.'' 7. T.C. Edwards, ``Foundations for Microstrip Circuit Design.'' 8. D.Lacombe and J. Cohen, ``Octave-Band Microstrip DC Blocks,'' pg.555, IEEE MTT, August G.D. Vendelin, A.M. Paviog and V.L. Rohde, ``Microwave Circuit Design,'' John Wiley, Chapt E.L. Kolberg in Chapt 2 of Vol. 2, ``Handbook of Microwave and Optical Components.'' ``Motivation and Background:'' At the heart of the majority of receiver systems is the mixer. This consists of a nonlinear element, commonly a Schottky diode, in which the signal (typically of very low level) at frequency s is combined with a strong local oscillator of frequency L which differs in frequency from s by the socalled intermediate frequency IF. Due to the diode nonlinearity, a signal is generated at IF which is amplified in an IF amplifier to a sufficient level for subsequent signal processing. Figure 1 below shows schematics of typical receiver systems.

2 ANT RF bandpass filter LNA Image Reject Filter Down Converter LO IF bandpass filter IF LO lowpass filter Voltage Control Oscillator System Specifications: (1) Operation frequency range: GHz to 14.5 GHz (2) Receiver gain: >18 db (3) Receiver noise figure: <3dB (4) IF frequency: 2.25 GHz (5) Bias voltage: +/- 12V Functional Block specifications: (1) Low noise amplifier Input frequency range: GHz to 14.55G Hz Small signal gain: >25 db Gain flatness: +/-0.25 db Noise Figure: < 2 db Output IP3: 10 dbm Input return loss: < -12dB (50 Ohm)

3 (2) Down Converter Mixer Input frequency range: GHz to 14.5GHz Conversion Loss: <6dB Input IP3: 12 dbm LO frequency: 11.5 GHz to GHz LO driving power: 10 dbm LO/RF Isolation: >30 db IF frequency: 2.25 GHz (3) Voltage Control Oscillator Frequency: 11.5 to GHz Output Power: 3 dbm Phase noise off carrier: 10 khz: 60 dbc/hz Figure 1a Ku-Band Satellite Receiver LNC. Figure 1b Sketch of a receiver used in anti-collision radar

4 Figure 1c 180 o 3-dB hybrid balanced mixer with microstrip stepped-impedance LPF Receivers: Superheterodyne Mixing In the course of this design, we would like to accomplish the following.

5 1. Design an impedance matching circuit for the diode (specifications provided) and implement the matching circuit in a balanced mixer configuration utilizing the rat-race coupler as a diplexing element. 2. Create an external IF matching circuit to properly match the IF output. 3. Design the DC blocks 4. Design the bias decoupling network 5. Design the image reject filter 6. Provide the layout for a complete microstrip realization of the mixer. The equivalent circuit for a Schottky diode is shown below. Components external to the dotted line result from the package around the diode chips and these do not vary. However, the components internal to the dotted line result from the physical geometry of the diode. These parameters can and do change as a function of bias voltage and incident RF power. Figure 2

6 To understand the mixer let us focus on a single diode (single-ended) mixer as shown below in Fig. 3. Note that before entering the mixing element the signal and LO must be combined together. This is called diplexing and can involve filters, hybrids, or simply a standard directional coupler. Figure 4 shows some typical configurations. Figure 3 Figure 4. Examples of single-ended mixer configurations: (a) local oscillator injected through a broadband directional coupler; (b) local oscillator injected via a narrow-band diplexer (e.g., a ring filter). From the typical values listed, you can make reasonable approximations to simplify the matching network design. The first choice is to neglect everything except for R j and C j. The difference in normalized diode impedance due to neglecting these is ~ 4 % as shown below, with the diode operating at I d = 1 ma at 10 GHz. Z d io d e j2.2 0 F u ll M o d e l

7 Z j C a n d R o n ly d io d e j j Another point to keep in mind is that the diode impedance will move along constant reactance circles with frequency and constant resistance circles with bias current and/or RF drive power (as long as we are operating in the small signal region). The simplification in circuit model will allow us to predict the diode admittance reasonably well. Unfortunately, since the diode is a nonlinear device this model will not predict exactly the input impedance with frequency and drive current. The diodes junction capacitance and junction resistance vary considerably over the period of one cycle. The junction capacitance can vary a factor of ~ 4-5 while the R j term can vary several orders of magnitude. A rough analysis of these effects appears in Appendix A. It is this effect that causes the mixing products to appear when two signal sources of different frequency are incident on the diode. The appearance of mixing products, which are signals at frequencies equal to the sum and difference of the RF and LO sources, cause the impedance to vary from the value calculated above. Figure 5 below shows the case for a large LO pump (but small rf signal). The situation becomes ever more complicated for large rf signal levels where so-called intermodulation products become important. Figure 5. Modulation spectra for a pumped nonlinear element allowing for both positive and negative frequencies Consider a diode with RF and LO sources applied to it, generating a sum frequency (RF+LO) and a difference frequency (RF-LO). The power reflected from the diode will contain four frequency components reflected from four different terminations, (Remember that the impedance seen is a function of frequency).

8 These different terminations are called embedding impedances. You can immediately see that this quickly complicates the task of creating a matching circuit. Actually a diode generates an infinite sum of sum and difference frequencies making this task seemingly impossible. However, the junction capacitance quickly dissipates the higher order frequencies. A method that exists to deal with a finite set of embedding impedances is harmonic balance. A description of this procedure is found in several publications including Maas' book referenced at the beginning of this design project write-up. Figure 6 illustrates the model. Since this is a very detailed procedure, it is beyond the scope of this course and we will use the approximation of matching to R j and C j only. However, more information is provided in Appendix A of this write-up. Figure 6. Equivalent circuit for mixer LO analysis with the large-signal diode model characterized in the time domain and with the series resistance, R s and embedding network Z e represented in the frequency domain. One can use the model described in Appendix A to estimate the input impedance of the diode at 8, ,..., 11.5, 12 GHz, using three bias currents of 0.5 ma, 0.75 ma and 1.0 ma. However, here we are providing you with the diode parameters. Another area of importance involves the noise which the mixer, local oscillator and IF amplifier can add to the signal (see Fig. 7). One defines the noise figure as the ratio of the signal-to-noise (S/N) ratio at room temperature at the signal input to the mixer to the S/N ratio at the output of the IF amplifier. Noise figure is the noise factor expressed in db.

9 Figure 7 The mixer diode has noise contributions due to shot noise, series resistance thermal noise and, sometimes, hot electron noise (if pumped sufficiently hard that intervally scattering occurs). 2. Measurement of S 11 of the diode The measurement of characterization of the diode is done in a special test fixture that allows one to calibrate the network analyzer at the device. This implies that the reference phase can be set inside the fixture and not at a connector. This allows us to make a more accurate measurement of device parameters. Next quarter, we will look at device characterization; here, we will simply use the manufacturer s specifications. 3. Mixer Configurations Although it is quite simple, the single-ended mixer suffers from a number of limitations including losses in the diplexing coupler, sensitivity to amplitude variations in LO level (AM noise) and reflections of LO power into the signal port. A solution is to employ a balanced mixer containing two diodes, which are driven in opposite phase. This is the configuration we will employ in our design project. In this case, the reflected LO power cancels, but the IF outputs add if the diodes are reversed. Figure 9 shows a comparison between singled-ended and balanced mixer configurations.

10 Figure 8. Common mixer topologies (a) single-ended; (b) single-balanced; (c) double balanced. Obviously, the LO power requirements for the single balanced mixers are doubled over the single diode case. Other interesting features include the fact that the noise figure of the single balanced mixer is reduced over the single diode case due to the fact that for well matched diodes the AM noise from the LO at the signal frequency cancels at the IF output. The double balanced mixer requires yet more LO power, but exhibits the best large signal handling capability as well as port-of-port isolation and spurious rejection. For this design project, we will design and fabricate a single balanced mixer, which is shown schematically in Fig. 9.

11 Figure 9. Single balanced mixer configurations: (a) the phase shifts of 90 o and 180 o hybrids; (b) schematic balanced mixer configurations; (c) equivalent circuit of the 180 o hybrid mixer. Examples of the 3-dB 90 o couplers are the branch-line coupler and the Lange coupler shown below in Fig. 10. Examples of the 180 o hybrid are the waveguide magic-t and the rat-race coupler. Figure 10. Microstrip Lange coupler configurations; (a) four-strip; (b) six-strip; (c) eight-strip.

12 An ideal balanced mixer needs to be able to route the RF and LO signals to two diodes and retrieve the desired IF signal. In addition, a DC bias is needed in some cases to drive the diodes to the proper operating point. The ideal mixer will allow these four signals, RF, IF, LO, and DC, to appear together in the mixer circuit and separately at their respective ports. The way we will configure the mixer in this design project is shown below in Fig. 11. A rat-race coupler is employed to combine the RF and LO at ports 1 and 4, respectively, to diode #1 and diode #2. Note that bias enters at diode #1 and exits through diode #2. A bias decoupling network at diode #1 allows only the DC to pass through, while appearing as a short at the RF, LO, and IF frequencies. This network consists of discrete capacitors and inductors that you can specify or the network can be made of distributed transmission line elements. DC blocks also appear at the LO, RF, IF ports. Also note that the IF is coupled out at a point in the rat race where both the RF and LO have reasonably high isolation. In addition, a low pass filter using quarter wavelengths of transmission line is employed. Higher LO/IF, RF/IF rejection could be obtained with a Chebyschev type low pass filter and you may choose this approach. Now the system has DC appearing in the mixer circuit but only at the bias port, not at the RF, LO, and IF. Likewise, only the IF appears at the IF power and not at the DC port. Proper choice of DC blocking capacitors will give a reasonable amount of rejection of the IF at the LO and RF ports. The nature of the rat race allows the RF to be rejected at the LO port and the LO to be rejected at the RF port. Note that the circuit shown in Fig. 11 does not include the diode rf matching circuitry or IF matching that you will include with your mixer design. Figure 11. Microstrip hybrid ring (rat race) mixer with dc bias. As you will recall from the discussions in class and in Pozar, the rat-race coupler has relatively narrow bandwidth. This can be improved by replacing 0.5 wavelengths of transmission line between ports 2 and 4 of the original design (Fig. 12) by a constant phase change of 180 o (see Fig. 13). The difficulty is that the transmission lines must be quite closely coupled. One of the common approaches is to use Lange couplers for this function.

13 Figure 12. Ring or ``ratrace'' hybrid circuit topology for microstrip or stripline applications. Figure 13. Circuit topology for broadband ring hybrid employing an inversely connected transmission line in order to achieve an additional 180 o phase``flip.'' Even though you are not asked to build one, it is appropriate to briefly mention the doubly balanced mixer. Here, by combining four diodes in a ring, bridge, or star it is possible to cancel the LO reflections and noise at both the signal and IF ports. This has the advantage that no filtering is required at the IF port but has the disadvantage that very broadband baluns or transformers must be employed. Figure 14 below shows a schematic representation.

14 Figure 14 Figure 15 shows a schematic taken from Maas of a microstrip realization of a balanced mixer employing baluns. Figure 15 Doubly balanced mixer using microstrip baluns. The capacitors prevent the IF from being short-circuited to ground through the microstrip lines. The second IF ground return is the LO input ground plane. A more detailed view of the microstrip balun is shown below in Fig. 16.

15 Figure 16 (a) Coaxial equivalent of the mixer balun; (b) equivalent circuit of the balun; (c) microstrip mixer balun. Before proceeding to our mixer design, there are a number of other mixer designs which should be mentioned. These are shown below in Figs Figure 17. Quadrature IF mixer outline. Figure 18. Single-sideband mixer using two balanced mixers and two 90 o hybrids.

16 Figure 19. Subharmonically pumped mixer using an antiparallel diode pair:(a) mixer circuit; (b) dc iv characteristic; (c) time dependence of the local oscillator voltage and the differential conductance. Figure 20. X-band MIC image rejection mixers.

17 Design Method: The basic steps in designing this mixer are: A) Characterization or acquisition of the diode S parameters (these are provided) B) Design of a wide band diode rf input matching network C) Design of IF low pass filter D) Design of Rat Race Coupler E) Design of the image rejection filter F) Design of bias circuitry G) Fabrication of the circuit (to be done next quarter) H) Testing (to be done next quarter) i) Conversion Loss ii) Noise Figure The completed mixer will (hopefully) have: f 1 0 G H z 2 G H z o IF F re q u e n c y = 1.4 G H z N o ise F ig u re ~ 8.0 d B The substrate material is Rogers RT/duroid 4350 (permitivity r is 3.66 and the thickness is ''). A data sheet may be found on the 132B website. For this design, we will be using HSCH-5340 Beam lead Schottky Diodes for Mixers and Detectors. The data sheet of the HSCH-5300 series is shown below. To find the gaps for diode mounting on the layout, you need to know the diode dimensions, which are listed in the data sheet. By simulating the diode model (equivalent lumped circuits), which is also in the data sheet, you can get the diode admittance characteristics for 1 ma self bias and external bias respectively. The diode Spice parameters are also listed in the data sheet; you need to put these Spice parameters on the ADS diode model from Device-Diode palette for diode DC bias simulation and mixer harmonic balance simulation in the future.

18 Below is the admittance curve of the ADS simulation results for 1 ma self bias.

19 m1 freq=26.00ghz S(1,1)=0.617 / admittance = Y0 * ( j1.554 m2 freq=2.000ghz S(1,1)=0.694 / admittance = Y0 * ( j0.082 S(1,1) m2 m1 freq (2.000GHz to 26.00GHz)

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26 To guide you, the following steps are provided which come from a lab design project employing an Alpha model DMK 6583 X-band GaAs Schottky barrier diode. This has a minimum noise figure of 7dB and a cutoff frequency of 350 GHz. The junction capacitance C j at zero bias has a minimum value of 0.1pF and a maximum value of 0.2 pf. At 1 ma the forward voltage is 600 mv minimum and 800 mv maximum. The package outline is shown below in Fig. 25. Figure 25 A. Characterization of the Diode To give you an idea of what your test data should look like, below are the manufacturers admittance characteristics for three different bias levels.

27 Figure 26 B. Design of a Wide Band Diode RF Input Matching Network Now that you have obtained S 11 for the diode, you will note (see Fig. 26) that the reflection from the diode is fairly large ( > 0.5). This can be matched in three steps: 1. A high impedance transmission line transformer is employed to move the admittance from Y 0 to Y 1 (see Fig. 27) 2. Shunt admittance (see Fig. 27) is employed to ``wrap'' the admittance into a circle (Y 2 ) 3. A quarter wavelength transformer is utilized to center the admittance (Y 3 ).

28 Figure 26 Figure 27

29 ``Step One:'' A transmission line is needed to move from Y 0 to Y 1 (see Fig. 26) so that the conductance at 8 and 12 1 GHz equals the inverse of the conductance at 10 GHz i.e., G 10GHz G GHz i) Normalize diode admittance of line, ( ) point B (see Fig. 28) ii) Rotate around Smith chart, (l), point C iii) Normalize to admittance of 50 line, point D. Step Two: Figure 27 We need to determine the length of a shunt stub at 10 GHz that will cancel the susceptance at 8, 12 GHz. This length should be around 90 0 at 10 GHz. Low impedance stubs are difficult to fabricate and place properly. Instead, it is suggested that you use a radial line stub. (See the note on radial line stubs provided for design information.) ``Step Three:'' The last step is to center the admittance around = 0. Understand that this is difficult (if not impossible) to do this exactly. The idea is to use a transformer to shift the center of the admittance circle. As a first order approximation a quarter wave transformer can be used to center the two resonance points ( Im(Y) = 0 ) around = 0. With wide band data and a computer, a better approximation could be achieved, but considering the inaccuracies in the characterization in step A, this would not necessarily work any better. If you are particularly ambitious, you may wish to write an interactive design program for more complete optimization.

30 In either case, provide a detailed description of your design approach in your laboratory write-up. C. IF Low Pass Filter Design As mentioned earlier, you should use your first lab experience to design a low pass IF filter which will block the X-band signal and LO while passing the 1.4 GHz IF output. D. Rat Race Coupler The other item which you need to design and fabricate is a rat-race coupler. You may wish to refer back to the notes for information on this device. To help you with your design, we have included below a copy of the artwork for a mixer utilizing a rat race. E. Fabrication of the Circuit Figure 28 Fabrication is assumed to done using the standard lithographic techniques available in the 132B laboratory (i.e., these set your design limits). F. Bias decoupling Network The network at the DC bias input is a lowpass filter to remove all AC signals and it provides an RF ground for the diode. The components for this network may be assembled from discrete components which you locate and document, or you may use microstrip equivalent circuits. G. IF Impedance Matching Here, one needs to employ the network analyzer to ``look'' back into the IF port to determine the impedance of the rf and dc biased diode at the IF frequency. Make sure that the power levels you employ for this measurement are sufficiently small that you do not change the diode dc current level. With this information you can design a matching circuit to match the IF output impedance to the input impedance of the next stage in the IF signal path. The next stage is typically a low noise amplifier (LNA), which does not necessarily have a 50 ohm input impedance. However, in this lab the LNA does have a 50 ohm input impedance. For the purposes of this lab, you should use a coaxial tuner terminated in 50 and adjust it for a conjugate match at the IF frequency at the tuner input. Now disconnect the tuner and connect it to the mixer IF output (no adaptors please!). If time permits you should also design and fabricate a microstrip IF matching circuit.

31 H. Testing If we had actually fabricated the mixer in 132B, the next step would be to provide detailed test data on your mixer. In particular,we would wish to experimentally determine the conversion loss and noise temperature (noise figure) of your mixer. There are various techniques for mixer characterization. These include the use of a noise figure meter (H.P. 8970B and H.P. 8971B), the configuration of the mixer at a heterodyne receiver for hot/cold load measurements, the determination of conversion loss using coherent sources and the use of coherent sources to find the minimum detectable signal. APPENDIX A: Mixer Fundamentals Since the LO power drives the diode alternately from forward bias to reverse bias the diode reflection coefficient varies periodically. An equivalent circuit of a diode chip (i.e., without package) is shown in Fig. 2(a) where R B I 28 I S I in m A B a rrie r R e sista n c e C J C BO C o V 1 c Ju n c tio n C a p a c ita n c e

32 and R R V B KI S R e v e rse B ia s R e sista n c e R S E p itaxy L ayer R esistance The junction capacitance and package parasitics result in a transformation of the source impedance to Z. o For the Alpha # package used for the DMK6583 diode, C p 0.06pF and L s 0.5 nh. Since diodes for mixer applications are typically chosen so that the cut-off frequency f 2 R C f and are either strongly forward biased R B c S J R or reverse biased R, the admittance can be written as S B 1 2 Y R R j C R R F B S j B S F o rw a rd B ia s 1 2 Y R R X j C R R S C J R e v e rse B ia s For an LO power level of P L we can express the generator voltage (see Fig. 3) as with V 2 Z P 1 2. If we approximate the diode conductance G as L o L 2V t 2V c o s t L L L G t 1 R R S B L F 2 2 C J, if 2V t V R S, o th e rw ise we obtain the time dependent reflection coefficient shown below in Fig. A-1. We can easily calculate R and F and find Figure A-1

33 R R Z S B 0 1 F S B 2 R R Z Z R 0 0 B R S Z C R 0 J S Z C R R J S 1 Z C R 0 J S As in power supply rectifier circuits, it is of importance to find the conduction angle which represents the normalized period of time in electrical degrees during which the LO drives the mixer diode into the conducting region. This is typically o and is given by V 2 a rc c o s F V 1 V F 2 a rc c o s 2 Z P 0 L To obtain a quantitative expression for the mixer conversion loss it is helpful to Fourier analyze the reflection coefficient (t): where 0 1 L 2 t c o s t c o s 2 t 2 s in 1 F R S B 2 2 Z C R 2 s in 0 L J S R L R Z 2 0 Recalling that we have a signal waveform V voltage wave S R c os t in addition to the LO we can write for the reflected S c o s V t t V t t V c o s t V c o s t c o s t 0 S 1 S L S S 1 t 0 V c o s t V c o s 1 c o s S S S t L S L S t 2

34 The term of particular interest for us in our mixer application is the L - S term since this is at the IF frequency IF. We can then define the conversion efficiency as the ratio of the power reflected at $\omega _ {IF}$ to that incident at S : The conversion loss is expressed in db: 2 2 V 1 S 1 P 0.5 IF 2 P V 4 S S 2 2 S B 2 1 Z C R sin 2 0 L j S 4 R R Z LC 1 0 log 10 To obtain a feeling for the diode design it is convenient to express in terms of the cut-off frequency f 2 C where R C 1. This yields C C S J 4 2 Z X f R 0 2 sin 1 C B 2 X Z f Z C 0 C 0 2 By choosing the X C corresponding to C J to be between Z 2 a n d 2 Z (i.e. 100 ) the quantity in the 0 0 parenthesis is 2. Then to optimize one wishes f / f C << 1, strong LO power to maximize G and low forward voltage to minimize R B. We then have and and 2 2 f 4 1 fc R B Z 0 L C 3.9 d B f 9 f C R B Z 0 Figure A-2 below shows actual results obtained at Alpha from silicon and GaAs single-ended mixers from 3-16 GHz.

35 Figure A-2 A further point of interest involves the match of the rf diode impedance to the LO source impedance. We have specifically V S W R Z Z fo r Z Z L O 0 L O 0 Z Z fo r Z Z 0 L O 0 L O To obtain Z LO we utilize the first order Fourier coefficients of the voltage and current waveforms: 2 V t V V c o s t V c o s 2 t D C L O L L where 2 I t I I c o s t I c o s 2 t D C L O L L and 2V sin L 2 2 I 2 C R V 2 Z R R L O J S L 0 S B We then have and V 2V Z I L O L 0 L O RS RB Z 0 Z V 2 1 L O L Z Z I sin C R Z 0 0 L O J S 0 V S W R Z LO Z 0 1 The IF impedance match is also important since we wish to couple the maximum amount of the IF signal into the amplifier. The IF impedance is given by the ratio

36 Z IF V I 1 didc dvf 2 Z R R 0 S This is always greater than 2 Z and typically ranges from 200 to 500 ohms. Figure B-3 displays mixer 0 parameters as a function of LO drive. F DC B Figure A-3