696 ieee transactions on ultrasonics, ferroelectrics, and frequency control, vol. 55, no. 3, march 2008

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1 696 ieee transactions on ultrasonics, ferroelectrics, and frequency control, vol. 55, no. 3, march 2008 Orthogonal Frequency Coded Filters for Use in Ultra-Wideband Communication Systems Daniel R. Gallagher, Student Member, IEEE, Donald C. Malocha, Fellow, IEEE, Derek Puccio, Member, IEEE, and Nancy Saldanha, Student Member, IEEE Abstract The use of ultra-short pulses, producing very wide bandwidths and low spectral power density, are the widely accepted approach for ultra-wideband (UWB) communication systems. This approach is simple and can be implemented with current digital signal processing technologies. However, surface acoustic wave (SAW) devices have the capability of producing complex signals with wide bandwidths and relatively high frequency operation. This approach, using SAW based correlators, eliminates many of the costly components that are needed in the IF block in the transmitter and receiver, and reduces many of the signal processing requirements. This work presents the development of SAW correlators using orthogonal frequency coding (OFC) for use in UWB spread spectrum communication systems. OFC and pseudonoise (PN) coding provide a means for UWB spreading of data. The use of OFC spectrally spreads a PN sequence beyond that of code division multiple access (CDMA) because of the increased bandwidth providing an improvement in processing gain. The transceiver approach is still very similar to that of a CDMA but provides greater code diversity. Experimental results of a SAW filter designed with OFC transducers are presented. The SAW correlation filter was designed using seven contiguous chip frequencies within the transducer. SAW correlators with a 29% fractional bandwidth were fabricated on lithium niobate (LiNbO 3) having a center frequency of 250 MHz. A coupling-of-modes (COM) model is used to predict the SAW filter response experimentally and is compared to the measured data. Good correlation between the predicted COM responses and the measured device data is obtained. Discussion of the design, analysis, and measurements are presented. The experimental matched filter results are shown for the OFC device and are compared to the ideal correlation. The results demonstrate the OFC SAW device concept for UWB communication transceivers. I. Introduction Ultra-wideband (UWB) communications is an emerging technology with numerous communications advantages. The ability to share the FCC-allocated frequency spectrum, large channel capacity, and data rate, simple transceiver architecture, and high performance in noisy en- Manuscript received May 15, 2007; accepted October 23, This work was partially supported by the National Aeronautics and Space Administration STTR # NNK05OB31C, Kennedy Space Center, and Microsystem Sensors, Inc. D. R. Gallagher, D. C. Malocha, and N. Lobo are with the School of Electrical Engineering and Computer Science, University of Central Florida, Orlando, FL ( danielgallagher@ieee.org). D. Puccio is currently with Quartzdyne, Inc., Salt Lake City, UT Digital Object Identifier /TUFFC vironments has paved the way for emerging wireless technologies such as wireless high definition video streaming, wireless sensor networks and more. The use of ultra-short pulses has become the widely accepted method for achieving the very wide bandwidths and low power spectral density needed for UWB communications. The impulse radio method of UWB communications is effective and simple; however, the implementation of more complex signals, such as continuous wave (CW) UWB, are not feasible with current silicon technologies. Surface acoustic wave (SAW) devices, however, allow for simple generation and detection of complex UWB communication. The numerous advantages of SAW devices for UWB communication transceivers were recently demonstrated using a pseudo-noise (PN)-coded SAW transducer to implement a code division multiple access (CDMA) coded signal on a single-frequency RF carrier [1]. Information is encoded by pulse-phase-modulation used to excite the SAW which can be amplified and then transmitted. Reception is achieved through the correlation of the matched filter received response and baseband envelope detection to extract the pulse phase. The introduction of orthogonal frequency coding (OFC) in UWB SAW correlators provides several advantages over CDMA including an increased data rate resulting from reduced compressed pulse ambiguity and greater multiple access operation due to greater code diversity [2]. The UWB OFC SAW device is capable of being used as both code generator and correlator in a UWB transceiver, as shown in Fig. 1. The use of a SAW correlator eliminates the need for costly high-speed silicon CMOS devices as well as many of the costly components needed in the IF section. This paper presents the development and evolution of SAW correlation filters using OFC for use in UWB communication systems. The evolution of the final device design and a discussion of problems and solutions are presented. The final device design is compared to coupling-ofmodes (COM) model predictions, and experimental data are correlated with its matched filter and compared to the ideal compressed pulse. II. Orthogonal Frequency Coding Definition OFC is a spread spectrum coding technique that has been successfully implemented in SAW tags and sensors using reflective structures [3] [5]. The technique uses multiple orthogonal chips, each τ chip long. In the frequency do /$25.00 c 2008 IEEE

2 gallagher et al.: orthogonal frequency coded filters 697 Fig. 1. Conceptual block diagram of UWB OFC transmitter and receiver. The OFC SAW filter is used as a code generator in the transmitter and correlation filter in the receiver. main, the local chip frequencies are separated by 1/τ chip. The final criteria require that f chip τ chip must equal an integer number of half carrier cycles. A well-known example signal is a linear stepped chirp, which contains a series of local chips with contiguous orthogonal frequencies and linear group delay. For OFC, a level of coding is achieved by shuffling the chips in time such that the adjacent chip carrier frequencies are no longer contiguous in time. Additionally, PN coding is available for an even higher level of code diversity. The OFC waveform is implemented in a SAW device embodiment using interdigital transducers or periodic reflector gratingswith local center frequencies and electrode counts necessary to meet the orthogonality conditions outlined above. The OFC technique permits multiple signals to occupy the same bandwidth with the data contained in the signal phase; a more complete description of the OFC technique is given in [2]. III. Device Design Parameters and Measurement All of the devices presented have the same basic design parameters. Devices were fabricated on YZ LiNbO 3 with aluminum electrodes and acoustic beam width (W a ) of approximately 0.5 mm. The filters were designed with a center frequency of 250 MHz, a fractional bandwidth of 29%, and an insertion loss of 30 db. The FCC defines UWB as a signal with a fractional bandwidth greater than 25% measured at the 10 db points, classifying these devices as UWB. The center frequency of 250 MHz is chosen for proof of concept and ease of implementation using conventional contact photolithography techniques. The devices can easily be scaled to higher frequencies using advanced fabrication techniques such as e-beam lithography. The dispersive and OFC transducers had seven chips with seven different chip frequencies which yield a time bandwidth product of 49. Each chip is 96 ns (τ chip ) long. A wide-band input transducer was used in conjunction with the OFC or stepped chirp transducer. The input transducer was placed on either side and at equal distance from the OFC transducer to allow the filter and its matched filter responses to be obtained using the same device. The frequency response data were obtained using an RF probe and measured two-port S-parameters of a single disper- Fig. 2. Schematic layout of the dispersive chirp transducer. Each chip is weighted based on (6) to achieve a constant conductance across the chirp response. Each chip has a constant length of τ chip.the direction of the up-chirp and down-chirp are marked with arrows, which indicate the direction of the traveling wave from the output transducer connected to port 2. Fig. 3. Schematic layout of the dispersive OFC transducer. The up and down directions of the code are marked with arrows, which indicate the direction of the traveling wave from the output transducer. sive transducer with an output transducer on either side. The time domain response was obtained via a fast Fourier transform (FFT). IV. UWB Dispersive Transducer Design The linear stepped chirp was used as a benchmark to identify design problems and to determine solutions due to its relative simplicity compared to an OFC device implementation. Using up-chirp and down-chirp signals facilitates the identification of issues such as bulk mode problems and other necessary transducer design modifications. To meet orthogonality conditions, every chip was a constant length (τ chip ). In order to keep the chip length constant, the number of electrodes in the chip must increase proportionately as the chip frequency increases. The conductance is proportional to the chip frequency (f chip ), the beam width (W a ), and the chip electrode count. Therefore, it is necessary to apodize the chips in order to obtain a uniform conductance for each chip across the inline dispersive transducer. The chip apodization is shown in Fig. 2 for a chirp configuration and Fig. 3 for an OFC configura-

3 698 ieee transactions on ultrasonics, ferroelectrics, and frequency control, vol. 55, no. 3, march 2008 tion. The tap weights for each chip can be determined by considering the conductance relationship between adjacent chip frequencies. To the first order, the center frequency acoustic conductance for a SAW transducer is given as [6] G 0 =8k 2 f 0 N 2 p C sw a, (1) where k 2 and C s are material properties of the chosen piezo-electric substrate, f 0 is the center frequency of the transducer, and N p is the effective number of electrode pairs. For each OFC chip, N p = 1 BW% = ( BWchip f chip ) 1 = τ chip f chip. (2) Substituting (2) into (1), the acoustic conductance for each chip is G 0chip =8k 2 τ chip 2 f chip 3 C s W achip. (3) From (3), the chip beam width, W achip, is the only available parameter for tuning the chip s acoustic conductance. The remaining parameters are constants defined by the orthogonality conditions and material properties. If the lowest frequency chip is chosen as the reference, the beam width of all subsequent chips will be calculated as a ratio to normalize the overall conductance. The ratio between adjacent chip frequencies is G 0n 1 G 0n = f 3 n 1 W an 1 f 3 n W an, (4) where the subscript n is the chip number. From (4), the ratio of conductances for adjacent chips is proportional to the cubed ratio of those chip frequencies, given as G 0n 1 G 0n f n 1 3 fn 3. (5) Therefore, to achieve constant conductance over the entire frequency band, the beam width for each chip can be calculated as W an 1 = f n 3 W an fn 1 3. (6) The lowest frequency chip, f 1, will have a tap weight of unity, or W a1 = 1. Using (6), and the known value for W a1, all remaining apodization tap weights are able to be determined. V. UWB OFC Device Design Evolution A. Initial Chirp Device Layout The initial chirp transducer design, shown in Fig. 4, was implemented using quarter-wavelength electrodes and was Fig. 4. Microscopic image of initial linear stepped chirp dispersive transducer design with quarter-wavelength electrodes. Chip weightings are visible across the device as electrode periodicity increases with frequency. paired with a wide-band, polarity weighted, output transducer. Polarity weighting was used in the output transducer to achieve higher conductance, broader bandwidth, and lower insertion loss through the device. The experimental results for the up-chirp and down-chirp directions are presented in Fig. 5. The frequency response for the up-chirp indicates that the inter-electrode reflections were not problematic for inline chips with orthogonal local center frequencies, a suspected issue with quarter-wavelength electrodes in the transducer design. For this device design, however, an important consequence in the design of wide-band chirp filters was observed. The device suffered from significant high frequency attenuation when operated in an up-chirp configuration. For up-chirps with fractional bandwidths above about 25%, bulk mode conversion effects within the dispersive transducer begin to cause the frequency response to roll off [7]. Bulk mode conversion occurs for high bandwidths since the bulk acoustic wave (BAW), which travels at a higher velocity than the SAW, can phase match the electrode pattern at the low frequency device side when a BAW mode frequency is higher than that of the synchronous SAW mode of the local electrode pattern. The radiating angle of the bulk wave varies with frequency, producing coherent bulk wave radiation that causes a BAW loss for an up-chirp device at the higher frequencies, but does not occur in down-chirp devices. The effect of coherent bulk mode conversion is evident in the experimental up-chirp frequency response shown in Fig. 5, and is best observed when comparing the timedomain magnitude shown in Fig. 6. The bulk mode conversion problem does not occur in down-chirp devices. For the proper correlation of the received signal and for optimum peak-to-side-lobe (PSL) ratio, the up- and downchirps should be reciprocal. The issue of bulk mode conversion is addressed in the following section. B. Revised Chirp Device Layout The attenuation resulting from bulk mode conversion can be eliminated by using more than two electrodes per period or by using a slanted geometry. Continuing with the in-line transducer geometry, all subsequent device designs use sixth-wavelength electrodes. The revised chirp design

4 gallagher et al.: orthogonal frequency coded filters 699 Fig. 5. Experimentally measured frequency response (S 21 ) of the initial chirp device design utilizing quarter-wavelength electrodes and a polarity weighted output transducer. Frequency response is obtained using an RF probe station and a network analyzer. Fig. 7. Experimentally measured frequency response (S 21 ) of revised chirp device design with sixth-wavelength electrodes and apodized (cos 1 ) output transducer. Fig. 6. Experimental time response of initial chirp device design. The time response is obtained using the FFT of the frequency response shown in Fig. 5. The time axis is normalized to chip length, τ chip. Fig. 8. Experimental time response of revised chirp device design. Time response obtained via FFT. The time axis is normalized to chip length, τ chip. experimental frequency response is shown in Fig. 7 and the experimental time response is shown in Fig. 8. The upchirp and down-chirp frequency responses are nearly reciprocalin frequency and time, showing that sixth-wavelength electrodes effectively eliminated the bulk mode conversion. Using previous experimental data, the velocity under the 3f 0 grating was experimentally determined to be 3418 m/s. A known velocity is an important design parameter needed in centering the output transducer to prevent unintentional in-band frequency attenuation. Various test devices were fabricated using a 3f 0 sampled polarity weighted and an inverse cosine apodized output transducer to ensure wave-guiding parasitics were not present in the device. Improvement in the device response from using a polarity weighted or apodized transducer was minimal; however, the use of the apodized output transducer was selected for subsequent designs due to its welldefined time domain symmetry. Although the revised design produces a reciprocal time response, the frequency response is distorted by the apodized output transducer at the passband edges. C. Final Chirp Device Layout Rather than attempt to flatten the response by redesigning the output transducer window, we used the preexisting apodization on the dispersive transducer chips to compensate for the output transducer window and flatten the overall response. To accomplish the compensation, the chip weights calculated using (6) are adjusted. The resulting apodization profile is evident in the device shown in Fig. 9. The final, compensated, linear stepped chirp frequency response is shown in Fig. 10. The frequency response magnitude for the up-chirp correlates very well with the downchirp, showing that bulk mode conversion effects were effectively eliminated. The chirp filter pass-band was effectively flattened; compensating for the nonuniform conductance in the output transducer. The improved passband shape is better observed in the time response, shown in Fig. 11. The linear time response, shown in Fig. 11(b), displays the continuous transition in chip cycles over the span of the stepped chirp which occur at the zero crossings.

5 700 ieee transactions on ultrasonics, ferroelectrics, and frequency control, vol. 55, no. 3, march 2008 Fig. 9. Microscopic image of final, compensated, linear stepped chirp dispersive transducer design with sixth-wavelength electrodes and apodized input transducer. The compensated chip weightings are visible across the device. Fig. 11. Experimental time response of final chirp device design. The time axis is normalized to chip length, τ chip. Fig. 12. Microscopic image of OFC dispersive transducer design. The optical diffraction grating effect reveals the chips. A. OFC Device Experimental Results Fig. 10. Experimentally measured frequency response (S 21 ) of final stepped chirp device design using compensated apodization weighting on each chip. The continuous chip transition is characteristic of orthogonal frequencies and the stepped chirp configuration yields a constant phase throughout the time response. VI. OFC Device Results Using the insight provided through troubleshooting the linear stepped chirp design, we implemented the OFC device by shuffling the chip frequencies, as described in Section II. The resulting device, with OFC sequence of {f 6,f 3,f 7,f 1,f 4,f 5,f 2 }, is shown in Fig. 12. Multiple code sequences were implemented without considering any code optimization. The OFC device also has output transducers on either side of the OFC transducer, allowing for the upanddownofcdirectiontobemeasuredonasingle device. An additional level of PN phase coding could be implemented in addition to OFC by changing the polarity of the individual chips, adding to the code diversity. The experimental OFC frequency response for the up and down directions are shown in Fig. 13. If we compare the up and down directions of the OFC sequence, the frequency response magnitudes compare extremely well at all frequencies and are nearly reciprocal as desired for proper matched filtering. The time response of the OFC device is shown in Fig. 14. The normalized magnitude response, shown in Fig. 14(a) in db, shows a relatively flat response, as desired. The linear time response, shown in Fig. 14(b), enables the determination of the code sequence by considering the relative carrier cycles of the experimentally measured signal. The transition of the chip frequencies occurs at the zero crossing for the OFC device, as with the chirp device, due to the properties of the orthogonal frequencies. B. Coupling of Modes Simulation Results The device was simulated using a coupling-of-modes (COM) model and compared to experimental results for each stage of design troubleshooting. An observational comparison of model prediction and experimental results was used to easily detect any device design issues. The COM model simulated frequency response is compared to the final experimental OFC device response in Fig. 15. The COM model prediction compares very well with the experimental frequency response, nearly matching the experimental OFC response across the entire passband.

6 gallagher et al.: orthogonal frequency coded filters 701 Fig. 16. Coupling-of-modes-simulated device response compared to the experimentally measured OFC device time response. The time response, shown in normalized magnitude, is obtained using an FFT of the frequency response shown in Fig. 15. The time axis is normalized to chip length, τ chip. Fig. 13. Experimental frequency response (S 21 )ofaseven-chipuwb OFC device with center frequency of 250 MHz and fractional bandwidth of 29%. The up and down directions of the OFC code sequence, as shown in Fig. 3, are comparable in frequency. The experimental up-direction OFC time magnitude response is compared with the COM model in Fig. 16, where the time axis on this figure has been normalized to the chip length. The spike in magnitude occurring at the transition between the third and fourth chips in the experimental up-direction OFC data shows evidence of inter-symbol interference in the time domain. This is a result of the OFC sequence chosen for the device and is accurately predicted by the model simulation. C. Experimental OFC Correlation Results Fig. 14. Experimental time response of UWB OFC device design. The time axis is normalized to chip length, τ chip. The OFC sequence of {f 6,f 3,f 7,f 1,f 4,f 5,f 2 } is labeled under each chip in the linear time plot and can be seen by observing the relative number of cycles in each chip. The experimental OFC correlation between the up and down directions is shown in Fig. 17 and is compared with the ideal OFC correlation. The experimental correlation results were obtained using s-parameter data from RF probe station measurements of both up-direction and down-direction OFC frequency responses. The ideal correlation response was generated using the mathematically ideal OFC time response. Experimental correlation results are in agreement with ideal predictions with respect to the time ambiguity, pulse width and sidelobe level. The correlation produces a compressed pulse approximately 2 N 1 chip τ chip or 0.28 τ chip long, which corresponds to a processing gain of 49 with respect to additive white Gaussian noise, resulting from the seven-chip, sevenfrequency OFC signal. This is seven times greater than the processing gain of a PN sequence of the same time length using a single frequency carrier. The traces shown in Fig. 17(a) and Fig. 17(b) are, respectively, narrow and wide time window scalings of the same correlation result. The figures show that the experimental correlation result coincides with the ideal case very well across both time ranges. Fig. 15. Coupling-of-modes-simulated device response compared to the experimentally measured OFC device frequency response. Both responses are the up-direction OFC code sequence. VII. Discussion and Conclusions This paper presented the development of SAW correlators using OFC for application in UWB spread spec-

7 702 ieee transactions on ultrasonics, ferroelectrics, and frequency control, vol. 55, no. 3, march 2008 and are nearly reciprocal, as desired. Experimental results were compared to simulated COM model results and agree very well in both frequency and time. The compressed pulse with a processing gain of 49 was verified from experimental data. These results demonstrate the feasibility of using an OFC SAW correlator in UWB communication transceivers. References [1] R. Brocato, E. Heller, J. Wendt, J. Blaich, G. Wouters, E. Gurule, G. Omdahl, and D. Palmer, UWB communication using SAW correlators, in Proc. IEEE Radio Wireless Conf., Atlanta, GA, Sep. 2004, pp [2] D. Malocha, D. Puccio, and D. Gallagher, Orthogonal frequency coding for SAW device applications, in Proc. IEEE Ultrason. Symp., vol. 2, Aug. 2004, pp [3] D. Puccio, D. Malocha, N. Saldanha, D. Gallagher, and J. Hines, Orthogonal frequency coding for SAW tagging and sensors, IEEE Trans. Ultrason., Ferroelect., Freq. Contr., vol. 53, no. 2, pp , [4] D. Puccio, D. Malocha, D. Gallagher, and J. Hines, SAW sensors using orthogonal frequency coding, in Proc. IEEE Int. Freq.Contr.Symp.Expo., 2004, pp [5] D. Puccio, D. Malocha, and N. Saldanha, Implementation of orthogonal frequency coded SAW devices using apodized reflectors, in Proc. IEEE Int. Freq. Contr. Symp. Expo., 2005, pp [6] C.S.Hartman,D.T.Bell,andR.C.Rosenfeld, Impulsemodel design of acoustic surface-wave filters, IEEE Trans. Sonics Ultrason., vol. 20, no. 2, pp , [7] D. P. Morgan, Surface-Wave Devices for Signal Processing. ser. Studies in Electrical and Electronic Engineering, vol. 19, New York: Elsevier Science Publishers, Fig. 17. Correlation results of experimental up and down direction OFC device data compared to ideal correlation response. Experimental device data are obtained from the same dispersive OFC transducer with an apodized output transducer located at equal distance on each side. The two plots show alternate time axis scalings of the same data. trum communications systems. A brief review of OFC and its inherent advantages was discussed, including the reduced compress pulse ambiguity and enhanced processing gain. The UWB OFC device development utilized a linear stepped chirp filter device embodiment, aiding in the identification of bulk mode problems and necessary transducer design modifications. It is shown that the shape of the transfer function can be enhanced by adjusting the conductance of chips within the dispersive transducer, thereby reducing unintended signal distortion. The experimental results of an UWB OFC SAW device with 250-MHz center frequency and a 29% fractional bandwidth were presented. The up- and down-direction OFC experimental responses correlate with each other very well Daniel R. Gallagher (S 02) was born in Orlando, FL, on September 7, He earned his B.S. and M.S. degrees in electrical engineering from the University of Central Florida, Orlando, FL, in 2003 and 2007, respectively. Currently, Daniel is pursuing his Ph.D. degree at the University of Central Florida. His research includes ultra-wideband communication systems and surface acoustic wave technology. He is a National Aeronautics and Space Administrations Graduate Student Researchers Program fellowship and Interdisciplinary Information science and technology (I 2 Laboratory) fellowship recipient. Donald C. Malocha (S 69 M 74 SM 84 F 93) earned his B.S. degree in electrical engineering/computer science and his M.S. and Ph.D. degrees in electrical engineering from the University of Illinois, Urbana, in 1972, 1974, and 1977, respectively. He joined the University of Central Florida (UCF) in 1982 and is a professor in the Electrical and Computer Engineering Department. He holds joint appointments in the Mechanical, Materials, and Aerospace Engineering Department and the Advanced Materials, Processing and Analysis Center (AMPAC). He has been a visiting scholar at the Swiss Federal Institute of Technology, Zurich, and the University of Linz, Austria.

8 gallagher et al.: orthogonal frequency coded filters 703 In industry, Don was a member of the Corporate Research Laboratories at Texas Instruments, Dallas; Manager of Advanced Product Development for Sawtek, Orlando; and a visiting member of the Technical Staff, Motorola, Phoenix, AZ, and Fort Lauderdale, FL. Don was an elected member of the IEEE UFFC AdCom and served for over 10 years in several chairs, including president. He received the Electronic Industries Association s 1998 David P. Larsen Award, the 2000 IEEE Third Millennium Medal, and the 2004 UCF Distinguished Researcher of the Year award. Currently, he is an associate editor for UFFC Transactions, and serves on the IEEE symposium technical program committees for ultrasonics and frequency control. He is the Technical Program Chair for the upcoming 2006 and 2007 International Frequency Control Symposium. His research emphasis is in materials, communications, device analysis, and fabrication related to acoustoelectronic technology. His group is currently working on SAW and BAW sensor devices and systems. He is an IEEE Fellow, a member of Tau Beta Pi, Eta Kappa Nu, and member emeritus of the Electronic Industries Association. Derek Puccio (S 99 M 06) was born in Ridgewood, NJ, on October 1, He received his B.S., M.S., and Ph.D. degrees in electrical engineering from the University of Central Florida, Orlando, FL, in 2001, 2003, and 2006, respectively. Early in his academic career, his research focused on the characterization of newly manmade SAW materials. Specifically, he studied langasite-structure compounds such as langatate and CTGS. His Ph.D. dissertation describes a coded, wireless SAW sensor embodiment using a technique called orthogonal frequency coding. Since graduating in the summer of 2006, Dr. Puccio has been employed as a staff scientist at Quartzdyne, Inc., in Salt Lake City, UT. Quartzdyne manufactures high precision, high pressure quartzcrystal-based transducers for the oil and gas industry. Dr. Puccio has been a member of IEEE and UFFC-S since 1999 and 2001, respectively. He is a National Aeronautics and Space Administration s Graduate Student Researchers Program fellowship recipient. He is also a member of Tau Beta Pi and Eta Kappa Nu. Nancy Saldanha (S 00) was born in May She received her B.S. and M.S. degrees in electrical engineering from the University of Central Florida in Orlando, FL, in 2001 and 2003, respectively. During her master s program, her research focused on the characterization of new SAW materials, specifically langasite structured compounds. She is currently pursuing her Ph.D. degree at the University of Central Florida. Her research interests include SAW and BAW sensor technology. Nancy has been a student member of IEEE and UFFC-S since 2000 and 2001, respectively. She is a National Aeronautics and Space Administration s Graduate Student Researchers Program fellowship recipient and a member of Tau Beta Pi.