High-Frequency Power Transformers with Foil Windings: Maximum Interleaving and Optimal Design

Transcription

1 0.09/TPEL , IEEE Transactions on Power Electronics High-Frequency Power Transformers with Foil Windings: Maximum Interleaving and Otimal Design Ernesto L. arrios, Student Member, IEEE, ndoni Urtasun, Student Member, IEEE, lfredo Ursúa, Member, IEEE, Luis Marroyo, Member, IEEE, and Pablo Sanchis, Senior Member, IEEE bstract Foil conductors and rimary and secondary interleaving are normally used to minimize winding losses in high-frequency transformers used for high-current ower alications. However, winding interleaving comlicates the transformer assembly, since tas are required to connect the winding sections, and also comlicates the transformer design, since it introduces a new tradeoff between minimizing losses and reducing the construction difficulty. This aer resents a novel interleaving technique, named maximum interleaving, that makes it ossible to minimize the winding losses as well as the construction difficulty. n analytical design methodology is also roosed in order to obtain free-cooled transformers with a high efficiency, low volume and, therefore, a high ower density. For the urose of evaluating the advantages of the roosed maximum interleaving technique, the methodology is alied to design a transformer ositioned in the 5 kw-50 khz intermediate high-frequency resonant stage of a commercial PV inverter. The roosed design achieves a transformer ower density of 8 W/cm with an efficiency of 99.8%. Finally, a rototye of the maximum-interleaved transformer is assembled and validated satisfactorily through exerimental tests. Index Te Foil windings, high-frequency, maximum interleaving, otimization, transformer design. F I. INTRODUCTION or alications requiring galvanic isolation, the linefrequency transformer has traditionally been the heaviest, most exensive and least efficient comonent art of an electronic ower converter. Nowadays, there is a large number of medium ower alications (-5 kv) with a limited weight and sace, such as electric traction systems, distributed generation systems (PV anels and mini-wind turbines) and ower sulies, in which cost and efficiency are aramount. In these alications, one of the most widely adoted solutions for achieving considerable reductions in weight and volume whilst significantly increasing efficiency, yet still maintaining the required galvanic isolation, is to Manuscrit received Setember 5, 04; acceted November 0, 04. Coyright 04 IEEE. Personal use of this material is ermitted. However, ermission to use this material for any other uroses must be obtained from the IEEE by sending a request to ubs-ermission@ieee.org. This work was suorted in art by the Sanish Ministry of Economy and Cometitiveness under Grants DPI00-67-C0-0 and DPI0-485-R and by the Public University of Navarre. The authors are with the Deartment of Electrical and Electronic Engineering, Public University of Navarre, Pamlona, Sain ( ernesto.barrios@unavarra.es; andoni.urtasun@unavarra.es; alfredo.ursua@unavarra.es; luisma@unavarra.es; ablo.sanchis@unavarra.es). increase the transformer oerating frequency to the range of -50 khz [] [4]. In order to reduce winding loss, the rimary and secondary windings of high-frequency (HF) transformers are usually sectioned and interleaved, and secial wire geometries, such as litz and foil conductors, are used [5] [7]. Comared to litz wire, a foil conductor is referred due to its higher width-tothickness ratios, which rovide lower DC resistances and higher fill factors, its better heat conduction which facilitates heat transfer to the environment, and its lower cost [], [8]. However, the conventional rocedure for interleaving foil windings comlicates the transformer assembly since tas are required to connect in series the beginning and end of each winding section. lthough secial techniques have been roosed to interleave windings and minimize both losses and assembly difficulty [8], [9], they have limitations since they are only alicable either to transformers with turns ratios that are close to the unity or to lanar transformers. Furthermore, the design of HF transformers needs to address the interdeendence between core sizing, foil thickness sizing and winding interleaving, which comlicates the design rocess. The roblem is usually solved by means of an iterative rocess that deends on the designer's exerience and may entail the omission of some of the comlex electromagnetic, thermal and construction interdeendencies existing amongst the design arameters [5] [7], [0] [4]. To address the challenge of otimal HF transformer design, this work rooses firstly an innovative technique, termed maximum interleaving, which makes it ossible to minimize losses whilst minimizing the number of tas required and, consequently, the construction difficulty. Then, a novel non-iterative analytical methodology to otimally design HF transformers is roosed, leading to the direct resolution of the design roblem as a non-linear otimization roblem of only four design variables. The methodology comrehensively formulates the comlex multi-hysical henomena resent in the oeration of a HF transformer and the deendencies that arise between these henomena in the transformer design rocess. Moreover, given the fact that winding interleaving may not be advisable in some alications, such as when a high leakage inductance or a low interwinding caacitance is required or in high voltage alications, the scoe of the design methodology roosed in this aer is extended to (c) 0 IEEE. 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2 0.09/TPEL , IEEE Transactions on Power Electronics include a non-interleaved aroach. This is then used as a reference oint to assess the advantages of the maximum interleaving technique. The aer is organized as follows. Section II resents the models for the calculation of the core loss, foil winding loss and transformer thermal resistance for use in the design rocess. Section III exlains the maximum interleaving and the winding loss calculation when this technique is alied. Section IV discusses the transformer design roblem and rooses the otimal design methodology. With this methodology two theoretical designs for a PV alication are obtained in Section V, one with maximum-interleaved and the other with non-interleaved windings. When comaring both designs, the maximum interleaving design achieves the best erformance in te of ower density and efficiency. rototye for the maximum interleaving design is then built in Section VI, aying articular attention to the assembly rocess and the use of low-cost standardized materials. The rototye is finally validated by means of exerimental tests with satisfactory results. two also deend on the lace where the windings are wound. Table I shows, for the two most common winding and core tyes, the five characteristic dimensions based on the three non-dimensional coefficients c, c, c and the dimensional factor a defined in Fig.. In the fourth column, the characteristic dimensions are exressed based on the characteristic coefficients mlt c, v e, a c, a w, v c and the dimensional factor a in order to facilitate its use in the design rocess. a V c w c a c a V c w a c a c a II. TRNSFORMER MODELING. Core Loss and Geometry In the transformer design, the calculation of the core loss is made in ractice through emirical formula based on the Steinmetz equation [5]. For tyical HF ower transformer alications, in which the voltage alied is rectangular in form, with or without zero voltage eriods, the modified Steinmetz equation (MSE) [6], the imroved generalized Steinmetz equation (igse) [7], and the imroved igse (i GSE) [8] have been shown to be accurate core loss emirical models [8] [0]. Each model imroves the accuracy of the revious one but with an increasing comlexity. Thus, the MSE is generally used to calculate the core loss in design rocesses due to its good trade-off between accuracy and simlicity []. Its exression is [6]: P C c m f f eq x y c T oe ct oe ct Vc () where f is the alied voltage waveform frequency, is the magnetic induction amlitude, τ oe is the oerating temerature of the magnetic material, V c is the magnetic core volume, C m, x, and y, and c T, c T and c T0 are the losses and temerature coefficients for the material, resectively, as rovided by the manufacturers in their datasheets, and f eq is the equivalent frequency: T d feq dt. () 0 dt Fig. shows the geometry of the double U and double E cores commonly used in ower alications. Traditionally, the core geometry is characterized by five characteristic dimensions: mean length turn of a winding that comletely fills the window (MLT c ), equivalent volume including the core and windings (V e ), effective cross-sectional area of the core ( c ), window area ( w ), and core volume (V c ). The latter three dimensions only deend on the tye of core whilst the first 0 c (a) (b) Fig.. Characteristic dimensions MLT c, c, w, and V c, dimensional factor a and form coefficient c, c, and c, for the main ower cores: (a) double U and (b) double E. y alying these generic form exressions, the core loss can be exressed as follows: y Pc K a () where coefficient K is: K k MLT c C f v c c (4) x mag m c T T T 0 where k mag is the ratio between the losses for a non-sinusoidal magnetic induction and those for a sinusoidal one, and can be exressed as a function of the length of the zero-voltage eriod in rad, θ, as follows [9]: k mag c a 8 c x yx. (5). Foil Winding Losses When oerating at high frequencies, the amlitude and nonuniformity of the current density distribution in the winding cross-sectional area increases due to the well-known skin and c MLT c c a TLE I. CHRCTERISTIC DIMENSIONS FOR MIN POWER CORES: DOULE E ND DOULE U Core characteristic dimensions Only one leg double U Shell tye double E Generic form MLT c (c+c+) a (c+c+) a mlt a V e (c+) (c+) (c+) (c+) (c+c) a (c+ c) a v e a c c a c a a c a w c c a c c a a w a V c c(c+c+)a c(c+c+5/4)a v c a (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

3 0.09/TPEL , IEEE Transactions on Power Electronics roximity effects. This can lead to an increase in winding loss. In order to reduce this increase as far as ossible foil conductors are used. The calculation of the foil winding loss has aroused great interest since Dowell s work [] and its generalization in [] until nowadays []. The ower loss in a foil winding section of layers filling the full window height with thickness h is calculated in this work by means of the exression roosed by Snelling in [4] based on the aroximation of the analysis made by Dowell in [] for h δ: 4 5 h Pw Rdc I (6) 45 where I is the winding current, δ is the skin deth, and R dc is the dc resistance. The exressions for these latter two arameters are: R f i MLT N, (7) w dc (8) h w f where f i is the current waveform frequency, MLT w is the mean length turn of the winding, N is the number of turns, σ is the material conductivity, μ is the magnetic ermeability and w f is the foil height. When the foil thickness is greater than the skin deth, this exression overestimates the winding loss. C. Thermal Modeling It is ossible to use both theoretical and emirical models to estimate the thermal resistance of the transformer in a steady state. wide range of theoretical thermal models are available, deending on the heat transfer mechanisms considered and their interretation. Emirical models achieve a similar accuracy to theoretical ones, but with greater simlicity [6], [], and are therefore generally referred for use in the design rocess. In fact, the value of the transformer thermal resistance R th is robably the most uncertain arameter in the entire transformer design []. Studies made by the magnetic material manufacturers show that it is ossible to establish an emirical relationshi between the thermal resistance of the transformer and the volume of its core [5], [6]. ased on the data included by the manufacturers in their alication notes for various double E and double U ferrite cores, for a temerature increase of 50 ºC, and including the core volume in its generic form as shown in Table I, the emirical formula for the thermal resistance R th of a naturally-cooled transformer is as follows: R th (9).56 v a 0.5 c where R th is exressed in ºC/W, v c is a non-dimensional factor and a is exressed in meters. Very similar exressions are commonly used in design rocesses for soft magnetic materials with relatively high thermal conductivity such as ferrites, nanocrystalline, and amorhous iron alloys [7]. III. MXIMUM INTERLEVING: OPTIML WINDING DISTRIUTION. Conventional Interleaving Thanks to the resence of a secondary winding, it is ossible to reduce the amlitude of the magnetomotive force (f mm ) in the window resonsible for the roximity effect. With this end, rimary and secondary windings are usually divided into sections and interleaved. The interleaving reduces the number of layers er section and, thus, as can be seen from alying (6), also the losses in the interleaved winding. However, this entails considerable construction difficulty as, in order to series-connect the last turn of one section with the start of the following section, tas need to be made. s a result, the maximum feasible interleaving is limited by this construction comlexity [5], [8]. Considering now an examle in which four turns for the rimary and eight for the secondary are required with rimary current value i, Fig. shows three different ossible winding configurations and their resultant f mm distributions. s indicated in Fig. (a), one otion is to make a 4-8 winding arrangement, i.e. no interleaving, having a rimary section with four turns (=4) and a secondary section with eight (=8). Consequently, maximum values are obtained for the f mm and roximity effect losses, and no tas are required, thereby the winding rocess difficulty is minimum. s shown in Fig. (b), another otion is to divide the windings into two grous with a -4 configuration. This is an intermediate interleaving comrising two rimary sections with two turns each (=) and two secondary sections with four turns each (=4). In this case, in comarison with the first otion above, the maximum f mm has been halved, leading to lower losses. However, the construction rocess is more comlicated, consisting in making two turns with the insulated rimary foil and then cutting it. The insulated secondary foil is then wraed around the rimary and cut. The rocess is then reeated, but when starting the second rimary section, the end of the first section needs to be connected to the beginning of this second section. This connection is called a ta. The same rocedure is followed for taing the secondary sections. This interleaving level is an accetable trade-off between losses and construction difficulty [5] and, therefore, manufacturers do not usually continue increasing the interleaving. When the number of layers er section for the rimary and secondary is minimized, the windings are fully interleaved and the f mm and the roximity effect losses are also minimized. However, this design creates the greatest construction difficulty, requiring the higher number of cuts and tas. s indicated in Fig. (c), in the examle studied, the windings are fully interleaved when four grous are formed with a - configuration. Consequently, six tas are needed; three tas to series-connect the rimary turns and another three to series-connect the end of each secondary section with the beginning of the next (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

4 0.09/TPEL , IEEE Transactions on Power Electronics 4 MGNETIC CORE TPS with the least number of turns (N ) is termed whilst the one with the greatest number of turns (N ) is named. One foil conductor is taken for winding, and foil conductors for winding, with equal to: TP f mm 4i i i i 0 x f mm i i 0 x (a) (b) (c) Fig.. Magnetomotive force in the cross section of a transformer window for three different winding interleaving arrangements: (a) non-interleaved, (b) intermediate interleaving, and (c) fully interleaved. It should be ointed out that the interleaving affects the arasitic elements of the transformer. The greater the interleaving, the lower the energy stored in the stray magnetic field and the greater the energy stored in the stray electric field between the rimary and secondary windings. In this way, the greater the interleaving, the lower the leakage inductance and the greater the caacity between the rimary and secondary windings [7]. Consequently, deending on the alication, the interleaving may not be advisable. For instance, when electromagnetic interference (EMI) needs to be minimized, when leakage inductance is used as a filter or resonant comonents [7], [8], or in high voltage alications in which rimary and secondary windings are groued into searate chambers due to isolation concern, a non-interleaved structure is referred.. Maximum Interleaving of Foil Windings In order to maximize the reduction of the roximity effect yet without a comlicated assembly, a new winding interleaving technique, named maximum interleaving, is roosed in this aer. With this technique, the f mm distribution is the same as for the conventional fullyinterleaved winding configuration, however, due to the construction rocess roosed, the number of tas required is reduced to the minimum technically necessary. In order to make it easier to understand the roosed technique, this is firstly alied to the examle above (see Fig. ) and it is then generically described. In the first ste, three insulated foils are wound around the magnetic core s central leg to make four turns. The first foil has the rimary winding thickness, while the second and third foils have the secondary thickness. Finally, and as detailed in Fig., the beginning of the third foil is soldered to the end of the second foil through a ta, so that the second and third foils are series connected and eight turns for the secondary winding are comleted. In so doing, the same minimum losses as for conventional full interleaving are achieved yet with a much simler construction, given the fact that only one ta is needed instead of the conventional six. The maximum interleaving technique roosed in this aer is generically described below. It can be easily alied to any transformer ratio, even a non-integer number. The winding f mm i 0 x (a) (b) Fig.. Maximum winding interleaving of a 4 rimary and 8 secondary turns foil transformer: (a) cross section of the magnetic core central leg, and (b) window cross section (Section - ). N round round (0) n N where generic transformation ratio n, taken as the quotient between the number of turns for windings and, equal to N /N s for ste-u transformers and N s /N for ste-down transformers. The conductor for winding and the conductors for are insulated from each other and then laced one on to of the other in order to roceed with the winding. From here onwards, two cases can be differentiated: - Decimal art of /n 0.5: Fig 4 shows the cross section of the central leg of a double E core with a generic winding distribution for this case. s can be seen, the conductors are wound jointly and continuously, with the conductor of the winding with the least number of turns ositioned inside, until z number of turns has been reached: N z floor. () + Fig. 4. Cross section of a double E core with maximum interleaving when /n is rounded uwards. Winding in blue and in green. Turn z indicates the end of the winding of the - external conductors of winding, where is: N z '. () N z z N Section (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

5 0.09/TPEL , IEEE Transactions on Power Electronics 5 In this way, - foils of winding are cut at turn z as indicated by an x in Fig. 4. Then, the remaining turns are wound until N is reached, however in this case conductors are used for winding. Once the winding has been comleted, the next ste is to use tas to connect in series the ends of the various layers of the winding. The end of layer is connected to the start of layer, and so on, until the end of layer - is connected to the start of layer. If is less than, then this means that when z turns have been wound, less turns need to be given to winding than to winding. In these cases, it is sufficient to equal to and make the remaining turns until N has been reached. Then, the remaining turns are made for winding until reaching N. n examle of this secial case is shown in Section VI. - Decimal art of /n 0.5: Fig 5 shows the cross section of the central leg of a double E core with a generic winding distribution for this case. Likewise, the windings are made jointly and continuously, however locating the conductors of the winding with the largest number of turns on the inside. So N turns are made around the central leg of the core, and then the winding of is comleted by one more turn with conductors: ' N N. () For this urose, the - external conductors of winding are cut and ended after turn N. Finally, the layers of winding must be series connected. s in the case above, the end of layer must be connected to the start of and so on until the end of layer - has been connected to the beginning of layer. If is greater than, then after turn N an additional turn should be given with conductors of winding, cutting the - external conductors and giving an additional turn with the remaining - conductors. + Fig. 5. Cross section of a double E core with maximum interleaving when /n is rounded downwards. It can be concluded that the maximum interleaving serves to minimize the roximity losses in the transformer windings whilst it also minimizes the construction comlexity, given the fact that the number of tas required is reduced to -. N N + C. Total Winding Losses Calculation ) Maximum-Interleaved Windings The exression for the total ower loss in the transformer windings when imlementing maximum interleaving is now obtained for use in the novel transformer design rocess that is roosed in Section IV. Firstly, the number of turns N for any winding can be exressed according to the magnetic induction amlitude with frequency f by means of the voltage equation V induced in the said winding: V N (4) 4 k sh f ac a where k sh is the waveform factor, equal to when a square voltage is alied and to. when the voltage is sinusoidal. For other voltage wavefo, k sh have to be secifically calculated as indicated in [7]. When the maximum interleaving is imlemented, it can be assumed that the mean length turn MLT w is equal for both transformer windings, and equal to the mean length of the core MLT c. The winding with a least number of turns, named, has a single layer of thickness h in all its sections, whilst the other, named, has layers of thickness h in most of its sections and in the rest. To simlify the design rocess it is considered that, across the length of the winding, there is a constant number of layers er section. Introducing (4) in (6), including the generic exression for the characteristic core dimensions develoed in Table I, exressing the foil height w f as the height of the window c a multilied by height fill factor k h, and articularizing for each of the windings, the losses are obtained for the winding with the least number of turns P w and for the winding with the greatest number of turns P w : mlt I, V, 4h P w 4, (5) k 45 4 sh f kh c ac a h mlt I, V, 5 h P w 4.(6) 4 ksh f kh c ac a h 45 y referring voltages and currents to the winding with the least number of turns by means of transformation ratio n defined in (0), introducing δ according to the current frequency f i in (7), and adding both exressions, the total losses in transformer windings P w,t are obtained: mlt n P w, t K K 4 h 5 n h (7) a h h where coefficients K and K have the following exressions: V, K 4 k k K sh h 45 I,, (8) c a f c f i. (9) (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

6 0.09/TPEL , IEEE Transactions on Power Electronics 6 s the length of the tas is negligible and the number of tas is minimum, the increase in losses due to the tas in maximum-interleaved windings can be disregarded. ) Non-Interleaved Windings s exlained above, sometimes a non-interleaved winding distribution is referred. In this case, the number of layers er section is equal to the number of turns of the winding N. Thus, articularizing (6): MLTw N I 5 N h Pw 4. (0) w f h 45 Furthermore, the mean length turns of the different windings are no longer equal. For the case of two windings wound around the central leg of a shell tye core, the winding and winding mean length turns, MLT and MLT resectively, can be aroximated as: MLT mlt a c c a, () MLT mlt a c c a. () Introducing (4) in (0), including the generic exression for the characteristic core dimensions develoed in Table I and in () and (), articularizing for each of the windings, and following a similar rocedure to the one indicated in the section above, the total losses in non-interleaved transformer windings are obtained: K mlt mlt n Pw, t K mlt h n mlt h a h h K mlt h 4 mlt h 6 a () n where coefficient K 4 is: K 4 f V I i,, 6 k sh k h c ac f. (4) If the waveform is not sinusoidal, and both for fully interleaved and for non-interleaved windings, then P w,t can be calculated as the sum of the losses due to each of the current harmonics P w,j [9]: n max P w, t j P w, j (5) where n max is the greatest harmonic considered. For each harmonic, the harmonic current I,,j and frequency f i,j need to be introduced in coefficients K, K, and K 4. IV. TRNSFORMER DESIGN METHODOLOGY For a design to be feasible, two requirements must be met. Firstly, the windings must fit into the magnetic core window. s the foils fully take the core height u, this requirement named the geometric limit in this aer, turns into an inequality exressing that the total winding width needs to be lower than the core width c a (Fig. ) as follows: N h N h k w c a (6) where k w reresents the loss of available sace due to both the coil former and the insulations between turns of the same winding and between windings. y articularizing this inequality for maximum interleaving, referring the number of turns for winding to winding, and relacing the number of turns for winding by its exression in (4), the geometric limit can be exressed as follows: K 5 h h K 6 K 7 c a 0 (7) a n where coefficient K 5 has the following exression: V K, 5. (8) 4 k sh f ac Coefficients K 6 and K 7 model the thickness of the electrical insulation. Thus, they are a function of the tye of interleaving imlemented. For maximum interleaving, K 7 is equal to the coil former thickness k cf and K 6 is: K 6 g g (9) where g is the insulation thickness between windings and and g is the thickness of the insulation between the turns of winding. For non-interleaved windings, K 7 is equal to the sum of k cf and g and K 6 is: g K g 6 (0) n where g is the thickness of the insulation between the turns of winding. The second requirement, named thermal criterion, is that the transformer must be able to transfer the heat roduced by its total losses P t to the environment with a temerature rise Δτ equal to or less than the maximum allowable rise Δτ max : R th P t max () where P t is obtained by adding the core loss P c in () to the total winding loss given by (7) for maximum-interleaved and () for non-interleaved windings. Furthermore, the transformer design needs to deal with two contradictory design criteria, which are minimum volume and maximum efficiency. For the ower range studied in this aer, natural ventilation is adoted as a good trade-off between these two design criteria, given the fact that its relatively low dissiation caacity ensures high efficiencies [0], []. Consequently, minimizing the transformer volume is considered to be a riority criterion in the free-cooled transformer design rocess. In this design rocess, the thermal rise is re-set to 50 C and, thus, the emirical exression for thermal resistance in (9) is alicable. s will be shown below, this exression also offers good results for different temerature rises tyical of design rocesses. If the emirical exression of the thermal resistance in (9) is introduced in the thermal criterion in (), the volume of the core can be exressed as: P t v c a. ().9 max Two conclusions can be drawn from (). Firstly, in order to minimize the core volume, the temerature increase must be maximized and, consequently, the thermal limit must be converted to an equality. Secondly, by minimizing the total (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

7 0.09/TPEL , IEEE Transactions on Power Electronics 7 losses, the transformer core volume is minimized to give an otimal design. Consequently, the design roblem becomes a non-linear otimization roblem in which the function of total losses is minimized, subject to the equality imosed by the thermal criterion and the geometric limit: min P s.t. s.t. t R N P th t h max N h thermal k w equality c a geometric () inequality characteristics, making it a highly interesting tool for designers of ower electronics converters. st nd DESIGN SPECIFICTIONS S V f n τ amb_max k sh c, c, c RNGES Mag. Mat. SCN CORE TYPES ND SHPE COEFFICIENTS SYSTEM RESOLUTION () s can be seen, the resolution of () is comlicated due to the considerable number of variables involved and the strong non-linear nature of the roblem. However, as the range of soft magnetic materials on the market is limited, there are few ossible values for C m, x, y and sat. Furthermore, shae coefficients c, c and c of the two high-ower core shaes are closely delimited. Therefore, the solution roosed consists in covering the ranges of all the ossible values for these coefficients and solving () with resect to the following four design arameters: dimensional factor a, magnetic induction amlitude, and rimary and secondary foil thicknesses h and h s, which are equivalent to h and h, resectively, for a ste-u transformer and to h and h for a ste-down transformer. The flowchart for the roosed design methodology is shown in Fig. 6. In the first ste, the design secifications are established, including the magnetic materials available, the core tyes and the shae coefficient ranges to be considered. The second ste scans the various combinations between the core tyes and shae coefficients considered. For each combination, the otimization roblem resented in () is resolved for each magnetic material. Once a set of core shae coefficients c, c, and c has been determined, the minimization of the core volume or, in other words, the minimization of dimensional factor a, is equivalent to the minimization of the equivalent volume of the comlete transformer. From the different magnetic materials, the one that achieves the minimum value for a is selected, thereby ensuring that the best design in te of minimum volume and minimum losses is selected. Finally, in ste three, once all the core tyes and ranges for the shae coefficients have been scanned, the designs stored in the second ste are comared and the design with the smallest equivalent volume is selected as the otimal overall design. Unlike traditional design methods [5] [7], [], [], the roblem resolution by means of the design methodology roosed is not iterative and is not based on generally acceted rules of thumb founded on exertise but not theoretically justified. Instead, it is ossible to make a comrehensive analysis of the design roblem thanks to its formulation through analytical models. Therefore, the result obtained is neither conditioned by the designer's rior exerience nor by market limitations (for instance, the commercially available magnetic cores and windings). The methodology makes it ossible to achieve otimal theoretical designs and to analyse the trends of the key design MG. MT. C m, x, y, sat. MG. MT. k C mk, x k, y k, satk rd (a,, h, h s ) ot. (a,, h, h s ) otk Otimal magnetic material design selection: MINIMUM a ot Fig. 6. Foil design methodology flowchart. V. DESIGN EXMPLE ll core tyes and c, c, c combinations scanned? In order to illustrate and validate the design methodology, this section discusses the design of a transformer for use in the commercial single hase resonant inverter for PV systems shown in Fig. 7. This inverter comrises an initial boost stage, an unregulated intermediate high frequency ZCS series halfbridge resonant converter in which the transformer is included and a final grid-connected full-bridge inversion stage. The main secifications for the design are a nominal ower S of 5 kw, a 50 khz oerating frequency and a ste-u transformation ratio /n of.6. s the transformer oerates in a 60 khz resonant tank, the alied voltage is a 5 V square waveform resulting in a triangular magnetizing current. Furthermore, as a result of the resonance between half bridge caacitors C R and inductor L R, the current has a truncated sinusoidal waveform mainly comosed of the fundamental and third harmonic comonents I 50 and I 50, resectively. Thus, when evaluating the winding loss, the nominal rimary current consisting of a 50 khz-4 fundamental and a 50 khz-4.7 third harmonic is considered as indicated in (5). The maximum ermitted ambient temerature is 50 C which, together with a maximum oerating temerature of 00 C, gives a maximum temerature increase of 50 C. Finally, the considered ranges for core shae coefficients c, c, c are 0.-, -4, and -6, resectively. Table II shows the characteristics and loss coefficients for the soft magnetic materials that are suitable for this alication. In order to investigate the benefits of the maximum interleaving technique when comared to a non-interleaved solution, the design methodology described in Fig. 6 is now imlemented in MTL, for both maximum-interleaved and non-interleaved windings. YES MIN V e selection: OVERLL OPTIML DESIGN (C m, x, y, sat, core tye, a, c, c, c,, h, h s ) ot NO (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

8 0.09/TPEL , IEEE Transactions on Power Electronics 8 C R L R HF Power Transformer Main Grid PV anels C R oost Stage Half-ridge ZCS Series Resonant Stage Full-ridge Stage Fig. 7. PV grid-connected ower system including the 50 khz 5 kw transformer. TLE II SOFT MGNETIC MTERILS FOR POWER PPLICTIONS: PROPERTIES ND LOSS COEFFICIENTS Material grade C94 R N87 FT-M 705M Material tye Mn-Zn Power Ferrites a Nanocrystalline morhous Ferroxcubtics Magne- Manufacturer Ecos Hitachi Metglas sat00 (T) Frequency <00 <500 <500 range (khz) 00 C m ( 0-4 ) Steinmetz Coefficients x y c T ( 0-4 ) c T ( 0 - ) c T a Minimum ower losses at C The design characteristics of the otimal designs obtained, named Maximum-Interleaved and Non-Interleaved resectively, are shown in Table III. When comared to the Maximum-Interleaved design, the volume of Non-Interleaved is 5% greater and the total losses at rated ower are % greater. For this alication, the otimal magnetic material is ower ferrite N87 for both designs and the otimal cores are double E with high, narrow windows. This is due to the fact that, as foil windings are used, an increase in the height of the windings leads to a decrease in winding losses, in other words it is ossible to increase the conductive area without increasing the losses due to the high frequency effects. However, the increased height entails an increase in the volume of the core since it has to enclose the windings, thereby increasing the core losses. There is therefore an otimal height at which the sum of the losses at the windings and core is minimal. In the Non-Interleaved design, in order to reduce the roximity effect losses, the thicknesses of the rimary and secondary foils are aroximately one fourth of the skin deth δ. Whilst, for the Maximum-Interleaved design, in which the roximity effect has been considerably reduced thanks to the winding configuration, these thicknesses are close to δ. Therefore, in order to obtain a reasonably low resistance R dc for the Non-Interleaved design a far greater winding height and window height/width ratio is required than for the Maximum-Interleaved design. Secifically, the otimum window height/width (ratio c/c) for the Non-Interleaved design has a disroortionate value of 7, whilst the Maximum-Interleaved design has a value of 4.4, which is a normal value for standardized cores. core such as the otimal one obtained for the Non- Interleaved design is not always readily obtained on the market and may not easily fit into the converter shell. Therefore, in order to areciate the real benefits of the maximum interleaving technique, this aer rooses a second non-interleaved winding design, named Non-Interleaved, with some core dimensions that are more commonly found on the market, but not otimal in te of efficiency. Reviewing the ossibilities available on the core market, we roose a maximum ratio of c/c of 6 to imlement the methodology. In this case, the imroved erformance resulting from the use of the maximum interleaving technique is logically greater. For the Non-Interleaved design, the volume is 78% greater and the losses are 9.5% greater than for the Maximum-Interleaved design. TLE III MXIMUM-INTERLEVED, NON-INTERLEVED ND NON-INTERLEVED DESIGN CHRCTERISTICS Design Characteristics Maximum- Interleaved Non- Interleaved Non- Interleaved Power density 8 W/cm W/cm 6 W/cm Volume, V e 80 cm 6 cm cm Core Windings Performance at rated ower (5 kw, τamb 50 C) Mag. Material Core Tye Ferrite EPCOS N87 Shell tye double E c/c/c 0.4/.75/.5 0.5/4/.5 0./.8/ a (mm) (T) N /N s 7.8/.5 0./6. 6./0 h /h s (mm) 0.4/0. 0./ /0. Fill factor, β P c 4.6 W 5.6 W 5.7 W P w 6.6 W 7. W 9.8 W P t 0.4 W.6 W 4.55 W R th 4.8 C/W.97 C/W.44 C/W Δτ max 50 C 50 C 50 C η % % 99.7 % It is also interesting to study the evolution of other design characteristics such as the fill factor, the magnetic induction amlitude, the rimary and secondary foil thicknesses, and the ower loss distribution between core and windings. In the first two designs, the core window is otimally shaed and, therefore, the window is fully filled with coer reflecting the fill factor the minimum sace required by the insulation. With regard to the last design indicated above, as the core is not of (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

9 0.09/TPEL , IEEE Transactions on Power Electronics 9 an otimum size, the window is not comletely filled with coer. greater fill, by increasing the foil thickness or the number of turns, would only lead to an increase in total losses. For the three designs, the magnetic induction amlitude is much lower than sat, as is usual in free-cooled ower alications [], [], and the rimary and secondary thicknesses are different. In the Maximum-Interleaved design, as the rimary winding only has one turn er section, it is not surrising that its thickness is close to the skin deth δ. In the secondary winding, as there are two turns er section, the roximity effect is greater and, therefore, its thickness is less than δ. ccording to [9], for this alication, the otimal thicknesses would be h =.7δ and h s =0.97δ. In actual fact, these values would be otimal if only the winding loss is taken into account. However, when the whole transformer is sized, the foil thickness affects the window width required and, therefore, the core volume, having a direct imact on its losses and on the total transformer volume. It is, therefore, logical that the foil thicknesses for the otimal transformer are less than the otimal values determined by [9]. The goodness of the aroximation carried out for the calculation of the winding loss in (6) is now assessed. It can be seen that, for the first harmonic, the winding thicknesses comly with the range for which the aroximation was carried out, h δ 50k. However, when evaluating the losses resulting from the third current harmonic, the following is obtained: h =.76δ 50k and h s =.4δ 50k. Therefore, a recise calculation of the losses resulting from this harmonic is now made and the results comared with the aroximated calculation. For the third harmonic, the exact losses for the rimary and secondary are 0.4 W in both cases, whilst the value obtained with the aroximation are 0.6 and 0.4 resectively. s reviously indicated, the aroximation overestimates the losses when the thickness exceeds the skin deth, so the design does not lose its validity. With regard to the ower loss distribution, an otimal ratio between magnetic and coer losses is sometimes roosed to minimize the total losses [] [4]. This otimal ratio is derived from a theoretical develoment in which the total losses are minimized with resect to only one design variable, being the number of turns in [4], or the equivalent magnetic induction amlitude in [] and []. oth cases give the same otimal ratio, equal to /y, which is 0.78 for the N87 ferrite used in this case. In the Maximum-Interleaved and the Non-Interleaved designs, the otimal ratios to obtain minimum losses are 0.7 and 0.74, resectively, both very close, but not equal to the conventional otimal ratio. The fact that this conventional ratio does not achieve the otential minimum losses is because the influence of only one design arameter is taken into account and, besides, no account is taken of the high frequency effects on the winding loss, which are also affected by the design arameters. In this aer, account is taken of the effects of high frequency on the windings and also the interdeendencies existing between the various design arameters. VI. PROTOTYPE SSEMLY ND EXPERIMENTL VLIDTION fter demonstrating the suerior erformance of the maximum interleaving transformer design in theory, in this section a rototye is constructed and then exerimentally validated. s shown in Table III, the analytical design methodology leads to a non-integer number of turns, which is difficult to imlement in ractice. Furthermore, the core and the foil conductor thicknesses must be adated to commercially available, reasonably riced roducts. Thus, the otimal design is now converted into a commercial one, with the characteristics shown in Table IV. s it is shown, the otimal design can be easily adated to the commercial environment, although, obviously, the exact characteristics of the theoretical design cannot be reroduced. The transformer assembly rocess, in which the roosed maximum interleaving technique is imlemented, is described below. Fig. 8 shows a longitudinal section of the transformer in which the width of the window is reresented with a scale of : and only one core drawn. Firstly, as /n is equal to.6, a rimary conductor must be wound together with two secondary conductors on the coil former, i.e. =. To do so, three conductors are cut, one for the rimary with a thickness of mm and a length equal to 8 times the MLT c, and two secondary ones with a thickness of 0.0 mm and a length that is 7 times the MLT c. Terminal P* is soldered to the start of the rimary conductor, terminal S* is soldered to the start of the first secondary conductor, and the terminal required for the taing is soldered to the second secondary conductor. The conductors are insulated from each other by means of adhesive tae comrising a olyester film and thermoset synthetic adhesive, called TECROLL. s the decimal art of /n is greater than 0.5, they are simultaneously wound with the rimary conductor located in the interior, until 6 comlete turns have been made, i.e. z=floor(/)=6. TLE IV DESIGN DPTED TO THE COMMERCIL ENVIRONMENT Equivalent Volume: 50 cm Efficiency (5kW): 99.8% Power density: 0 W/cm Magnetic core: Windings: Material: Ferrite R tye Foil thickness: h /h s (mm) Core shae: xee55/8/ 0.406/0.0 Dimensions: Turn numbers: N /N s c/c/c 0.6/.5/.66 8/ a 0.07 m Fill factor, β: 0.5 Flux density, : 0.4T Oerating characteristics at 5 kw: P c 4.44 W P w 5.97 W P t 0.4 W TP 4.47 K/W R th Δτ max 45 C Secondly, as is less than, the outmost conductor of the secondary is cut and terminal S is soldered to its end, i.e. = and the number of conductors to be cut - =. Then a further turn is made with the rimary conductor and with the remaining secondary conductor. t this oint, the end of the ta is soldered to the end of this secondary conductor. This ta series connects the end of the first secondary conductor with (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

10 0.09/TPEL , IEEE Transactions on Power Electronics 0 the start of the second secondary conductor. This gives turns for the secondary and 7 for the rimary, therefore an extra turn is given to the rimary in order to achieve 8 turns. Finally, the three magnetic cores are stacked, the windings are inserted in the window and each E is closed on the other, leaving no air ga. Thus, thanks to the maximum interleaving technique, the windings are fully interleaved with only ta required, instead of the tas required by the conventional interleaving technique. P* P The hot-sot temerature was monitored, and its steady state value was taken when the temerature variation remained below 0.5 C for at least 0 minutes. The ambient temerature τ amb in the climatic chamber was ket at 50 C throughout the test. Finally, the comlete transformer model is alied to estimate its steady state temerature τ estimated at different oerating ower levels P by means of an iterative rocess in which the emirical thermal resistance in (9) is used. Estimated and measured steady state temeratures are comared in Table V. From this Table, it can be concluded that the estimated and exerimental temeratures are in good agreement, showing a good accuracy of the emirical thermal model even at different temerature increases. The estimation error is always negative, meaning that the temerature is always overestimated. Furthermore, a maximum relative error of 4% at rated ower was recorded. In short, the selected models for the transformer and the roosed design methodology have been shown to work satisfactorily and to lead to an otimal design in te of ower density and efficiency. S* Fig. 8. Winding distribution: longitudinal transformer section with the window width in scale : and only one E core. Once the transformer had been assembled, its correct oeration was validated on the test bench shown in Fig. 9. The HF transformer was laced in a climatic chamber in order to reroduce the worst case ambient design temerature of 50 C. The transformer was connected to the intermediate stage of the PV alication shown in Fig. 7. Its measured secondary voltage and rimary current wavefo at rated ower are shown in Fig. 0. S () HF Transformer () PV Converter () Climatic Chamber (4) GILENT Data Logger (5) Oscilloscoe Fig. 9. HF transformer validation test bench. 0 /div TP Fig. 0. Transformer secondary voltage and rimary current wavefo. 5 4 TLE V TRNSFORMER ESTIMTED ND EXPERIMENTL TEMPERTURES P τ estimated τ hot-sot error (W) (C) (C) (%) VII. CONCLUSIONS This aer resents an innovative foil winding interleaving technique, termed maximum interleaving, which minimizes the winding loss and facilitates construction by minimizing the number of tas required. In addition, an otimal design methodology is develoed for medium-ower high-frequency foil transformers, directed at meeting the demands for high efficiency and low volume in order to obtain high ower densities. For its use in the design methodology, the most suitable transformer loss calculation and thermal erformance models in te of a trade-off between accuracy and simlicity are selected and reformulated. Then, the methodology is develoed and alied to the design of a 50 khz transformer for use in a 5 kw PV converter for both maximum interleaved and non-interleaved windings. The non-interleaved design has 78% higher volume and 9.5% higher losses than the maximum interleaving design. Secifically, the maximum interleaving design has an efficiency of 99.8% and a ower density of 8 W/cm. Finally, a rototye is assembled and, thanks to the maximum interleaving technique, the windings are fully interleaved through only one ta instead of the required with the conventional interleaving technique. The rototye is satisfactorily validated through exerimental tests, obtaining a maximum error at the estimated oerating temerature of 4% (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

11 0.09/TPEL , IEEE Transactions on Power Electronics CKNOWLEDGMENT The authors gratefully acknowledge INGETEM POWER TECHNOLOGY for their financial and ermanent suort, and Centro Politécnico Salesianos Pamlona for their suort in the rototye construction rocess. REFERENCES [] J. iela, U. adstuebner, and J. W. Kolar, DC DC Converter for Telecom lications, IEEE Trans. Power Electron., vol. 4, no. 7, , Jul [] H.-S. Kim, M.-H. Ryu, J.-W. aek, and J.-H. Jung, High-Efficiency Isolated idirectional C DC Converter for a DC Distribution System, IEEE Trans. Power Electron., vol. 8, no. 4, , r. 0. [] J. Everts, F. Krismer, J. Van Den Keybus, J. Driesen, S. Member, and J. W. Kolar, Otimal ZVS Modulation of Single-Phase Single-Stage idirectional D C DC Converters, IEEE Trans. Power Electron., vol. 9, no. 8, , ug. 04. [4] T. esselmann,. Mester, D. Dujic, and S. Member, Power Electronic Traction Transformer : Efficiency Imrovements Under Light-Load Conditions, IEEE Trans. Power Electron., vol. 9, no. 8, , ug. 04. [5] N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics: Converters, lications, and Design, rd ed., New York: John Wiley & Sons, 00, ch. 0. [6] M. K. Kazimierczuk, High-Frequency Magnetic Comonents, nd ed., Chischester, U. K.: John Wiley & Sons, 04, ch. 5 and. [7] W. G. Hurley and W. H. Wölfle, Transformers and Inductors for Power Electronics: Theory, Design and lications, st ed., Chischester, U. K.: John Wiley & Sons, 0, ch. -6. [8] M. Pavlovsky, S. W. H. de Haan, and J.. Ferreira, Partial Interleaving: Method to Reduce High Frequency Losses and to Tune the Leakage Inductance in High Current, High Frequency Transformer Foil Windings, in Proc. IEEE PESC, 005, [9] D. C. Pentz, Overview of helical foil winding design for lanar magnetic comonents, 0 IEEE Int. Conf. Ind. Technol., 0, [0] Z. Ouyang and M.. E. ndersen, Overview of Planar Magnetic Technology Fundamental Proerties, IEEE Trans. Power Electron., vol. 9, no. 9, , Se. 04. [] C. W. T. McLyman, Transformer and Inductor design Handbook, rd ed., New York: Marcel Dekker, 004, ch [] R. Petkov, Otimum design of a high-ower, high-frequency transformer, IEEE Trans. Power Electron., vol., no.,. 4, Jan [] W. G. Hurley, W. H. W, and J. G. reslin, Otimized Transformer Design: Inclusive of High-Frequency Effects, IEEE Trans. Power Electron., vol., no. 4, , Jul [4] N. R. Coonrod, Transfonner Comuter Design id for Higher Frequency Switching Power Sulies, IEEE Trans. Power Electron., vol. PE-, no. 4, , Oct [5] C. P. Steinmetz, On the law of hysteresis, Proc. IEE, vol. 7, no.,. 97, Feb [6] S.. Mulder, Fit formulae for ower loss in ferrites and their use in transformer design, in Proc. PCIM, 99, [7] K. Venkatachalam, C. R. Sullivan, T. bdallah, and H. Tacca, ccurate rediction of ferrite core loss with nonsinusoidal wavefo using only Steinmetz arameters, in Proc. IEEE COMPEL, 00, [8] J. Mühlethaler, J. iela, J. W. Kolar, and. Ecklebe, Imroved Core- Loss Calculation for Magnetic Comonents Emloyed in Power Electronic Systems, IEEE Trans. Power Electron., vol. 7, no., , Feb. 0. [9] I. Villar, U. Viscarret, I. Etxeberria-Otadui, and a. Rufer, Global Loss Evaluation Methods for Nonsinusoidally Fed Medium-Frequency Power Transformers, IEEE Trans. Ind. Electron., vol. 56, no. 0, , Oct [0] T. Hatakeyama and K. Onda, Core Loss Estimation of Various Materials Magnetized With the Symmetrical / symmetrical Rectangular Voltage, IEEE Trans. Power Electron., vol. 9, no., , Dec. 04. [] P. L. Dowell, Effects of eddy currents in transformer windings, Proc. Inst. Electr. Eng., vol., no. 8, , ug [] J.-P. Vandelac and P. D. Ziogas, Novel roach for Minimizing High-Frequency Transformer Coer Losses, IEEE Trans. Power Electron., vol., no., , Jul [] M.. ahmani, S. Member, and H. Ortega, n ccurate Pseudoemirical Model of Winding Loss Calculation in HF Foil and Round Conductors in Switchmode Magnetics, IEEE Trans. Power Electron., vol. 9, no. 8, , ug. 04. [4] E. C. Snelling, Soft Ferrites: Proerties and lications, st ed. London, U. K.: Iliffe ooks Ltd, 969, ch.. [5] Ferroxcube Comonents, lication note on the design of low rofile high frequency transformers, 990. [6] EPCOS. G., lication notes. Ferrites and accessories, 006. [7] F. Musavi, M. Craciun, D. S. Gautam, W. Eberle, and W. G. Dunford, n LLC Resonant DC DC Converter for Wide Outut Voltage Range attery Charging lications, IEEE Trans. Power Electron., vol. 8, no., , Dec. 0. [8] J.-H. Jung, ifilar Winding of a Center-Taed Transformer Including Integrated Resonant Inductance for LLC Resonant Converters, IEEE Trans. Power Electron., vol. 8, no., , Feb. 0. [9] W. G. Hurley, E. Gath, and J. G. reslin, Otimizing the C Resistance of Multilayer Transformer Windings with rbitrary Current Wavefo, IEEE Trans. Power Electron., vol. 5, no., , Mar [0] M. Pavlovsky, S. W. H. de Haan, and J.. Ferreira, Reaching High Power Density in Multikilowatt DC DC Converters With Galvanic Isolation, IEEE Trans. Power Electron., vol. 4, no.,. 60 6, Mar [] W. G. Odendaal and J.. Ferreira, thermal model for high-frequency magnetic comonents, IEEE Trans. Ind. l., vol. 5, no. 4,. 94 9, Jul./ug Ernesto L. arrios (S ) was born in Pamlona, Sain, in 988. He received the.sc. and M.Sc. degrees (with honors) in Electrical Engineering from the Public University of Navarre, Pamlona, Sain, in 009 and 0, resectively. In 0, he joined the Research Grou in Electrical Engineering, Power Electronics and Renewable Energy (INGEPER) of the Public University of Navarre, where he is currently ursuing his Ph.D. His main research interests include high frequency magnetics, wide bandga ower semiconductor devices and ower converters for renewable energies, articularly for hotovoltaics and fuel cells. ndoni Urtasun (S ) was born in Pamlona, Sain, in 987. He received the M.Sc. degree in electrical engineering from the Public University of Navarre, Pamlona, Sain, and from the Institut National Polytechnique de Toulouse, Toulouse, France, both in 00. In 00, he joined the Electrical Engineering, Power Electronics and Renewable Energy research grou, Public University of Navarre, where he is currently ursuing his Ph.D. His research interests include ower electronics and renewable energies. lfredo Ursúa (M 04) received the.sc. and M.Sc. degrees, both with honors, in Electrical Engineering in 00 and 004, resectively, and the Ph.D. degree in Electrical Engineering in 00, all from the Public University of Navarre, Sain. In 00 he joined the Deartment of Electrical and Electronic Engineering at the Public University of Navarre, first as a researcher and since 00 as ssociate Professor. He is Vice Dean of the Technical School for Industrial Engineering and Telecommunications and member of the Steering Committee of the university Chair for Renewable Energies. He has been involved in several research rojects both with rivate and ublic funding, and mainly related to renewable energy systems, hydrogen technologies, ower electronics and electric microgrids. He has also co-authored more than 50 journal aers and conference contributions, and holds atents (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See

12 0.09/TPEL , IEEE Transactions on Power Electronics Dr. Ursúa is member of the IEEE and the Sanish Hydrogen ssociation. Luis Marroyo (M 04) received the M.Sc. degree in electrical engineering in 99 from the University of Toulouse, France, and the Ph.D. degree in electrical engineering in 997 from the Public University of Navarre (UPNa), Sain, and in 999 from the LEEI- ENSEEIHT INP Toulouse, France. From 99 to 998, he was ssistant Professor at the Deartment of Electrical and Electronic Engineering of the UPNa, where he currently works as ssociate Professor, since 998. He is the head of the Electrical Engineering, Power Electronics and Renewable Energy research grou (INGEPER). He has been involved in more than 60 research rojects mainly, in co-oeration with industry, he is the co-inventor of international atents and co-authored of more than 75 aers in international journals and conferences. His research interests include ower electronics, grid quality and renewable energy. Pablo Sanchis (M 0, SM ) received the M.Sc. and Ph.D. degrees (with honors) in Electrical Engineering in 995 and 00, resectively, and the M.Sc. degree in Management and usiness dministration in 994, all from the Public University of Navarra, Pamlona, Sain. From 996 to 998, he worked as a Guest Researcher at Delft University of Technology, The Netherlands. In 998, he joined the Deartment of Electrical and Electronic Engineering at the Public University of Navarra, where he is currently ssociate Professor. He is also the Head of the Chair for Renewable Energies of the university. He has been involved in more than 6 research rojects mainly in cooeration with industry, and is the co-inventor of 8 atents. He has also coauthored around 40 journal aers and more than 70 conference contributions. Dr. Sanchis is member of IEEE and the Sanish Hydrogen ssociation. His research interests include renewable energies, ower electronics, hydrogen technologies, electric grid integration and electric microgrids (c) 0 IEEE. Personal use is ermitted, but reublication/redistribution requires IEEE ermission. See