Design and Measurement of 2.5GHz Driver Amplifier for IEEE e Mobile WiMAX using a Small-Signal Method

Transcription

1 Design and Measurement of 2.5GHz Driver Amplifier for IEEE 82.16e Mobile WiMAX using a Small-Signal Method S. KASSIM, F. MALEK School of Computer and Communication Engineering, University Malaysia Perlis (UniMAP), No 12 & 14, Jalan Satu, Taman Seberang Jaya, Fasa 3, 2 Kuala Perlis, Perlis, Malaysia. mfareq@unimap.edu.my Abstract In this paper, the crucial and the needs for a highly linear driver amplifier for IEEE 82.16e Mobile WiMAX is being addressed. This was done by demonstrating a high linearity and low power driver amplifier which is adequate to accommodate a high Peak-to-Average Power Ratio (PAPR) and high efficiency requirements in Mobile WiMAX systems. A discrete E-pHEMT transistor is used to design a 5V, 3mA, 2.5GHz amplifier with a Small- Signal design method. The amplifier exhibits an Input and Output Return Loss (IRL & ORL) that are better than 1dB, Small Signal Gain (SSGain) of 12dB, Third-Order Intercept Point (OIP3) of 44dBm and Output 1-dB Gain Compression (OP1B) of 27dBm. The results demonstrate that it can provide a reasonable efficiency, linearity and good output power. Keywords IEEE 82.16e, High linearity, OIP3, PAPR, Driver Amplifier. 1 INTRODUCTION 1.1 The Needs for High Linearity in WiMAX Transmitter Similar to Power Amplifier (PA), driver amplifier is one of the key components in IEEE 82.16e Mobile WiMAX system. It needs some characteristics, such as high output power, high linearity and high power added efficiency (PAE) in order to support high Peak-to- Average Power Ratio (PAPR) results from complex digital modulation i.e. 64-Quadrature Amplitude Modulation (64QAM) used in Mobile WiMAX. This paper intended to highlights a 2.5GHz highly linear driver amplifier that is suitable for adaptation into Mobile WiMAX applications [2 & 5]. As for comparison, Mobile WiMAX needs to have a better linearity than IEEE (WLAN) in order to supply robust link at high data rates. Mobile WiMAX supports multiple modulation formats, including 64QAM, 16QAM and Quadrature Phase-Shift Keying (QPSK), with 64QAM systems demand the highest linearity performance. Since Mobile WiMAX standards have been enhanced with additional features, such as quality of service (QOS), the amplifier in transmitter chain require a very low levels of distortion for a given modulation, as defined by system Error Vector Magnitude (EVM). EVM is a measure of the distortion in a QAM constellation diagram and the resulting uncertainty (or error) for each point therein. The IEEE 82.16e Mobile WIMAX standard specifies that Relative Constellation Error (RCE) be held to -31dB, based on a 1% packet error rate. EVM and RCE are interchangeable terms since RCE describes EVM measurement that is calculated over an entire Fixed WiMAX frame. So as to achieve a robust link, Mobile WiMAX driver amplifier or PA must not exceed an EVM of 2.5% for 64QAM and 4% for 16QAM modulation [9]. 2 METHODOLOGY 2.1 E-pHEMT biasing The enhancement mode technology provides superior performance while allowing a DC grounded source amplifier with a single polarity power supply to be easily designed and built. As opposed to a typical depletion mode phemt where the Gate must be made negative with respect to the Source for a proper operation, an enhancement mode phemt requires that the Gate be made more positive than the source for normal operation. Biasing an enhancement mode phemt is as simple as biasing a Bipolar Junction Transistor (BJT). Instead of a.7v Base to Emitter Voltage (V be ), the enhancement mode phemt requires about a.6v potential between the Gate and Source, V gs, for the target drain current, I ds [1]. 2.2 Circuit Description DC biasing is accomplished by the use of a Resistive Voltage Divider network consisting of R 2 to R 9. The voltage for the divider is derived from the Drain Voltage (V d ) which provides a form of voltage feedback ISBN:

2 to help keep Drain Current (I d ) constant. At the input side, the combination of R 1 and C 7 enhance the ATF s stability by terminating the gate resistively at low frequency. L 1 and C 5 form the bias-decoupling network. In order to reduce circuit loss, L 1 should have high unloaded Q, (Q UL ) and operated below its Self Resonant Frequency (SRF). Capacitor C 5 is dimensioned for low reactance at the operating frequency (f opr ). Capacitors C 1 and C 2 form a capacitive tap matching for ATF-5189 s input. board. Multiple via-holes around ATF-5189 help to increase the thermal dissipation effectiveness of the evaluation board. 2.4 Linear Simulation An RF simulator from Agilent Technologies, Advanced Design System (ADS) allows the input and output matching networks to be designed accurately given the evaluation board was properly modeled i.e. dielectric materials and RF connector used were taken into considerations. In addition, critical parameters such as Rollet Stability Factor (K-Factor) and Small Signal Gain (SSGain) can be predicted during the preliminary design stage i.e. if simulation forecasts a strong tendency to self-oscillation, the designer can anticipate the problem by incorporating additional stabilization network into the preliminary circuit. Figure 1: 2.5GHz Driver Amplifier Schematic At the output, the ferrite bead, L 3, works in combination with C 8 to provide a resistive termination down to the tens of MHz range. Although a resistor can provide the same function, the power dissipation will be high. Inductor, L 2 and capacitor, C 6 form the biasdecoupling network. L 2 and C 6 are chosen with the same criteria as L 1 and C 5. Capacitors C 3 & C 4 form a capacitive tap matching for ATF-5189 s output. 2.3 Evaluation Board Design In order to demonstrate the performance, an evaluation board was designed with the material cost and actual space constraints were taken into considerations. The evaluation board was designed using.31inch (31mils) of FR-4 dielectric and 63 small surface mount components. RF connections to the evaluation board were made via PCB edge-mounted microstrip to SMA coax transitions, J 1 and J 2. Similar to bipolar transistors, which exhibit a wide variation in hfe within the same part number, the ATF-5189 s forward transconductance, g m, can also vary from unit to unit. For that reason, the resistor network, R 3 to R 7, on the evaluation board allows a finetuning the Gate Bias Voltage, V g, to be made in order to cover the range of g m variation. The individual PCB traces connecting to R 3 to R 7 is cut one at a time until the evaluation board draws the target current range of 315 ± 15mA. This would results 4.5V of V ds and 28mA of I ds at the device-under-test, ATF Two 3mm holes are provided for a heat sink to be mounted to the ground plane on the bottom side of the evaluation Figure 2: ADS Simulation of Input Matching Network Figure 3: Trajectories of input impedance during the various phases of matching design The trajectories of the input match can be systematically verified as shown in the Figure 3. The curve marked as 1 represents the initial impedance at the position of the first matching component, C 1. Subsequently, the addition of C 1 moves the input-side impedance along the constant resistance circle to 2. The shunt capacitor, C 2, shifts point 2 to the final position 3 near to the Smith chart centre whilst traveling along the constant admittance circle. The trajectories of the output match are shown in Figure 5. The curve marked as 1 represents the initial impedance at the position of the first output matching component, C 3. Subsequently, the addition of C 3 moves the trace along the constant resistance circle to 2. The last matching component, C 4 nudges the curve along the constant admittance circle to position 3 in the vicinity of the chart centre. ISBN:

3 Figure 4: ADS Simulation of Output Matching Network Small Signal Gain and Reverse Isolation (db) Small Signal Gain Reverse Isolation Figure 7: Small Signal Gain and Reverse Isolation Performance Figure 5: Trajectories of output impedance during the various phases of matching design 3 RESULT AND DISCUSSIONS 3.1 Measured Performance The evaluation board performance was measured under the following test conditions: V ds of 4.5V, I ds of 28mA and Operating Frequency f c of 2.5GHz. The gain was approximately 12.5 db in the middle of the pass-band. Slightly more gain can be obtained at the expense of higher cost by using high Q inductors and/or a PCB substrate with lower loss. 22dB of reverse isolation will be a good figure when cascading the ATF into the transmitter line-up. Any tuning or optimization on the amplifier s output will not significantly affect the input of the amplifier and viceversa. 48 Return Loss (db) OP1dB & OIP3 (dbm) OIP3 OP1dB -25 Input Retun Loss Output Return Loss Figure 6: Input and Output Return Loss Performance The ATF-5189 evaluation board amplifier exhibits good input and output return losses. Return losses of greater than 1dB of return losses as shown in Figure 6 will sufficiently minimizes detuning effects when the amplifier is cascaded with other stages in WiMAX transmitter chain. As for example, filters and antennas are commonly vulnerable to the adverse effects of reflective terminations. In view of that, designing the amplifier s input and output for a close match to 5Ω over the operating bandwidth, prevents unpredictable shift in the cascaded frequency response [8] Figure 8: OP1dB and OIP3 Performance The 1 db gain compression point, P1dB, indicates the upper limit of either the input or the output power level at which saturation has started to occur. Non-linear effects become increasingly prominent as the amplifier is driven to this limit. Linear modulation schemes like 64QAM OFDMA in Mobile WiMAX requires the power to be backed off several dbs from this limit [3]. The P1dB is measured by progressively increasing the input power while noting the point when the gain became compressed by 1 db. In transmit environment, P1dB is usually referred to the output. The evaluation board nominal output P1dB is approximately 27dBm as can be seen from Figure 8. The intercept point is another measure of ISBN:

4 amplifier s linearity. The intercept point is a theoretical point when the fundamental signal and the third order Intermodulation Distortion (IMD) are of equal amplitude is the third-order intercept point, IP3. The distortion level at other power levels can be conveniently calculated from the amplifier s IP3 specification. Two test signals spaced 5MHz apart with the power of -5dBm per signal were used for evaluating the ATF evaluation board. The large dynamic range between the fundamental tones and the intermodulation products meant that the latter is barely above the spectrum analyzer s noise floor. In order to measure the 3 rd order product amplitude accurately, a narrow sweep span need to be used to improve the signal to noise ratio performance of the analyzer. As a tradeoff from the narrow sweep span, only one fundamental and one 3 rd order intermodulation output signals can be practically displayed on the same graph. The IP3, referenced to the output, can be calculated from: where IM OIP3= Pout + (1) 2 P = P + SSGain (2) fund in and P out is the amplitude of either one of the fundamental outputs and IM is the amplitude difference between the fundamental tones (P out ) and the third intermodulation products [4, 6 & 1]. P out is the function of ATF-5189 s small signal gain and the input power applied during the two-tone test. Based on this formula, the output intercept point, OIP3, as illustrated is approximately 44dBm. demonstrates increasing gain corresponding with decreasing frequency. If this phenomenon is not tamed with the appropriate countermeasures, the amplifier can break into self or parasitics oscillation below its operating frequency - in the tens of MHz range. In order to assess the effectiveness of the low frequency circuit stabilization described previously, the Rollett stability criterion was calculated from the measurement of the evaluation board s S-parameters. The ATF-5189 evaluation board exhibits unconditional stability (K > 1) up to 2GHz which is sufficient to take the third harmonics into considerations. Small Signal Gain (db) Figure 1: Wideband Gain Sweep Unintentional coupling between the input and output of amplifier in transmitter chain plus component parasitic can lead to instability in the upper microwave region. If there are pronounced gain peaks above its operating frequency, the amplifier may oscillate under certain operating conditions [7]. In a wideband sweep of the ATF-5189 evaluation board up to 18 GHz, no abnormal peak was observed in the frequency response as shown in Figure 1. A reasonable 42% of PAE at P1dB was obtained as shown in Figure Figure 9: Rollet Stability Factor, K Like all microwave transistors, the ATF Pout 28 PAE 42 SSGain Input Power (dbm) Figure 11: PAE, Output Power and SSGain over Input Power at 2.5GHz PAE (%) ISBN:

5 4 CONCLUSION A 2.5GHz high linearity driver amplifier design for IEEE 82.16e Mobile WiMAX applications was demonstrated using a systematic Small-Signal S- Parameters design approach without relying to a typical large-signal i.e. load-pull data. A Small-Signal (linear) simulation, crucial factors that affects the driver amplifier performance i.e. component Q until evaluation board layout was thoroughly discussed. With a GaAs E- phemt transistor chosen, the Class A driver amplifier able to produce Input and Output Return Loss (IRL & ORL) that are better than 1dB across the 2.5GHz WiMAX band, a Small Signal Gain (SSGain) of 12dB, Third-Order Intercept Point (OIP3) of 44dBm and Output 1-dB Gain Compression (OP1B) of 27dBm. The amplifier not only having a high linearity performance but with low power (5V, 3mA) consumption, it should be able to cope for the high efficiency demanded in Mobile WiMAX i.e. CPE applications. In addition, a practical measurement method on the critical driver amplifier parameters such as P1dB and P1dB were methodically discussed. [9] Saleh A, Salz J. Adaptive Linearization of Power Amplifier in Digital Radio Systems. Bell Systems Technical Journal 1983; April; 62: [1] Vendelin, G. D., A. M. Pavio, and U. L. Rohde, Microwave Circuit Design Using Linear and Nonlinear Techniques, New York: John Wiley & Sons, 199. References [1] Application Note 1281, A High IIP3 Balanced Low Noise Amplifier for Cellular Base Station Applications, Agilent Technologies, February 22. [2] Bailey, M. J., Intermodulation Distortion in Pseudomorphic HEMTs and an Extension of the Classical Theory, IEEE Trans. on Microwave Theory and Techniques, Vol. 48, No. 1, January 2, pp [3] Crescenzi, E.J. Wood, S.M. Prejs, A. Pengelly, R.S. Pribble, W. A 2.5 Watt GHz Power Amplifier for WiMAX Applications using a GaN HEMT in a Small Surface-Mount Package, IEEE International Microwave Symposium, 27. [4] Cripps, S., RF Power Amplifiers for Wireless Communications, Norwood, MA: Artech House, [5] De Carvalho, N. B., and J. C. Pedro, Compact Formulas to Relate ACPR and NPR to Two-Tone IMR and IP3, Microwave Journal, December 1999, pp [6] Gonzalez, G., Microwave Transistor Amplifiers Analysis and Design, 2nd ed., Englewood Cliffs, NJ: Prentice Hall, 1997 [7] Ingruber, B., et al., Rectangularly Driven Class-A Harmonic-Control Amplifier, IEEE Trans. on Microwave Theory and Techniques, Vol. 46, No. 11, November 1998, pp [8] Leenaerts, D., J. van der Tang, and C. Vaucher, Circuit Design for RF Transceivers, Boston, MA: Kluwer Academic Publishers, 21. ISBN: