Fine Voltage Control Based on Frequency Separation Two-Degrees-of-Freedom Control for Single-Phase Inverter

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1 IEEJ Journal of Industry Applications Vol.5 No.6 pp DOI: /ieejjia Fine Voltage Control Based on Frequency Separation Two-Degrees-of-Freedom Control for Single-Phase Inverter Hitoshi Haga Member, Kenta Sayama Non-member Kiyoshi Ohishi Senior Member, Takayuki Shimizu Non-member (Manuscript received June 26, 2015, revised Feb. 29, 2016) Paper This paper proposes a new control system that facilitates two-degrees-of-freedom (2DOF) control to reduce the output voltage distortion in the self-sustained operation mode of a photovoltaic generation system. The proposed control system is configured by combining the disturbance observer using a notch filter (notch type disturbance observer) and a sinusoidal tracking controller. This configuration of the system facilitates high tracking performance and high disturbance suppression performance. This system is a new frequency-separation-type 2DOF control system that has the complete tracking performance of only the fundamental frequency, and has the desired recovery performance of the other harmonic frequency and the quick inserting load current. The numerical simulation results and experimental results confirm that the proposed control system reduces the output voltage distortion in an effective manner. Furthermore, the proposed control system is highly robust to the changes in the load. Keywords: single-phase inverter, sinusoidal tracking controller, disturbance observer, two-degrees-of-freedom control 1. Introduction A single-phase voltage type inverter is primarily used to convert a DC voltage such as a solar cell or a battery to a AC voltage. As an example of a voltage type single-phase inverter applications, a grid-connected inverter for photovoltaic (PV) generation system is enumerated. In most cases, the power conditioner consists of the combining a DC-DC converter and a single-phase inverter with an LC filter, which is connected to the utility grid, as shown in Fig. 1. Normally, the power conditioner runs at grid-connected operation mode (1). Meanwhile, if a power failure occurs due to a lightning strike or emergency, the power conditioner runs in a self-sustained operating mode. In both operating modes, a high-quality of the output waveform of a single-phase inverter is required.this paper focuses on a output voltage control system in the self-sustained operation mode. Basically, a proportional-integral (PI) or a proportionalintegral-derivative (PID) controller are used as controller of the single-phase inverter (2) (3). These controller has an integral element, it has the complete tracking performance on step reference. Therefore, these controller cannot track to sinusoidal reference completely. It has the high gains and carries out the almost complete tracking for sinusoidal reference. However, the high gain PI or PID controller often has large influence on the noise frequency and the other harmonic frequency. In order to obtain a high-quality output voltage waveform, Nagaoka University of Technology , Kamitomioka-machi, Nagaoka, Niigata , Japan SHARP Niigata Electronics Corporation 1310, Kamihachimai, Minami-ku, Niigata , Japan Fig. 1. Simplified structure of single-phase PV generation system the control system should achieve high tracking performance with respect to the sinusoidal reference. As conventional control method of the inverter to obtain a high-quality output waveform (3) (9), deadbeat control and repetitive control has been proposed (4) (5) (10) (11). Deadbeat control can realize highspeed response (4) (5). However, if the parameters of the main circuit are varied, the control system often shows vibrations in the response waveform. In a system with repetitive control, a band-pass filter is often used to ensure the stability of the system (10) (11). However, it is difficult to design the bandwidth or cutoff frequency of the filters, and this method often suffers from problems such as poor transient response. Meanwhile, a conventional 2DOF control system is configured with the PI controller and the disturbance observer (9) (12) (13), the tracking performance and disturbance suppression performance are determined by both PI controller and disturbance observer, respectively. Therefore, the 2DOF control system always must keep these two performance requirements independently. However, it is sometimes difficult for a conventional 2DOF control systems to realize and keep both these two performances separately and independently (14) (15). In order to overcome these problems, this paper proposes a new frequency separation 2DOF control by using the notch type disturbance observer and the sinusoidal tracking c 2016 The Institute of Electrical Engineers of Japan. 413

2 controller (16). The proposed sinusoidal tracking controller tracks completely only sinusoidal reference (50 Hz in this paper) independently with the disturbance observer. The proposed control system has a high versatile system for a singlephase inverter. Thus, the proposed control system is also possible to apply to grid-connected operation mode of the PV generation system. This paper confirms the validity of the proposed control system by the numerical simulations and the experiments, where the load comprised resistance and a rectifier. The various load changes are assumed in the self-sustained operating mode. Therefore, the high robustness of the proposed control system using disturbance observer is confirmed by experiments of the resistance load change. 2. Principle of Proposed Control System 2.1 PI Controller and Sinusoidal Tracking Controller In the control system of a single-phase inverter, the reference is a sinusoidal waveform. The regulator of the control system requires high tracking performance to the sinusoidal reference. Hence, the regulator should be designed to track only the fundamental frequency completely. A previous report (7) is mentioned about two conditions that the output of plant system tracks the reference. Two conditions are as follows. The closed-loop system is asymptotically stable. The open-loop transfer function of the system includes a mathematical model to generate the reference signal. A PI controller or a PID controller has been used as the conventional regulator in a single-phase inverter (2) (3). However, it is sometimes difficult for these controllers to track to the sinusoidal signal completely, because the mathematical model of both PI controller and PID controller has no sinusoidal waveform function and has only step function. These controller do not satisfy the complete tracking performance to the sinusoidal reference, it is often difficult for these regulators to design the controller gains. In order to track the sinusoidal reference, a sinusoidal tracking controller has been proposed (6) (8). Figure 2 shows a tested single-phase inverter circuit. This circuit consists of a single-phase inverter and an LC filter. L f and C f denote filter inductance and filter capacitance, respectively. V dc and i load represent the input voltage and the output current, respectively. v and i Lf are detected by the sensor. In this paper, the load has non-liner load (diode rectifier) or resistance load. When the load current i load has harmonics components, the output voltage v cf has harmonics distortion because of the output filter. The difference between v ref and v is controlled by the regulator, and its output is the voltage reference v inv.thegate signals are determined by comparisons with the v inv and the triangular carrier. Table 1 lists the parameters of the tested inverter circuit. These parameter use as a condition of numerical simulations and experiments in Sects. 4 and 5. Figure 3 shows the control block diagram of the conventional controller. The reference signal of the control system is a sinusoidal waveform of 50 Hz. Figure 3(a) shows the control system based on PI controller (2). The transfer function of the system shown in Fig. 3(a) is given by Fig. 2. Specification of tested IGBT single-phase in- Table 1. verter v v ref Fig. 3. Tested inverter circuit of self-sustained operation Input voltage V dc Output voltage v Output frequency Rated power PWM switching frequency Sampling frequency LC filter resonance frequency Dead time of inverter 195 V 100 Vrms 50 Hz 300 W 25 khz 25 khz 1.6 khz 2 μs Configuration of regulator in control system sk p + K i s 3 L f C f + s 2 (C f Ki Lf ) + s(k p +1) + K i (1) where K p, K i,andki Lf represent the proportional gain, integral gain, and the gain of the current i Lf current feedback, respectively. Figure 3(b) shows the control system based on sinusoidal tracking controller. z 1, z 2, i Lf,andv denote the state variables, respectively. f 1 to f 4 represent the state feedback gains. The optimal mathematical model G s (s) in the sinusoidal tracking controller is as follows: s G s (s) (2) s 2 + ω 2 o where ω o represents the angular frequency. This paper uses a cosine-type mathematical model of the sinusoidal tracking controller. As the proposed sinusoidal tracking controller is a standard feedback control system, its desired response performance is determined by the state feedback gains f 1 to f 4. Therefore, the proposed sinusoidal tracking controller has the cosine-type mathematical model and determines its desired response performance by f 1 to f 4. The state equation using 414 IEEJ Journal IA, Vol.5, No.6, 2016

3 Table 2. Gains of PI controller and sinusoidal tracking controller Proportional gain PI controller Integral gain Feedback gain Ki Lf Feedback gain f Sinusoidal tracking controller Feedback gain f Feedback gain f Feedback gain f the state variables of the system is expressed as where ẋ(t) Ax(t) + Bu(t), y Cx(t). (3) A 0 ω L f C f 0 C [ ], x(t), B z 1 z 2 i Lf v L f 0 The transfer functions in Fig. 3(b) are obtained in Eqs. (4) and (5). v v ref f 1 s + f 2 d 4 s 4 + d 3 s 3 + d 2 s 2 + d 1 s + d 0 (4) v (s 2 + ω 2 o )( L f s + f 3 ) (5) i load d 4 s 4 + d 3 s 3 + d 2 s 2 + d 1 s + d 0 The denominator coefficients d 4 to d 0 denote as follows: d 4 L f C f (6a) d 3 f 3 C f (6b) d 2 L f C f ω 2 0 f (6c) d 1 f 1 f 3 C f ω 2 0 (6d) d 0 f 2 + ω 2 0 f 4ω 2 0 (6e) In this system, the state feedback gains are calculated using the coefficient diagram method (CDM) (17).Thesegainare parameters that determine the poles of the sinusoidal tracking controller. The stability indexes of the CDM are standard values, which is based on the previous reports (15). Table 2 lists the parameters of PI controller and tracking controller. In this paper, the poles of the tested PI controller and the tested sinusoidal tracking controller are set at 6280 rad/s. The inverter switching frequency is 25 khz. It is preferable for the poles for controllers to set 2.5 khz or less for the stable operation of the system. It is necessary to make the pole the same value to compare the control methods. Figure 4, shows the frequency characteristics of closedloop transfer function on the sinusoidal tracking controller and the PI controller, whose poles are 6280 rad/s bytradeoff condition of the experimental system optimally. When the pole set to large value, the inverter reduce the distortion of the output voltage. However, the system is easy to unstable by a noise. The sinusoidal tracking controller and the PI controller in this paper keep 0 db on the fundamental frequency 50 Hz in the closed-loop transfer function. Using Fig. 4. Frequency characteristic of closed-loop transfer function of system Fig. 5. Frequency characteristic of open-loop transfer function of system Fig. 6. Principle of Proposed control system the PI controller causes a phase delay at the fundamental frequency, whereas using the sinusoidal tracking controller does not cause a phase delay at the fundamental frequency. The input of the control system is a step reference, the PI controller is preferred because this regulator respond to the entire frequency. If the input of the control system is the sinusoidal reference, the sinusoidal tracking controller is preferred because this regulator responds to only the fundamental frequency. Therefore, in order to track to the reference signal completely, the regulator must meet the mathematical model with respect to the reference. Figure 5 shows the open-loop frequency characteristics. Figure 5 indicates that the gain characteristic has a peak value at the fundamental frequency when using the sinusoidal tracking controller. Thus, the sinusoidal tracking controller is suitable for the regulator of the system because this controller tracks the sinusoidal signal completely. 2.2 Configuration of Proposed Control System Figure 6 shows the principle of the proposed control system. v ref, v and, v inv denote the voltage reference of the control system, the output signal of the plant system, and the output signal of the regulator, respectively. i Lf represents the 415 IEEJ Journal IA, Vol.5, No.6, 2016

4 filter inductance current. The voltage reference v ref is a sinusoidal waveform at the fundamental frequency of 50 Hz. Therefore, the sinusoidal tracking controller requires completely track to the fundamental frequency. In control system of the single-phase inverter for a PV generation system, a DC frequency component and AC frequency components except for the fundamental frequency are disturbance. Therefore, these frequency components should be suppressed. Thus, the control system requires the enhanced disturbance suppression and high tracking performance to the sinusoidal reference. This paper defines the notch type disturbance observer as the total system of the disturbance observer and the complete suppression feedback loop of harmonic frequency without fundamental frequency, which includes the notch filter (16). The configuration of the suitable control system to satisfy both performances is shown in Fig. 6. A typical 2DOF control system configured such as PI controller and disturbance observer is designed for all frequencies and does not consider the frequency separation. The proposed frequency separation type 2DOF control system achieves high tracking performance and high disturbance suppression performance separately and independently, in comparison with typical 2DOF control system. Hence, this system differs from the typical 2DOF control system. 3. Details of the Proposed Control System based on Notch Type Disturbance Observer Figure 7 shows a block diagram of the proposed control system. The proposed control system combines the sinusoidal tracking controller and the notch type disturbance observer. The sinusoidal tracking controller regulates only fundamental frequency component of the output voltage (50 Hz in this paper). The output of the regulator v r almost has fundamental voltage component. On the other hands, the proposed notch type disturbance observer suppress only harmonics components in the output voltage. The output of regulator v dis has harmonics components. The load current i load is estimated by using a disturbance observer in which the inputs are the filter reactor current i Lf and output voltage v.the estimated load current is calculated by using disturbance observer as follows: i load i Lf sc f v g dis i load s + g dis g dis s + g dis i Lf sg dis s + g dis C f v î load g dis s + g dis (i Lf + g dis C f v ) g dis C f v (7) The estimated load current î load is shown in Fig. 7. The pole g dis of the disturbance observer can be set to independently and separately. In this paper, in order to confirm the performance of proposed system, the pole g dis of the disturbance observer is the same as the poles of the sinusoidal tracking controller and the PI controller. The notch type disturbance observer is based on the disturbance observer (12) (13), which estimates the output current i load. The output of the notch type disturbance observer v dis is added to the output signal of the sinusoidal tracking controller v r as the compensation value for the voltage dimension. Therefore, the estimated value excluding the fundamental frequency is feedback by multiplying the inverse model of the plant system. However, the order of the numerator is increased by the derivative elements. Thus, in order to stabilize the system, a first order low-pass filter G lpf (s) is added to the notch type disturbance observer. The transfer function of G lpf (s) and the notch filter G notch (s) are given by these equations. G lpf (s) g lpf (8) s + g lpf G notch (s) s2 + d2πδ fs+ (2π f o ) 2 (9) s 2 + 2πΔ fs+ (2π f o ) 2 where d, Δ f, f o,andg lpf denote the depth of the notch, the width of the notch, the center frequency of the notch, and the pole of the G lpf (s), respectively. The transfer functions of the system are expressed as these equations. v v ref v î load f 1 s 2 + ( f 2 + f 1 g dis )s + f 2 g dis (d 4 s4 + d 3 s3 + d 2 s2 + d 1 s + d 0 )(s + g dis) (10) (s 2 + ω 2 0 )(n 2 s2 + n 1 s + n 0 ) (d 4 s4 + d 3 s3 + d 2 s2 + d 1 s + d 0 )(s + g dis) (11) The numerator coefficients n 2 to n 0 in Eq. (11) denote as follows: n 2 L f (12a) n 1 f 3 L f g dis (12b) n 0 f 3g dis + g dis G fb (s) (12c) Fig. 7. Block diagram of frequency separation 2DOF control system by notch type disturbance observer 416 IEEJ Journal IA, Vol.5, No.6, 2016

5 (a) Resistance and rectifier (b) Resistance load change Fig. 9. Load circuits Fig. 8. Frequency characteristic of notch filter G notch (s) The denominator coefficients d 4 to d 0 in Eqs. (10) and (11) denote as follows: d 4 d 4 L f C f (13a) d 3 d 3 f 3 C f (13b) d 2 d 2 L f C f ω 2 0 f (13c) d 1 d 1 f 1 f 3 C f ω 2 0 (13d) d 0 d 0 f 2 + ω 2 0 f 4ω 2 0 (13e) where G fb (s) is the feedback elements of the notch type disturbance observer in Fig. 7. The transfer function of G fb (s) is expressed as G fb (s) G lpf (s)sl f G notch (s) g lpf s+g lpf sl f s 2 +d2πδ fs+(2π f o ) 2 s 2 +2πΔ fs+(2π f o ) 2 (14) Equations (10) and (11) indicate that the poles of the observer and the regulator are divided in the denominator polynomial. Therefore, the sinusoidal tracking controller and the disturbance observer are designed and constructed by using the separation principle. The construction of the proposed control system realizes the frequency separation type 2DOF control system. 4. Numerical Simulation Results The validity of the proposed control system is confirmed by the numerical simulation results using the tested IGBT inverter as shown in Fig. 2. The numerical simulation is implemented with sufficiently fine pitch width using the Runge- Kutta method with fixed integration pitch. The pitch width is set at 0.1 μs. The elements of the main circuit use ideal devices. All poles of the PI controller, the sinusoidal tracking controller, and the disturbance observer are set to 6280 rad/s by trade-off condition of the experimental system. In the proposed control system, the depth of notch d is determined to sufficiently suppress the frequency components except for the fundamental frequency. d is set at ( 60 db). By determining the notch depth, the approximate bandwidth of the notch Δ f is determined. Δ f issetat20hz. Further, the center frequency of notch f o is set at 50 Hz, and the poles of low-pass filter g lpf is set at 6280 rad/s. Figure 8 shows the frequency characteristics of the notch filter G notch (s). Figure 9 shows load circuits. A lot of diode rectifier circuits are connected as the load of the single-phase inverter. Hence, this paper uses non-linear load (diode rectifier circuit) shown in Fig. 9(a) as load to clarify the effectiveness of Table 3. THD of output voltage waveform in numerical simulation results THD [%] Ratio [%] PI controller Sinusoidal tracking controller Proposed control system proposed control method in the steady state. The load comprised resistance and a rectifier is used, as shown in Fig. 9(a), where the values are 290 Ω and 90 μf, respectively. The peak value of the inverter output current is set at 5 A based on the specifications of the tested inverter. The output power is set at 140 VA. The disturbance observer using the proposed control system is known as one of the robust control method. Therefore, the robustness of the notch type disturbance observer is confirmed by the resistance load change. The resistance load has a long conduction period of the output current than the diode rectifier circuit, and the influence of of the load change appears. Figure 9(b) shows the method of the load change, where the load is changed from 200 W to the full load of 300 W. 4.1 Load of Resistance and Rectifier Figures 10(a) 10(c) show the results of the numerical simulations using the PI controller, sinusoidal tracking controller, and proposed control system, respectively. These waveforms show the voltage reference v ref, the output voltage v, and the output current i load. As shown in Fig. 10(a), the tested PI controller has some phase error. Hence, the complete tracking could not be achieved, because this PI controller does not satisfy the complete tracking performance to the sinusoidal reference. By contrast, the complete tracking is achieved using the tested sinusoidal tracking controller and the tested proposed control system, as shown (b) and (c) in Fig. 10, respectively. The voltage ripples appear around every voltage peak because a rush currents flows to the load at the area. In order to reduce the voltage ripple, it is necessary to have highly switching frequency and to raise the pole of the controller. Table 3 summarizes the THD of the output voltage in the numerical simulations. Here, the definition of the THD is follow. V2 2 + V2 3 + V2 40 THD 100[%] (15) V 1 Where V n is n-order RMS value. The THD ratio of the output voltage is improved by 16% by using the proposed control system. Here, the definition of the THD ratio is follow. THDratio THD B V 1 100[%] (16) 417 IEEJ Journal IA, Vol.5, No.6, 2016

6 Frequency Separation Two-Degrees-of-Freedom Controlled Inverter Hitoshi Haga et al. Fig. 11. Numerical simulation results of resistance load change (increase of the load) and 1.5 ms, respectively. The characteristic of the decrease of the load and the increase of load is almost the same. Therefore, the robustness of the proposed control system is verified by the numerical simulations of resistance load change. Fig. 10. Numerical simulation results in case of resistance and rectifier Where T HDA is the THD of the proposed control system. T HDB is the THD of the PI controller or sinusoidal tracking controller or proposed control system in Table 3. Therefore, the effectiveness of the proposed control system is verified by numerical simulations. 4.2 Resistance Load Change Figures 11(a) 11(c) show the simulation results using the PI controller, the sinusoidal tracking controller, and the proposed control method, respectively. This paper carried out the resistance load change using the circuit parameter shown in Fig. 9 The waveforms indicate the voltage reference vcreff, the output voltage vc f and the output current iload. In this paper, the validity of the proposed control system is evaluated by the load current response time. The load current response time is calculated by that time before the differences between the voltage reference and actual voltage converging in a zero. In PI controller, the response time is calculated by that time before the differences between the voltage reference and actual voltage converging to constant. The load current response time in the case of using each control system are 4.0 ms, 3.5 ms, and 1.5 ms, respectively. Figures 12(a) 12(c) show the results with decrease of the load. The load current response time in the case of using each control system are 5.0 ms, 2.0 ms, 5. Experimental Results This paper confirms the effectiveness of the proposed control system by the experiments all results using the tested IGBT inverter as shown in Fig. 2. All of the tested control systems are implemented by DSP (TI TMS320C6713) software algorithm, whose sampling time is 40 μs. 5.1 Load of Resistance and Rectifier The experimental conditions and parameters for resistance and a rectifier are same as those used in the numerical simulations. Figures 13(a) 13(c) show the experimental results using the PI controller, the sinusoidal tracking controller, and the proposed control system, respectively. The waveforms indicate the voltage reference vcreff, the output voltage vc f and the output current iload. Similarly, using the tested PI controller, the complete tracking could not be achieved, as shown in Fig. 13(a). The experimental results of tested PI controller also has some phase error and some harmonic frequency. By contrast, the complete tracking is achieved using the tested sinusoidal tracking controller and the tested proposed control system, as shown in Figs. 13(b) and 13(c), respectively. Table 4 summarizes the THD of the output voltage in the experimental 418 IEEJ Journal IA, Vol.5, No.6, 2016

7 Frequency Separation Two-Degrees-of-Freedom Controlled Inverter Hitoshi Haga et al. Fig. 12. Numerical simulation results of resistance load change (decrease of the load) Table 4. THD of output voltage waveform in experimental results PI controller Sinusoidal tracking controller Proposed control system THD [%] Ratio [%] Fig. 13. Experimental results in case of resistance and rectifier results. The THD ratio of the output voltage is reduced by 16% by using proposed control system. The validity of the proposed control system is also confirmed by the FFT analysis results as shown in Fig. 14. Figure 14 shows the FFT analysis results of the output voltage in the case of using resistance and a rectifier. The FFT analysis results indicate the performance of PI controller, the performance of sinusoidal tracking controller and the performance of proposed control system, respectively. The system using the PI controller has small voltage amplitude at the fundamental component, in comparison with the proposed system. Therefore, the proposed control system realizes the fine tracking performance for the fundamental frequency reference. On the other hand, the control system using a PI controller has harmonics with large amplitude compared with the proposed control system. By contrast, the proposed control system decreases the amplitude of harmonics. The proposed control system has the fine disturbance suppression performance, in comparison with the system using the PI controller. Therefore, the effectiveness of the proposed control system is verified by the experimental results. 5.2 Resistance Load Change The experimental Fig. 14. FFT analysis of experimental results in the case of output voltage at the resistance and rectifier conditions and parameters for resistance load change are same as those used in the numerical simulations. Figures 15(a) 15(c) show the experimental results using the PI controller, the sinusoidal tracking controller, and the proposed control system, respectively. The waveforms indicate the voltage reference vcreff, the output voltage vc f and the output current iload. Using the tested PI controller and the tested sinusoidal tracking controller, the load current response time is 2.5 ms, as shown (a) and (b) in Fig. 15, respectively. By contrast, the load current response time using the tested proposed control system is 1.5 ms, as shown in Fig. 15(c). The improvement in the load current response time is 40%. Thus, the proposed control system improves the 419 IEEJ Journal IA, Vol.5, No.6, 2016

8 Fig. 15. (a)picontroller Experimental results of resistance load change load current response time when the resistance loads change. The proposed control system with a notch type disturbance observer enhances the robust control performance. 6. Conclusion In order to improve the output voltage distortion in a single-phase inverter, this paper a new frequency separation 2DOF control system which has a complete tracking performance of only fundamental frequency, and has the desired recovery performance on the other harmonic frequency and the quick inserting load current. As the sinusoidal tracking controller has the sinusoidal mathematical model of fundamental frequency, the sinusoidal tracking controller has a large peak gain of only fundamental frequency. The notch type disturbance observer has the complete suppression performance of the frequency bands without fundamental frequency. This paper verifies the effectiveness of proposed control system by using the numerical simulation results and experimental results. Using the proposed control system, THD of the output voltage is reduced by 16% in experiments that employs resistance and a rectifier. Furthermore, the current response time is reduced by 40% in load change experiments. References ( 1 ) Z. Guo and F. Kurokawa: Inverter Dead-Time Compensation and Control Scheme for Reducing Harmonic Distortion and Improving Conversion Efficiency, T. IEE Japan, Vol.130-D, No.1, pp (2010) (in Japanese) ( 2 ) T.B. Lazzarin, G.A.T. Bauer, and I. Barbi: A Control Strategy for Parallel Operation of Single-Phase Voltage Source Inverters: Analysis, Design and Experimental Results, IEEE Trans. Ind. Electron., Vol.60, No.6, pp (2013) ( 3 ) M. Pascual, G. Garcera, E. Figueres, and F. Gonzalez-Espin: Robust Model- Following Control of Parallel UPS Single-Phase Inverters, IEEE Trans. Ind. Electron., Vol.55, No.8, pp (2008) ( 4 ) T. Yokoyama, Y. Igarashi, T. Haneyoshi, and T. Izumi: A Study of Digital Instantaneous Value Control with Filter Capacitor Current Compensation for PWM Inverter, T. IEE Japan, Vol.123-D, No.5, pp (2003) (in Japanese) ( 5 ) P. Mattavelli: An Improved Deadbeat Control for UPS Using Disturbance Observers, IEEE Trans. Ind. Electron., Vol.52, No.1, pp (2005) ( 6 ) T. Kato, K. Inoue, and S. Kuroda: Sinusoidal Waveform Following Method for Digital Control of PWM inverter, T. IEE Japan, Vol.126-D, No.3, pp (2006) (in Japanese) ( 7 ) S. Fukuda and T. Yoda: A Current Control Method for Active Filters Using Sinusoidal Internal Model, T. IEE Japan, Vol.120-D, No.12, pp (2000) (in Japanese) ( 8 ) L.F.A. Pereira, J.V. Flores, G. Bonan, D.F. Coutinho, and J.M. Gomes da Silva, Jr.: Multiple Resonant Controllers for Uninterruptible Power Supplies A Systematic Robust Control Design Approach, IEEE Trans. Ind. Electron., Vol.61, No.3, pp (2014) ( 9 ) Y. Ito, M. Iwata, and S. Kawauchi: Digital Control Method Using Full- Order Observer on Three-Phase Inverter for UPS, T. IEE Japan, Vol.113-D, No.12, pp (1993) (in Japanese) (10) K. Inazuma, K. Ohishi, H. Haga, M. Sazawa, and S. Kondo: High Power Factor Control Regulating Inverter Output Power in IPM Motor Driven by Inverter System without Electrolytic Capacitor, T. IEE Japan, Vol.131-D, No.7, pp (2011) (in Japanese) (11) P. Mattavelli, L. Tubiana, and M. Zigliotto: Torque-ripple Reduction in PM Synchronous Motor Drives Using Repetitive Current Control, IEEE Trans. Power. Electron., Vol.20, No.6, pp (2005) (12) T. Miyazaki and K. Ohishi: Design Method of Robust Stable Two-Degreesof-Freedom Control System Based on Disturbance Observer and Coprime Factorization, T. IEE Japan, Vol.117-D, No.5, pp (1997) (in Japanese) (13) K. Ohnishi, M. Shibata, and T. Murakami: Motion Control for Advanced Mechatronics, IEEE/ASME Trans. Mechatronics, Vol.1, No.1, pp (1996) (14) T. Yokoyama and A. Kawamura: Disturbance Observer Based Fully Digital Controlled PWM Inverter for CVCF Operation, IEEE Trans. Power Electron., Vol.9, No.5, pp (1994) ( 15) Y. Ito and S. Kawauchi: Microprocessor-Based Robust Digital Control for UPS with Three-phase PWM Inverter, IEEE Trans. Power Electron., Vol.10, No.2, pp (1995) (16) K. Sayama, S. Anze, K. Ohishi, H. Haga, and T. Shimizu: Robust and Fine Sinusoidal Voltage Control of Self-sustained Operation Mode for Photovoltaic Generation System, in Proc. 40th Annu. Conf. IEEE Ind. Electron. Soc., IECON2014, pp (2014) (17) S. Manabe: Controller Design of Two-mass Resonant System by Coefficient Diagram Method, T. IEE Japan, Vol.118-D, No.1, pp (1998) (in Japanese) Hitoshi Haga (Member) received B.S., M.S. and D.Eng. degrees in energy and environmental science from the Nagaoka University of Technology, Nagaoka, Japan, in 1999, 2001, and 2004, respectively. From 2004 to 2007, he was a Researcher with Daikin Industries, Ltd., Osaka, Japan. From 2007 to 2010, he was an Assistant Professor with the Sendsai National College of Technology, Sendai, Japan. Since 2010, he has been with the Department of Electrical Engineering, Nagaoka University of Technology. His research interests include power electronics. Kenta Sayama (Non-member) received the M.S. degree in Electrical, Electronics and Information Engineering from Nagaoka University of Technology, Nagaoka, Japan in He was a Ph.D. student at Nagaoka University of Technology, Nagaoka, Japan, until His research interests include power electronics. 420 IEEJ Journal IA, Vol.5, No.6, 2016

9 Kiyoshi Ohishi (Senior Member) received the B.S., M.S., and Ph.D. degrees in electrical engineering from Keio University, Yokohama, Japan, in 1981, 1983, and 1986, respectively. From 1986 to 1993, he was an Associate Professor with Osaka Institute of Technology, Osaka, Japan. From 1993 to 2003, he was an Associate Professor with Nagaoka University of Technology, Niigata, Japan. Since August 2003, he has been a Professor at the same university. He is an administration committee member of the IEEE Industrial Electronics Society, the Institute of Electrical Engineers of Japan (IEEJ), the Japan Society of Mechanical Engineers (JSME), the Society of Instrument and Control Engineers (SICE), and the Robotics Society of Japan (RSJ). Takayuki Shimizu (Non-member) received the M.S. degree in electrical engineering from Niigata University, Niigata, Japan, in He is currently with SHARP Niigata Electronics Corporation, Niigata, Japan. His research interests include power electronics. 421 IEEJ Journal IA, Vol.5, No.6, 2016