Highly Linear GaN Class AB Power Amplifier Design

Transcription

1 1 Highly Linear GaN Class AB Power Amplifier Design Pedro Miguel Cabral, José Carlos Pedro and Nuno Borges Carvalho Instituto de Telecomunicações Universidade de Aveiro, Campus Universitário de Santiago Aveiro Corresponding author: Pedro Miguel da Silva Cabral Instituto de Telecomunicações Campus Universitário de Santiago Aveiro Portugal Phone: Fax: pcabral@av.it.pt Appropriate Conference Topics: 1. Solid State Devices and Circuits 3. High-Power Devices and Techniques 26. Wide Band Gap Semiconductor Devices

2 2 Highly Linear GaN Class AB Power Amplifier Design Pedro Miguel Cabral, José Carlos Pedro and Nuno Borges Carvalho Instituto de Telecomunicações Universidade de Aveiro, Campus Universitário de Santiago Aveiro Extended Abstract: Modern digital telecommunication systems demand a steady improvement of the RF frontend s performance in terms of bandwidth, power added efficiency, and signal fidelity. This is especially true in microwave power amplifiers for which many advances have been made public. In this respect, one of the most promising device technologies is the one based on wide band-gap materials like GaN HEMTs. These devices already offer power transistors capable of delivering very high output powers, which are expected to be accompanied by also interesting linearity figures. This paper presents a highly linear 2W Class AB GaN Power Amplifier. All design stages are explained, from the matching networks (input and output) up to the bias circuitry. For supporting the nonlinear design of the PA, a convenient GaN HEMT equivalent circuit model was especially built and integrated in a harmonic-balance simulator. To validate the design strategy, the measured power amplifier performance was compared with the one predicted by the nonlinear PA model. This way, its linear and nonlinear predictive capabilities could be studied. The linear predictions are illustrated with S-parameter data, while the nonlinear ones focused on CW output power, Gain, power added efficiency and two-tone intermodulation distortion values. The observed performance data and the simulated and measured results comparison were considered very good, fully validating, this way, the adopted PA design methodology.

3 3 Highly Linear GaN Class AB Power Amplifier Design Pedro Miguel Cabral, José Carlos Pedro and Nuno Borges Carvalho Instituto de Telecomunicações Universidade de Aveiro, Campus Universitário de Santiago Aveiro Abstract This paper presents a highly linear 2W Class AB GaN Power Amplifier. All design stages are explained, from matching networks up to bias circuitry. The obtained Power Amplifier performance is compared with the one predicted by a nonlinear device model especially built for these transistors. This model s linear and nonlinear predictive capabilities are studied. The former are illustrated with the comparison between measured and simulated S-Parameters while the latter with output power, power added efficiency and intermodulation distortion data. I. INTRODUCTION The deployment of modern digital telecommunication systems, with continuously increasing capacity, has demanded a steady improvement of the RF front-end s performance in terms of bandwidth, power added efficiency, PAE, and signal fidelity. This is especially true in microwave power amplifiers, PAs, for which many advances have been made public. In this respect, one of the most promising device technologies is the one based on wide band-gap materials like GaN HEMTs. Despite the recognized device processing infancy, it already offers power transistors of unbeaten breakdown voltages, therefore capable of delivering very high output power, Pout, figures [1]. Also significant is the high linearity provided by these GaN HEMTs. In fact, the observed valleys of intermodulation distortion, IMD, versus input drive level patterns, which have been frequently observed in class AB PA designs, constitute a great help in achieving the aimed compromise between nonlinear distortion and power added efficiency [2],[3]. Unfortunately, the critical dependence of these IMD valleys, the so-called large-signal IMD sweet-spots, on almost unsuspected issues like: out-of-band terminations [4], device s strong and mild nonlinearities [2],[3] and quiescent point (not unusually in ranges of only a few tenths of Volt) have raised the quality standards of common PA design methodologies and nonlinear device models. This paper addresses the design of a microwave GaN PA, paying particular attention to the prediction of small- and large-signal Pout and IMD. Section II introduces the most important design stages and Section III presents all tests made and compares the results obtained from measurements with the ones predicted by a nonlinear global model especially conceived for this kind of active devices. Finally, Section IV summarizes all the work done. II. 9 MHZ CLASS AB POWER AMPLIFIER DESIGN The active device chosen for our PA was a 2mm GaN HEMT on Si substrate, encapsulated in a standard high power microwave package, similar to the one used in [5]. Our PA design goals were to simultaneously optimize Pout, PAE and signal to intermodulation ratio. Gate voltage was selected to maximize signal to IMD ratio and, as it is widely known, an active device operating under class AB is able to provide the best compromise between linearity, efficiency and output power. Drain voltage was set to 2 V taking full profit of this wide band-gap device output voltage and current excursion capabilities. After a few V GS tests around V T (i.e., close to class B and AB) it became clear that best IMD performance could be achieved when the HEMT presented an IMD vs Pin pattern with double minima. This led to a quiescent point V GS1 = -4.2 V or 4% of I DSS. Maximization of Pout and PAE demanded a careful selection of the Cripps load-line and fine tuning of the even harmonics [6]. Fig. 1a shows the schematic used to determine the output matching network requirements in order to achieve drain constraints. Fig. 1b shows the simulated I DS vs V DS characteristics, for six different V GS biases and superimposed to it the desired and obtained drain load line.

4 4 R d L d_b L d Drain R 31 OUT_Match 5 Ω 1. C ds I DS (A).5 C MHz 18 MHz a V DS (V) b Fig. 1. (a) Schematic used to determine output matching network requirements. (b) Simulated I DS vs V DS characteristics, for six different V GS biases (-) and desired (--) and obtained (-x-) drain load line. A two-stub output matching network was designed to guarantee the calculated intrinsic 34 Ω load-line at 9 MHz central frequency and a short-circuit at the 1.8 GHz (2nd harmonic). Fig. 2 shows the simulated output match response at the drain from 9 MHz to 18 MHz. 18 MHz 9 MHz 34 Ω Fig. 2. Simulated output match response seen at the drain from 9 MHz to 18 MHz. After designing the output network, next stage was to conceive an input network capable of providing possible source matching and optimized gain without in-band instability. As it is known, that is important to compensate for the expected gain loss caused by the PA output mismatch. After this, a broad band stability analysis was conducted which showed potential problems at VHF. This was solved by the design of convenient lossy gate and drain bias networks. However, since it is known that the bias circuitry also determines the device terminations at the envelope frequencies and thus nonlinear distortion performance [3], they were retuned to guarantee very low impedances at most of the envelope bandwidth (4 MHz). Fig. 3 shows the simulated output match response at the drain from 3 khz to 4 MHz. 4 MHz 3 KHz Fig. 3. Simulated output match response seen at the drain from 3 khz to 4 MHz. In fact, it could be confirmed during the simulation, and then in the PA testing, that these low frequency terminations can either jeopardize IMD performance or even introduce undesired sideband asymmetries (a symptom of long term PA memory effects) [7]. The PA was then implemented in MIC technology using a RT/Duroid high frequency laminate with a ε r =1,2. Fig. 4a shows the complete PA schematic and Fig. 4b a photograph of the implemented amplifier board.

5 5 V GS V DS Input Output a Fig. 4. (a) Complete PA schematic. (b) Photograph of the implemented PA MIC board. b III. 9 MHZ CLASS AB POWER AMPLIFIER TESTING In order to validate our PA design, several experimental tests were conducted and the results thus obtained compared with the ones predicted by the model presented in [5]. A. Small-Signal S-Parameter Measurements The first PA test was a set of broad band small-signal S-parameter measurements. 2 S 11 (db) S 21 (db) 1-1 a b c Freq (GHz) Freq (GHz) Freq (GHz) Fig. 5. Measured (x) and modeled ( ) PA S11 (a), S21 (b) and S22 (c). S 22 (db) As seen in Fig. 5a, Fig. 5b and Fig. 5c, there is a reasonable good agreement between measured and modeled results. B. Large-Signal One-Tone Measurements. The second test step consisted in several CW experiments to evaluate Gain, Pout and PAE versus input drive level. As seen in Fig. 6a, the PA presents a 1dB compression point of 2 W with an associated Gain of 15 db, and a PAE of nearly 32 %. Compared to the model predictions, it is clear that the efficiency came somewhat lower than expected, while the output power and gain deviations were within the measurement error. Nevertheless, one remarkable result that should be pointed out is the correct prediction of the Gain versus Pin pattern, Fig. 6b, despite the rather complex behavior of gain compression followed by gain expansion to end up again in gain compression. This is a direct consequence of the selected bias point as it can be related to the double minima IMD pattern aimed at the PA design phase [3] Pout (dbm) a PAE (%) Fig. 6. (a) Measured (x) and modeled ( ) Pout and PAE under CW operation. (b) Measured (x) and modeled ( ) Gain vs Pin under CW operation. Gain (db) b

6 6 C. Large-Signal Two-Tone Nonlinear Distortion Measurements. In nowadays communication systems, the wide variety of distinct modulation schemes and wideband signals present a statistical amplitude distribution that is quite different from the one of a simple CW or two-tone excitation [8]. Therefore, tailoring the IMD versus Pin pattern is crucial. Fortunately, the GaN HEMTs under study showed a very flexible IMD control via V GS bias. In order to evaluate our device s model flexibility of accurately reproducing the dramatic changes observed on the IMD vs Pin pattern with bias variations, several two-tone excitations (tones centered at 9 MHz, with a frequency separation of 1 khz) were applied at our PA input, for three different V GS values (V GS1 = -4,2 V, V GS2 = -4,15 V and V GS3 = -4,1 V). All these gate voltages guaranteed class AB PA operation. Fig. 7a, Fig. 7b and Fig. 7c, show Pout and IM3 vs Pin for the three V GS values previously mentioned. Pout & IM3 (dbm) Pout & IM3 (dbm) a b c Fig. 7. Measured (x) and simulated ( ) PA Pout and IM3 vs Pin for V GS1 (a), V GS2 (b) and V GS3 (c). As seen from the data depicted in Fig. 7a, Fig. 7b and Fig. 7c, there is again a good agreement between the predicted and measured results. As V GS values are increased, it is possible to see a displacement in the minima position. In Fig. 7a there are two minima, Fig. 7b is similar to the previous one but now the minima are closer and in Fig. 7c they are overlapped. Therefore, controlling the IMD pattern represents an enormous advantage when designing a PA with linearity and efficiency constraints. Pout & IM3 (dbm) IV. CONCLUSIONS A 2W class AB power amplifier circuit was built and all goals and design stages explained. The test results were compared with the ones predicted by a nonlinear global model. Indeed, a remarkable good agreement between measured and simulated Pout, PAE and two-tone IM3, was obtained in a practical circuit. This represents an important achievement as GaN modeling studies are still in their early stages of development. ACKNOWLEDGMENT The authors would like to acknowledge Eng. João Paulo Martins for the development of the automatic measurement benches extensively used throughout this work, Nitronex Corporation for providing the GaN HEMT devices, Portuguese Science Bureau, F.C.T., for the Ph.D. grant Ref /22, given to the first author and financial support provided under Project POCTI/ESE/455/22 MEGAN. This work was also partially supported by EC NoE TARGET. REFERENCES [1] K. Joshin, T. Kikkawa, H. Hayashi, T. Maniwa, S. Yokokawa, M. Yokoyama, N. Adachi and M. Takikawa, A 174 W high-efficiency GaN HEMT power amplifier for W-CDMA base station applications, in Proc. IEEE International Electron Devices Meeting Technical Dig., pp , Dec. 23. [2] N. B. Carvalho and J. C. Pedro, Large and Small Signal IMD Behavior of Microwave Power Amplifiers, IEEE Trans. on Microwave Theory and Tech., vol. 47, No. 12, pp , Dec [3] J. C. Pedro and N. B. Carvalho, Intermodulation Distortion in Microwave and Wireless Circuits, Artech House, 23. [4] F. Palomba, M. Pagani, I. De Francesco, A. Meazza, A. Mornata, G. Procopio and G. Sivverini, Process-Tolerant High Linearity MMIC Power Amplifiers, in Proc. Gallium Arsenide Applications Symposium Proc., Munich, pp , Oct. 23. [5] P. M. Cabral, J. C. Pedro and N. B. Carvalho, New Nonlinear Device Model for Microwave Power GaN HEMTs, to be published in the IEEE Int. Microwave Symposium Dig., Fort Worth, Jun. 24. [6] S. C. Cripps, RF Power Amplifiers for Wireless Communications, Artech House, [7] N. B. Carvalho and J. C. Pedro, "A Comprehensive Explanation of Distortion Sideband Asymmetries", IEEE Trans. on Microwave Theory and Tech., vol. 5, No. 9, pp , Sept. 22. [8] J. C. Pedro and N. B. Carvalho, Designing Band-Pass Multisine Excitations for Microwave Behavioral Model Identification, to be published in the IEEE Int. Microwave Symposium Dig., Fort Worth, Jun. 24.