Off-line LED applications

Transcription

1 Off-line LED applications From PFC basics to constant current LED drive CompelFest 2013 EU Design Services Roberto Scibilia 1

2 AGENDA Review of Power Factor EU Specs and Energy Star Limits for lighting equipments: EN Class C Selecting the right topology: Buck in average current mode Buck + Charge pump Boost Buck + Voltage Feed Forward Continuous, Discontinuous and Transition Mode Boost Flyback Closing the loop on the output current DC current sensing Transformer on secondary side Peak current stabilizing Low and Mid-Power LED driver Portfolio 2

3 Review of Power Factor Power Factor is the Ratio of Real Power (Watts) to Apparent Power (RMS Volt-Ampere product) Real Power (W) PF Apparent Power ( VA) Power Factor has two components Vin Iin Displacement Factor (DispF) Distortion Factor (DF) DF I I DispF cos 1 rms 1 1 THD 2 Current Lags Voltage by (Displacement Factor) Power Factor PF is the product of DF and DispF Vin Iin PF 1 1 THD 2 cos Current has high harmonic content (THD) 3

4 Review of Power Factor Reactive currents, either capacitive or inductive, including reactive harmonics result in circulating currents and associated I 2 R losses in the power transmission system but do not develop power in the load. Loads presenting the AC line with high power factor minimize unnecessary power losses in the transmission system. Loads presenting the AC line with low current THD minimize losses and interference with adjacent loads. Ideal resistive loads have a power factor of 1.0 and generate no harmonics (THD = 0). Legacy incandescent lamps are nearly ideal resistive loads. (temperature coefficient of filament causes some distortion,. Line Voltage Line Voltage Volt/ Amp/ PF = 1.0 THD = 0 Volt/ Amp/ PF = 0.90 THD = 43.5% AC AC Line Current Line Current 4

5 E U Specs and Energy Star Classes EN sets harmonic content for any power supply sold in EU 4 classes D ~ personal computers and TVs C ~ lighting equipment B ~ portable tools A ~ everything else Energy Star Power supplies with greater than or equal to 100-W input power must have a true power factor of 0.9 or greater at 100% of rated load when tested at 115 V, 60 Hz 5

6 EN and Energy Star Limits Power Factor not important! Only harmonic currents are limited, up to 39th harmonic, nominal 230 VAC Limits depend on Class: Absolute limit in amps (Class A, B) Percentage of fundamental (Class C lighting) Amps/watt up to absolute max limit (Class D 75 W P in 600 W) Energy Star Power factor drives the limit here 6

7 Lighting limits: EN Class C 7

8 Lighting limits: EN Class C (P<25W) For class C equipment with an input power smaller or equal than 25W either: 1. The limits of table 3 (column two) apply 2. Or the third harmonic current shall not exceed 86% and the fifth harmonic current shall not exceed 61% of the fundamental current (for further details refer to the standard). 8

9 Fixing the maximum limits (P>25W) 1st, 3rd, 5th, 7 th, 9 th Harmonic (7 th is 180 deg. shifted) 9

10 Fixing the maximum limits Approximation with a simple waveform: A trapezoidal line 10

11 Fixing the maximum limits The trapezoidal line fulfill the class C limit since it s an approximation of the built waveform 11

12 Is it enough also for Energy Star? The THD of the waveform generated from the EN limits is ~ 11.5% Consequently the PF, assuming there is no displacement factor, is 94.7% This value is well higher than the 0.9 dictated from the Energy Star regulations BUT: how do we get this waveform? 12

13 Selecting the right Topology: Buck An average current mode Buck converter might do the job, but the conduction angle is not enough to fulfill Class C 13

14 Selecting the right Topology: Buck Peak current mode, sensing on Mosfet s current Constant output current: you get the smiling input current 14

15 Selecting the right Topology: Buck Constant output current and voltage = Constant Power The smiling current has even worse PF (~0.55) and bad THD Real Measurement 15

16 Solution with Feed Forward injection Absorbed current almost sinusoidal (PF>0.85) Low cost, no feedback loop Output current slightly dependant on input AC voltage (±20% on Vin translates into ± 10% on Iout) This dependence can be reduced by injecting a DC bias Vin 17

17 CCM, TM, DCM what s the difference? I AVERAGE Some very important differences: (a) CCM (b) DCM I PEAK I AVERAGE I PEAK I AVERAGE Ripple Current Drives filtering requirements, ac losses in magnetics Peak current Drives semiconductor stress, losses, peak flux in magnetics Frequency CRM is variable frequency Helps on EMI, but enters quickly the 150KHz lower limit (C) DRM 18

18 CCM Loss Analysis In CCM diode experiences large reverse recovery current PFC boost is the worst case for diode MOSFET losses are increased Ripple currents are small low ac losses low peak current stress RMS currents are lower conduction losses are lower L I L I Q Q1 (a) CCM I D C I L_ pk_crm Switch Current Diode Current V OUT I L_ pk_ccm IL_valley_ccm Switch Current Diode Current t (b) CRM t 19

19 TM and DCM Loss Analysis Peak current in TM is 2 x CCM (even higher in DCM) High ac losses High peak current stress on Mosfet and Diode TM and DCM benefit: the diode stars conducting always when the energy in the inductor is zero: no reverse recovery issue Reduced risk of MOSFETs failure due to shoot through by reverse recovery of the Diode RMS currents are higher than CCM conduction losses are higher Worst case in DCM 20

20 TM Current Loop TM current loop employs hysteretic type control Lower boundary is zero Upper boundary is set by multiplier No loop to design Simply chose sense resistor based on large signal considerations Need to sense when zero current crossing occurs Signal taken from already existing auxiliary winding Small inductance of TM is traded-off for increased filter size Use low losses cores and Litz wire, when possible ($$$) Variable frequency operation can help 21

21 CCM: Closing the loop The inner current loop corrects the Power Factor The outer voltage loop regulates the output voltage The input voltage feed-forward speeds-up the voltage regulation 22

22 Input Average Current: why don t we get PF=1? The current loop tries to stabilize a constant output voltage (or current) Since we have 100Hz ripple on output cap, the error signal (Verr) will have also that ripple This error signal will add distortion on the absorbed current while trying to regulate inside it The consequence is a reduced Power Factor and higher THD 23

23 Boost Topology: TM mode L I L I D V OUT I Q C Q1 I L_ pk_ccm IL CRM_PK Vin* sin(ωt) L *T ON (a) CCM (b) CRM IL_valley_ccm Switch Current Diode Current t I L_ pk_crm Switch Current Diode Current t Ripple current in CCM is small Peak current in CRM is 2x CCM If TON is constant, the peak input current is proportional to the sinusoidal input voltage The cycle-average input current is half the triangular switching waveform area resulting in a sinusoidal input current 24

24 Flyback Topology: TM mode If T ON is constant, the peak input current is proportional to the sinusoidal input voltage The cycle-average input current is the average of the switch current, which is NOT sinusoidal because T OFF depends on IL CRM_PK 25

25 Duty Cycle and Frequency vs. Time The output power is then calculated by integration of the transferred energy cycle by cycle 0 P OUT 1 2 L P I PKprim ( x) 2 F S ( x) dx F.S ( x) Duty_Cycle( x) x

26 Input Average Current: why don t we get PF=1? The average input current is NOT a half of the peak inductor current but it is averaged with duty-cycle I inavg ( x) 1 T ON ( x) 2 I PKprim ( x) T ON ( x) T OFF ( x) I0.333 inavgvmin ( x) I.inavg ( x) x x 27

27 Typical application LED Driving The loop is closed to the output current, so VSENSE pin needs only a bias voltage Primary aux. winding used for zero current switching Secondary aux. winding for biasing and sensing the reflected input voltage (Triac dimming) 28

28 UCC28810 Transition Mode Controller Suitable for Boost, Flyback, Sepic, Buck, as PFC controller and constant output current generator as Buck controller. 29

29 Closing the loop on the output current: Current sense resistor + op-amp + TL431 R5 senses the output current U1-B is a differential amplifier U1-A has a TL431 (pin 3) + opamp inside U3 transfers the error signal to the EAOUT pin of UCC28810 D9, Q3 provide the Bias voltage for U1: note the dots on transformer; the forward voltage is rectified, which is independent of the output voltage (variable from 40V to 120V) 30

30 Closing the loop on the output current: Current sense transformer + TL431 C6 + Llk of T1 define the resonant frequency (LLC) T3 senses the output current R11 contains the information of the output current U3 + U4 close the loop and create the error voltage, which modulates the switching frequency U2, D5 work as OVP 31

31 Closing the loop on the output current: Stabilizing the peak current on the output inductor (TM) R1 senses the switch current (same peak value of the inductor) The value is compared inside UCC28811 with a fixed reference An hysteretic mode allows the converter to work into Transition Mode (Bang-Bang modulation) The average output current is half of the stabilized peak 32

32 Closing the loop on the output current: Stabilizing the peak current on the output inductor (CCM) The peak current through R4 is compared to the internal maximum limit CCM and duty-cycle > 50% need slope compensation (R5) C4, D3, D4, supply the Bias 33

33 Flyback Loop Compensation: Power Stage Gain 34

34 Loop Compensation: Power Stage Gain Calculate the power stage gain at 95Vdc input and full load of a 80Vout@2A, Flyback converter Transformer primary inductance Lp=70uH and turns ratio 1:1 The input power is always: If the efficiency is 89%, Pin = 180W VIN PK = 120.2V, the 95Vdc is reached at 0.91 radians From the switching frequency graph we pick the equivalent frequency at 0.91 rad.: ~ 60KHz Flying back to the TL431, the total gain is: Gps = 30.5dB V IN ( t) 95V DC Pin 1 2 Lp Ipk2 Fsw V IN ( t) V INPK sin( t ) 35

35 Gain (db) Loop Compensation: Closing the loop Hz 18Hz, 17.9dB 18Hz, -17.9dB 160Hz 7.24Hz Frequency (Hz) Power Stage Compensation Total Open Loop Choose a crossover frequency << 100/120Hz Place the zero to gain enough phase margin 36

36 Loop Compensation: Closing the loop ΦEA = -180º amplifier) ΦINT = -90º same amplifier θz = Tang -1 (Fc/Fz) TL431 inverting input (error due to the integration of the Fc= crossover frequency, Fz = zero freq. θp = Tang -1 (Fp/Fc) - 90º Fp = pole frequency The phase margin without compensation is: Mφ = 360+ ΦEA + ΦINT + θp = 13.2º..not enough! We start choosing a pole at 160Hz (we loose here 6.4º) We need then a zero that has a phase lead, at least: θz = 75º - (13.2º - 6.4º) = 68.2º, where 75º is our ideal phase margin 37

37 Loop Compensation: Real Measurement The real measurement has only a slightly higher phase margin than calculated (85º instead of 75º) The crossover frequency and the -1 slope match the measurement 38

38 Efficiency (%) Test results on a 42W isolated LED string driver Output Voltage (V) 180Vdc 230Vdc 265Vdc 39

39 TRADITIONAL LINEAR DRIVING METHOD Driven by a constant current power supply with ballast resistors Unbalanced channel current Poor efficiency Driven by a constant voltage power supply with constant current linear LED drivers Balanced channel current, but Poor efficiency 40 40

40 DRIVING LED BY A CONSTANT CURRENT POWER SUPPLY Constant Current Power Supply Unbalance current through each channel V LED1 V LED2 VLED4 V LED5 V LED3 V LED6 Lowest current through the highest LED voltage channel Largest current through the lowest LED voltage channel 41 41

41 DRIVING LED BY LINEAR CONSTANT CURRENT DRIVERS Constant Voltage Power Supply Fix Rail Voltage V LED1 V LED2 V LED4 V LED5 V LED3 V LED6 Linear constant current driver V REG1 V REG2 V REG3 V REG4 V REG5 V REG6 Significant power dissipation across linear drivers =>Poor efficiency 42 42

42 DEVICE HIGHLIGHT LM3466 is a linear LED driver which acts like an intelligent ballast resistor. Each IC communicates with other IC s to equalize the current in each channel which derives from a constant current power supply, i.e., divides the current equally. Thus it is very easy to construct a high power lighting fixture by combining an off-the-shelf constant current power supply and LM3466 s

43 LM3466 OVERVIEW Works with a constant current power supply Equalizes the current of every active LED string automatically Maintains constant output power if some strings open (inactive), and the current of remaining active LED strings will be equalized automatically No communication to/from the constant current power supply is required Operating with minimum voltage overhead to maximize power efficiency (up to 99%) Wide operating voltage from 6-70V, drives up to 20 LEDs per string Up to 1.5A driving current Protections : Thermal shutdown, fault status output Linear circuitry does not deteriorate EMI Package: PSOP8 Target application : Street lamp, tunnel lamp, parking lot lamp, panel lamp 44 44

44 LM3466 BUILDING BLOCK 45 45

45 TYPICAL APPLICATION 46 46

46 Lighting Power Products Combined Portfolio 47

47 Performance AC-DC LED Driver Controller < 5W GU10, E14 Candles < 10W A19, PAR 20, downlights < 25W A19, PAR 20/38 < 50W PAR 38, Industrial Fixtures, Area Lights TPS92070 (Q3/11) TPS92210 Market Base Requirements TPS92010 Dimming, dissipative UCC28810 PF >80 % LM3445 LM3450 THD < 40% Eff > 65% LM3444 TPS92001 Integrated Dimmer Detection Output Power 48

48 TPS92070 LED lighting Driver controller System Block Diagram (Isolated driver) Valley Fill TRIAC Dimmer detect Near Lossless Dimmer Triggering Dimmer Angle detect 49

49 LM3445 TRIAC Dimmable Offline LED Driver Features TRIAC Dimming Decoder for LED Dimming Master/Slave Operation Application Voltage Range (80-277Vac) Controls LED Currents of Greater than 1A Adjustable Switching Frequency Adaptive, Programmable Off-Time Control Thermal Shutdown, UVLO, Current Limit Applications Dimmable Residential LED Lighting Drivers: A19 (E26/27, E14), PAR30/38, GU10 Lighting Applications: Light Bulb Replacement, Wall Sconces, Wall Washers, Architectural and Display Lighting, Commercial Troffers and Downlights Benefits Integrated TRIAC Detection Reduces Component Count and Solution Size Single TRIAC Controls Multiple Strings with Consistent Dimming Performance Supports Residential and Commercial LED Lighting Applications 50

50 LM3444 AC/DC Offline LED Driver Features Application Voltage Range (80-277Vac) Controls LED Currents of Greater than 1A Adjustable Switching Frequency Adaptive, Programmable Off-Time Control Thermal Shutdown, UVLO, Current Limit Applications Non-Dimming Residential LED Lighting Drivers: A19 (E26/27, E14), PAR30/38, GU10 Lighting Applications: Light Bulb Replacement, Wall Sconces, Wall Washers, Architectural and Display Lighting, Commercial Troffers and Downlights Benefits Supports Residential and Commercial LED Lighting Applications Supports a Wide Variety of LED Configurations 51

51 TPS92210 PFC Offline LED Lighting Driver Controller Features Flexible Operating Modes: Peak Primary Current, Constant On-Time, or both Cascoded MOSFET Configuration Works with TRIAC Dimmers Transformer Zero Energy Detection Discontinuous Conduction or Transition Mode Operation Advanced Over-Current Protection and Integrated Over-voltage Protection Applications Residential LED Lighting Drivers: A19 (E26/27, E14), PAR30/38, GU10 Lighting Applications: Light Bulb Replacement, Sconces, Wall Washers, Architectural and TPS92210EVM-647 Display Lighting, (110V) TPS92210EVM-613 (230V) Commercial TOOLS Troffers and Downlights Benefits Constant On-Time implements Single Stage Power Factor Correction (PFC) Fast start up; Line Surge Ruggedness Better Than Internal HV FET Continuous Exponential Dimming High Efficiency, Low EMI No Reverse Recovery Loss in Output Rectifier Protects Driver Against Fault Conditions 52

52 TPS92010 High Efficiency Offline LED Lighting Driver Controller Features High Efficiency LED Lighting Current Quasi resonant and low power modes High Performance TRIAC dimming with application circuit Programmable Overvoltage Protection Internal Over-temperature Protection TrueDrive Gate Drive 1A sink, 0.75A Source Current Limit Protection Cycle-by-cycle Power Limit Primary Side Over-current Hiccup Restart Mode Applications Residential LED Lighting Drivers Lighting Applications: Wall Sconces, Pathway, Overhead Lighting, TPS92210EVM-592 wall washing (110V) and display lighting TOOLS TPS92210EVM-631 (230V) Benefits 87% Achievable Efficiency Higher than Standard Flyback Topologies Less than 400mA Standby Power Allows Efficient Deep Dimming 20% More Efficient Dimming Compared with Other Methods Safely Shuts Down Driver if Open or Over Temperature is Condition is Present Lower Switching Losses Reduces System Cost Protects Driver from Fault / Abnormal Conditions 53

53 TPS92001/2General Purpose LED Lighting Driver Features Ideal for Single Stage LED Driver Designs Isolated and Non-Isolated Topologies TRIAC Dimmable Application Circuit with Low External Component Count Convenient 5V Reference Two Under-Voltage Lockout Options (10V or 15V) Integrated Gate Drive: 0.4A Source / 0.8A Sink Applications Residential LED Lighting Drivers: A19 (E26/27, E14), PAR30/38, GU10 Lighting Applications: Light Bulb Replacement, Wall Sconces, Wall Washers, Architectural and Display Lighting, Commercial Troffers and Downlights Benefits Power Factor >0.7 Supports Wide Configuration of LED Loads Low Cost Deep Dimming Solution with Small Form Factor Power for MCU or Linear Circuits Protection from Abnormal Operating Conditions External Gate Drivers Not Required 54

54 TPS92310 Features Description Single Stage Flyback AC-DC Controller: Primary side flyback LED current regulation Doesn t require opto-coupler or secondary side circuitry Adaptive ON-time control with inherent PFC Critical-Conduction-Mode (CrM) with Zero-Current-Detect (ZCD) for valley switching Reduces EMI filter design complexity LED current setting with external sense resistor No loop compensation required Gate driver with slew rate control Eases EMI filter design Output voltage protection (OVP) through ZCD VCC Under Voltage Lock Out (UVLO) Thermal Shutdown SOIC-10 package 55 55

55 Typical schematic (COT) 56 56

56 Typical schematic (PCM) 57 57

57 What is difference between COT and PCM operation. IAC VAC Constant ON time >> High power factor Peak Current Mode (Pull MODE1 pin to GND) >>Low output ripple current 58 58

58 Design consideration Fly wheel Shottky diode Snubber OVP Resistor Current sense Resistor 59 59

59 Design example and component selection 1. Snubber circuits: use 600V ultra fast diode. 2. Flywheel diode : use the 100V 200V 2A schottky. 3. ZCD / OVP resistor : Suggest set the Normal voltage is ~3.5V. Current is 1mA. 4. Current sense: IOUT = N*I REF / R CS N is turn ratio of transformer (primary : aux ): (3.8:1) REF = ma = 3.8 x 0.14 /1.5Ω. (*PS: 0.14 is Internal reference.) 5. Power MOSFET : 600V or upper. 2A e.g. 3N60E 6. TDLY resistor selection: 1. First step : 1/4 (TDLY )= 1/(2 Pi x SQRT(Lm x C )). Assume Cds= 200p, L = 5mH. 2. Calculation : TDLY = 2/(pi*SQRT(Lm x C) =636ns = 8K. 3. Based on the MOSFET drain to source waveforms. Fine tune RDLY

60 Design example and component selection 7. Pull up resistor (R4): The current through resistor R4 must less than 1mA. Otherwise OVP can t restart. (can not pull low by device itself) 8. Input capacitor (C2): C2 can t too small, We assume the VIN is constant within a switching cycle. If value of C2 too small. The input current can t estimate the output current correctly. For 220VAC the base is 0.15uF > C2> 0.1uF, The C2 too large will impact the power factor, too small will impact the current regulation. 9. Output capacitor (C3): C3 can smooth the IOUT current more smooth. Bigger C3 value can reduce the output current, but will impact the start up time. 430uF or 330uF are typical. 10. Diode (D7,D6): Diode D7 can limit the ZCD pin > -0.3V, it can avoid a negative current through the IC device. D6 is a zener diode for protection the VCC voltage

61 Design transformer consideration: Transformer:- Use a better couple transformer. Consider the transformer layout. e.g. use the interleaved windings. Interleaved winding has low eddy current loss. Full cover the bobbin length. Switch-node pin should be connect the transformer inside for reduce the EMI

62 Iout (A) TPS92310 line regulation (I LED = 350mA) Iout vs AC Input Voltage LIM_HI LIM_LO 0,450 0,425 0,400 0,375 0,350 0,325 0,300 0,275 0, AC Input (V) 63 63

63 I_LED (%) TPS92310 I_LED vs Temperature 3% I_LED vs Temperature 0% -3% Temperature ( C) 64 64

64 Input voltage and current IAC VAC PF = 0.91, Efficiency = 83% 65 65

65 Start up operation waveforms VCC VLED VIN IOUT Test condition: - Apply V DC = 0 to 250V DC, measure the output LED voltage and current. CH1 :- Controller input Voltage VCC CH2:- VLED voltage CH3 :- VDC Input DC Supply Voltage, CH4 :- ILED output current. Test result: - Normal, In upper start up case, LED light up time is 0.6sec. In typical operation, (220V AC ) start up time within 1.2 sec by current silicon

66 Typical operation waveforms V_Lx I_TX Test condition :- Apply 250V DC to light bulb supply pin, measure from drain to source voltage and Transformer primary side current. CH1 : NC, CH2 : NC, CH3: Drain to source voltage at external main MOSFET, CH4 : Main switch current, Test result :- Normal. The convertor can achieve CRM switching and zero current switch. Thus, design EMI and EMC filter are easy