ADVANCED METAMATERIAL CIRCUITS FOR MICROWAVE AND MILLIMETER WAVE APPLICATIONS

Transcription

1 ADVANCED METAMATERIAL CIRCUITS FOR MICROWAVE AND MILLIMETER WAVE APPLICATIONS By DAVID ELIECER SENIOR A DISSERTATION PRESENTED TO THE GRADUATE SCHOOL OF THE UNIVERSITY OF FLORIDA IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF DOCTOR OF PHILOSOPHY UNIVERSITY OF FLORIDA

2 2012 David Eliecer Senior 2

3 To my wife and daughter Chen Chen and Annabeth, my parents Lercy and Eliecer, and my sisters Aminta y Lercy 3

4 ACKNOWLEDGEMENTS This work was supported in part by the National Science Foundation (NSF) ECCS The following institutions in Colombia also supported my graduate studies: Universidad Tecnológica de Bolivar, COLCIENCIAS, Departamento Nacional de Planeación (DNP) and Fulbright Commission. I would like to start by thanking my chairman and advisor, Dr. Yong-Kyu Yoon, whose continuous support, encouragement and guidance provided me with many opportunities during these five years to expand my knowledge, improve my technical and communication skills and have a clear and insightful understanding of my research. Dr. Yoon has not only been an excellent mentor, but also a friend for me and my family, first at the University at Buffalo and then, at the University of Florida. This work would not have been possible without his guidance, support and patience. I would also like to thank the rest of my committee members Dr. David Arnold, Dr. Jenshan Lin and Dr. Peng Jiang for their valuable suggestions and their time to serve as the reviewers of this research. This work would not be possible without the support of the current and previous members of Multidisciplinary nano and Microsystems Laboratory that is part of the Interdisciplinary Microsystems Group (IMG). I would especially like to thank Dr. Jungkwun Kim, Xiaoyu Cheng, Pitfee Jao, Cheolbok Kim, Melroy Machado and Dr. Kyoung-Tae Kim for their continuous assistance on the simulation, fabrication and characterization of my devices, as well as the valuable technical discussions. I am thankful to Jessica Meloy who gave me the first training on the milling machine and has been very helpful with the organization and maintenance of IMG labs. 4

5 This work could not be possible without the assistance of the Nanoscale Research Facility members Al Ogden, Bill Lewis, David Hayes and Brent Gilla. I am grateful to all of them for their training on microfabrication and the continuous technical support. I would also like to thank my colleagues in Colombia, Professors Jorge Duque, Eduardo Gomez, Enrique Vanegas, Oscar Acuña, Gonzalo López, Jose Luis Villa and Ricardo Arjona for their academic support during this time. In the same way, I would like to thank the president of Universidad Tecnólogica de Bolivar, Dr. Patricia Martinez Barrios for her great vision of the future of the university and for giving me the opportunity and the financial support to study abroad. During the last year of my research, I have been especially supported and inspired by the love of my wife Chen Chen and my beautiful daughter Annabeth Nicole. They have brought the happiness and joy I need in the most difficult moments. Finally, I am grateful to my parents Lercy and Eliecer, my sisters Lercy and Aminta and my parents in law, Xiangliang and Guihua for their continuous love and support. 5

6 TABLE OF CONTENTS Page ACKNOWLEDGEMENTS... 4 LIST OF TABLES LIST OF FIGURES LIST OF ABBREVIATIONS ABSTRACT CHAPTER 1 INTRODUCTION Basic Concepts on Electromagnetic Metamaterials Towards Compact Practical Metamaterial Applications Motivation Research Objectives and Contributions Dissertation Organization ELECTROMAGNETIC METAMATERIALS Theoretical Prediction of Metamaterials Negative Refraction Phase Velocity and Group Velocity Experimental Demonstration of Metamaterials Negative Permittivity Medium Negative Permeability Medium Left-Handed Material Demonstration Transmission Line Approach of Metamaterials The Left-handed Transmission Lines The Composite Right/Left-handed Transmission Lines Periodic Structure Implementation Review of CRLH Transmission Line Applications Metamaterial Couplers and Filters Multiband Components

7 2.6.3 Metamaterial Resonators Compact Multilayer Components CRLH Substrate Integrated Waveguide Components Micromachined CRLH Applications MICROMACHINED TRANSMISSION LINES Bulk Micromachined Transmission Lines Surface Micromachined Transmission Lines Multilayer and Microfabricated CRLH Transmission Lines Applications Proposed Multilayer Unit Cell for the Implementation of CRLH Metamaterial Applications Proposed Multilayer Embedded Substrate Integrated Waveguide Filter Architecture Proposed Multilayer Architecture for Micromachined Wideband Bandpass Filters SU8 and BCB as Dielectric Materials for RF Circuits COMPOSITE RIGHT/LEFT-HANDED (CRLH) METAMATERIAL APPLICATIONS Compact Dual Band Three Way Bagley Polygon Power Divider Using CRLH Transmission Lines Compact Bagley Polygon Power Divider Dual Band CRLH Transmission Line Theory Review Design of the CRLH and Conventional Dual Band Quarter Wavelength Transmission Lines Implementation Summary Surface Micromachined CRLH Unit Cell on SU8 for Microwave Applications Unit Cell Structure and Modelling Implementation of CRLH Unit Cells Fabrication Process Measurement Results Summary Bridged Composite Right/Left-Handed Unit Cell with All-Pass Behavior Proposed Bridged-CRLH Analysis

8 4.3.3 Physical Implementation Summary MICROMACHINED METAMATERIAL UNIT CELLS ON BCB Analysis of Finite Ground Plane Coplanar Waveguide Transmission Lines on BCB Design and Implementation of the Dual Band CRLH Unit Cell and Transmission Line Loss Analysis Fabrication Measurement Results Summary METAMATERIAL LOADED THE HALF MODE SUBSTRATE INTEGRATED WAVEGUIDE Single and Dual Band Bandpass Filters Using CSRR Loaded Half Mode Substrate Integrated Waveguide Theoretical Backround Proposed CSRR Loaded Half Mode Substrate Integrated Waveguide Evanescent Mode Resonators Two Pole Filter Implementation and Measurement Results Summary Electrically Tunable Evanescent Mode Half Mode Substrate Integrated Waveguide Resonators The Tunable CSRR loaded HMSIW Resonator Implementation and Measurement Results Summary Dual-Band Filters Using CSRR and Capacitive Loaded Half-Mode Substrate-Integrated-Waveguide The Dual Band CSRR and Capacitive Loaded HMSIW Resonator Dual Band Bandpass Filter Design and Measurement Results Summary Wireless Passive Sensing Application Using a Cavity Loaded Evanescent Mode HMSIW Resonator The Evanescent Mode Resonator Proposed Sensor Structures

9 6.4.3 Effect of the Air Gap in the Resonance Frequency Mechanical Simulation of the Deflection Wireless Interrogation Measurement Results Summary MICROMACHINED EMBEDDED EVANESCENT MODE HMSIW BANDPASS FILTER D SU8 Embedded Resonator Design Two Pole Embedded Filter Design Fabrication Process Measurement Results Summary EVANESCENT MODE BROADBAND BANDPASS FILTERS Broadband Bandpass Filters using CSRR Loaded Eighth-Mode Substrate Integrated Waveguide Cavities General Requirements for Broadband Bandpass Filter Design The Eighth-Mode SIW Cavity The CSRR-loaded Eighth-Mode SIW Cavity Resonator Analysis and Design Two Pole Filters Designs Results and Discussion Summary Surface Micromachined Broadband Millimeter Wave Bandpass Filters Using CSRR Loaded Quarter Mode Substrate Integrated Waveguide Cavities The CSRR loaded Quarter Mode Substrate Integrated Waveguide Cavities The Design of CSRR loaded QMSIW Bandpass Filters on Flexible LCP The Design of CSRR loaded QMSIW Bandpass Filters on BCB Summary CONCLUSIONS Summary of Research Contributions

10 9.2 Future Work List of Related Publications APPENDIX : MICROMACHINED FABRICATION PROCEDURES ON BENZOCYCLOBUTENE LIST OF REFERENCES BIOGRAPHICAL SKETCH

11 LIST OF TABLES Table Page 3-1 Properties of the dielectric materials Parameters of the unit cell Summary of power divider measurements Design parameters of the MIM capacitors Design parameters of the inductor Parameters of the unit cells Dimensions of the unit cells in m Dimensions in m of the CRLH unit cell on BCB Dimensions of the proposed HMSIW-CSRR resonators Calculated parameters and dimensions of the two pole HMSIW-CSRR filters Parameters of the varactor diode Dimensions in mm of the resonator Summary of measurements results Dimensions of the resonator Parameters of the embedded filter Comparison of resonator size in different technologies Specification and calculated parameters of the filters Design specification for the filters on LCP Design parameters of the filters on LCP Design specifications and calculated parameters of the filters on BCB A-1 Curing of BCB

12 LIST OF FIGURES Figure Page 1-1 Wave propagation Demonstration of left-handed materials in free space Classification of electromagnetic materials Planar composite right/left-handed transmission line Overview of the dissertation structure Reflection and refraction of EM waves in the RH and LH media Negative permittivity medium Negative permeability medium Variation of the permeability of the SRR with the frequency Split ring resonator Left-handed materials in free space Modeling of a uniform right handed transmission line Modeling of a uniform left-handed transmission line Planar composite right/left-handed transmission line CRLH transmission line CRLH dual band components CRLH SIW slot antennas Cross section of some transmission lines used in microwave and millimeter wave circuits Bulk micromachined transmission lines Surface micromachined transmission lines Unit cells of multilayer CRLH transmission lines Implemented CRLH transmission line on SU

13 3-6 Multilayer CRLH unit cell on SU8 or BCB Cross section of the proposed dielectric embedded resonators and filters Cross section of the proposed micromachined cavity resonators and filters Conventional Bagley polygon three way power divider CRLH phase response Dual band quarter wavelength transmission line with shunt connections of open and short stubs Layout of the dual band CRLH /4 transformer Implemented dual band Bagley polygon power dividers Return loss (S11) and insertion loss (S21 or S41) for two implemented Bagley polygon power dividers Insertion loss at each port for the two implemented Bagley polygon power dividers The CRLH unit cell MIM capacitors Equivalent electrical circuit model of the two cascaded MIM capacitors Simulation of extracted parameters for the MIM capacitors Embedded meander line inductor Inductor modeling Extracted parameters for the inductor Complete equivalent electrical circuit of the CRLH unit cell including parasitic contributions General representation of the unit cell structure Comparison of the electromagnetic and circuital simulation for the broadband CRLH unit cell with a 90 phase at 2.4 GHz Fabrication process for the multilayer CRLH devices Photographs of the microfabricated broadband CRLH unit cell

14 4-20 Measurement setup consisting of a Cascade Microtech probe station and an Agilent E8361A VNA Insertion and return loss for a balanced microfabricated CRLH unit cell Dispersion relation for the microfabricated CRLH unit cell Proposed topology Physical configuration of the B-CRLH unit cell Simulated and measured results of the B-CRLH unit cell Triband two B-CRLH unit cells /4 open-stub Cross section view of the CPW structures Insertion loss of a CPW line on a low resistivity silicon substrate with a BCB interface layer Electric field in the cross-section for a CPW line with W = 55 m, G = 20m Multilayer CRLH unit cell on BCB Extracted parameters for the MIM capacitors on BCB Simulated performance of the unit cell Simulated loss factor of the unit cell for ideal dielectric, ideal conductor, lossless silicon, radiation loss and total loss Current distribution in the CRLH unit cell Fabrication process of the unit cell SEM Image of the CRLH unit cell Photomicrographs of the microfabricated devices Measured performance Measured phase constant of the two unit cells transmission line showing a dual band behavior around 12.8 GHz and 36 GHz Geometry of the rectangular waveguide Simulated electric field distribution for the TE 10 mode in a rectangular waveguide The substrate integrated waveguide

15 6-4 The half mode substrate integrated waveguide (HMSIW) Electric field distribution of the dominant TE mode A typical frequency response for the SIW and HMSIW structures Proposed HMSIW resonator with a series of vias for electric walls and complementary split ring resonator (CSRR) on the top surface Simulated results for a conventional split ring resonator Simulated results for a CSRR loaded HMSIW resonator External quality factor Q e of the CSRR loaded HMSIW resonator Internal coupling coefficient Extracted internal coupling coefficients Two pole bandpass filters Measurement and simulation results Fabricated resonators and filters The electrically tunable resonator Simulated results Photograph of the fabricated tunable resonators Measured results with applied DC voltage Measured results with applied DC voltage for the resonator C Dual band resonator Simulated performance of the dual band resonator Proposed dual band filter Simulated and measured performance of the implemented devices Photographs of the fabricated dual band devices Simulated and measured results for the resonator used in our study Proposed sensor configurations Variation of resonance frequency as a function of the air gap

16 6-29 Mechanical simulation for the deflection as a function of an applied pressure Broadband antenna Measured results Fabrication and test Electrical equivalent circuit of the implemented CSRR loaded HMSIW resonator Cross section of the proposed SU8 embedded resonator D view of the embedded resonator Simulated performance of the resonator External Q factor variation with the input distance l Top view of the two pole embedded filter Electromagnetic simulation results of the two pole filter Fabrication process Scanning electron microscopy (SEM) images of the embedded resonator and filter Measurement results The eighth-mode substrate integrated waveguide (EMSIW) Parameters of the EMSIW cavities Characterization of the EMSIW cavities Two-pole filters Measured and simulated results The quarter mode substrate integrated waveguide (QMSIW) cavities Extracted external quality factor (Q e ) of the resonator Extracted internal coupling coefficient k Simulated frequency response of a single ended CSRR loaded QMSIW cavity Physical layout of the bandpass filters on LCP Simulated results for the two pole bandpass filter on LCP

17 8-12 Simulated results for the three pole bandpass filter on LCP Proposed fabrication process of the LCP filters Extracted external quality factor of a CSRR loaded QMSIW cavity on BCB Extracted internal coupling coefficient for magnetically coupled CSRR loaded QMSIW cavities on BCB Layout of the proposed filters Simulated frequency response of the two pole 60GHz bandpass filter on BCB Simulated frequency response of the four pole 60 GHz bandpass filter on BCB

18 LIST OF ABBREVIATIONS BCB B-CRLH CRLH CSRR DNG EMSIW HMSIW LCP LH PCB QMSIW RH SIW SRR TL Benzocyclobutene Bridged- Composite right/left-handed Composite right/left-handed Complementary split ring resonator Double negative Eighth mode substrate integrated waveguide Half mode substrate integrated waveguide Liquid crystal polymer Left-handed Printed circuit board Quarter mode substrate integrated waveguide Right-handed Substrate integrated waveguide Split ring resonator Transmission line 18

19 Abstract of Dissertation Presented to the Graduate School of the University of Florida in Partial Fulfillment of the Requirements for the Degree of Doctor of Philosophy ADVANCED METAMATERIAL CIRCUITS FOR MICROWAVE AND MILLIMETER WAVE APPLICATIONS Chair : Yong-Kyu Yoon Major : Electrical and Computer Engineering By David Eliecer Senior December 2012 The exploration of various new architectures and fabrication techniques for implementing miniaturized metamaterial circuits for radio frequency (RF) applications is presented. The possibility of using low resistivity silicon, glass, liquid crystal polymer (LCP) and conventional printed circuit board (PCB) organic substrates for compact advanced metamaterial applications is addressed. The design, modeling, simulation and fabrication processes of the compact metamaterial devices are discussed in detail. As a first step, the composite right/left-handed (CRLH) approach for implementing metamaterials RF circuits is applied to the design of dual band applications using commercial lumped elements on a conventional PCB. Next, a multilayer surface micromachined fabrication process that utilizes the negative tone photopatternable epoxy SU8 and the negative tone photopatternable resin Benzocyclobutene (BCB) as dielectric interface layers on low cost organic carrier substrates, is employed for implementing highly compact CRLH transmission lines for broadband and dual band operation up to 40 GHz. Our study shows that, SU8 and BCB are good candidates for implementing compact metamaterial applications. The study continues with the implementation of compact resonators that make use of reduced mode versions of the substrate integrated waveguide (SIW) for narrow band, wideband 19

20 and dual band bandpass filters. The half mode SIW (HMSIW), quarter mode SIW (QMSIW) and eight mode SIW (EMSIW) loaded with a metamaterial particle, the complementary split ring resonator (CSRR), are proposed to implement compact bandpass filters working below the original waveguide cutoff frequency. Theoretical analysis and experimental demonstration are provided for bandpass filters working at S and X frequency bands on a conventional PCB substrate. Additional experimental implementations include a surface micromachined SU8 embedded CSRR loaded HMSIW bandpass filter working at 12 GHz. At the end, the proposed cavities are also applied for the design of a set of bandpass filters for operation at 25 GHz and 60 GHz using the flexible substrate LCP and the BCB resin as dielectrics. Finally, since the conventional printed circuit board (PCB), low resistivity silicon and glass are selected as the supporting substrates for the micromachined CRLH devices and filters; the compatibility with conventional microwave PCB implementations and CMOS integrated circuits is maintained. 20

21 CHAPTER 1 INTRODUCTION In the last decade, the scientific and engineering communities developed a great interest in a new area of study: Metamaterials. By definition, metamaterials are artificially created materials exhibiting unusual electromagnetic properties not readily found in nature. Mostly based on arrays of periodically organized structures, namely unit cells, their unusual electromagnetic properties - represented by the electric permittivity, the magnetic permeability and the refractive index n - make them interesting for devising new and creative applications for physics and engineering. Wireless communications greatly benefit from this new area, not only with the development of bulky metamaterials created to behave as conventional macroscopic materials, but also with planar and non-planar implementations of new applications working at microwave, millimeter-wave and terahertz frequencies. Moreover, modern wireless communication technologies demanding new applications, miniaturized devices with enhanced performance, integration with CMOS/MEMS/MMIC processes and the integration with digital circuitry, seem to be the motivation for the continuous exploration of new metamaterials circuits, systems and concepts in engineering. In this chapter basic concepts on electromagnetic metamaterials are visited. Section 1.1 introduces the electromagnetic metamaterials. Section 1.2 presents a review of the pioneer work on metamaterials for RF applications, emphasizing on the Composite Right/Left-handed (CRLH) Transmission Line (TL) approach for implementing planar and multilayer metamaterials. Section 1.3 discusses the motivation of this research. Section 1.4 provides the research objectives and, finally, section 1.5 summarizes the dissertation organization. 21

22 1.1. Basic Concepts on Electromagnetic Metamaterials The existence of metamaterials was first speculated in 1967 on the existence of substances with simultaneously negative and by Viktor Veselago [1]. In his paper, such kind of substances are called left-handed (LH) ones to express the fact they propagate electromagnetic waves in which the electric field, the magnetic field and the phase constant vectors build a lefthanded triad; in comparison with the conventional materials, also called right-handed (RH) ones, where the triad is right-handed, as illustrated in Figure 1-1. However, it was after three decades that Smith et al. [2] succeeded in the first demonstration of engineered left-handed materials by using negative permittivity thin wires (TW) and the negative permeability split ring resonators (SRR) previously proposed by Pendry et al. [3], but with limited number of unit cells and measurements in a waveguide environment. Later, Osbay et al. [4] performed the first experimental demonstration of left-handed materials in free space by using intercalated planar arrays of split ring resonators and thin wires, as illustrated in Figure 1-2. Although this resonant approach is lossy and narrow banded to be of practical interest for engineering applications, it opened a new research area for multiple and unique applications on microwave, millimeter wave, terahertz and optics, placing the concept as a good candidate for the next generation devices. A B Figure 1-1. Wave propagation. A) Right handed medium. B) Left-handed medium 22

23 A Figure 1-2. Demonstration of left-handed materials in free space. A) The left-handed structure consisting of planar implementations on printed circuit board (PCB) of intercalated arrays of SRRs and TWs. B) The transmission and reflection responses of the lefthanded or double negative (DNG) material. (Adaptation and reprint of Figures 1 and 4 from E. Ozbay, K. Aydin, E. Cubukcu and M. Bayindir, Transmission and reflection properties of composite double negative metamaterials in free space, IEEE Trans. Antennas and Propagat.,vol.51, no.10, pp , Copyright 2003, with permission from IEEE). To further understand the concept of metamaterials, Figure 1-3 illustrates the classification of a medium according to the possible sign combinations in the pair (, ). It is important to understand that the response of a system to the electromagnetic field is mainly determined by the properties of the constitutive material [5, 6]. These properties are described by the macroscopic parameters of the material: The electric permittivity and the magnetic permeability. These properties are related to the refractive index n by x y z B n, (1-1) r r where r and r are the relative permittivity and permeability related to the free space permittivity and permeability by 0 = / r = F/m and 0 = / r = H.m -1, respectively. For conventional right-handed materials, the refractive index n is positive, while for the left-handed materials it is negative. 23

24 Reflected II Plasmas below cutoff, Metals at high frequencies Wire structure Reflected wave I Conventional Dielectric materials Refracted wave Air Incident III Evanescent wave propagation < 0, > 0 n 0 Air Incident wave IV RH/forward-wave propagation > 0, > 0 n 0 Reflected Left-handed (LH) Materials < 0, < 0 Reflected Ferrites, Ferrimagnetic materials, Split ring resonators Air Incident Refracted LH/Backwardwave propagation n 0 Air Incident Evanescent wave propagation > 0, < 0 n 0 Figure 1-3. Classification of electromagnetic materials. (Adaptation of Figure 1 from C. Caloz and T. Itoh, Metamaterials for High-Frequency Electronics, Proceedings of the IEEE, vol.93, no.10, pp , Copyright 2005, with permission from IEEE). A medium with both positive permittivity and permeability ( > 0, > 0) is known as the Double Positive material (DPS) or Right-Handed material (RH), which is well known naturally occurring conventional isotropic dielectrics [5]. A medium with negative permittivity and positive permeability ( < 0, > 0) is known as the Epsilon Negative medium (ENG). Some plasmas and metals at optical frequencies behave in this manner. Conventional ferromagnetic and gyrotropic materials behave as a medium with positive permittivity and negative permeability ( > 0, < 0), which is called the Mu-Negative medium (MNG) [5]. Some DPS, ENG and MNG artificial materials exist in nature and were also demonstrated. A medium with 24

25 both negative permittivity and permeability is called as a Double Negative medium (DNG) or Left-Handed medium (LH), which is the material theorized by Veselago [1]. There is no clear evidence of this kind of material occurring naturally, but it can be constructed artificially, which can be categorized as a new class of materials. Such Left-Handed materials are classified as Metamaterials from the electromagnetic material viewpoint, where they are artificially fabricated in order to be effectively homogeneous (p < g /4) and exhibit unusual electromagnetic properties, such as negative permittivity, permeability and refractive index. In Chapter 2, a more detailed study of metamaterials is provided Towards Compact Practical Metamaterial Applications In the search for practical implementation of metamaterials for engineering applications, in 2002, the transmission line approach for implementing planar metamaterials on microstrip technology was proposed, almost simultaneously, by three different groups [7-10]. The approach led to the conception of planar negative-refractive index transmission lines or composite right/left-handed (CRLH) transmission lines, as shown in Figure 1-4. Due to their controllable unusual properties, such as non linear dispersion, broad bandwidth, infinite wavelength regime and positive phase shift, among others, this kind of transmission lines can be carefully designed to satisfy specific requirements depending on the particular application. After their introduction, many new different applications have been implemented by using either surface mounted devices (SMD) or integrated planar components, confirming the advantages of using negative properties in microwave engineering such as size reduction, backward propagation, bandwidth enhancement, zeroth and negative order resonances and multiband operation [11-34]. The need for additional size reduction and the integration with 3D and multilayer devices extended the transmission line approach to the implementation of super compact multilayer CRLH 25

26 transmission lines in the frame of conventional printed circuit board processes and ceramic LTCC processes, where MIM capacitors and meander lines are mostly used [26,27]. These well established PCB and LTCC technologies are greatly benefitted from the new CRLH architectures for miniaturized microwave device and system implementation. A Figure 1-4. Planar composite right/left-handed transmission line. A) Microstrip implementation of a 5 unit cells CRLH Transmission line. (Adaptation of Figure 8 from A. Lai, T. Itoh and C. Caloz, Composite right/left-handed transmission line metamaterials," IEEE Microwave Mag., vol.5, no.3, pp , Copyright 2004, with permission from IEEE). B) Unit cell (Reprint of Figure 5 from C. Caloz and T. Itoh, Transmission line approach of left-handed (LH) structures and microstrip realization of a low-loss broadband LH filter, IEEE Trans. Antennas Propagat., vol. 52, no. 5, Copyright 2004, with permission from IEEE). At the same time, the demand for miniaturized devices, higher operating frequencies and CMOS-MEMS integrable devices seems to be the motivation for using microfabrication techniques in the implementation of left-handed transmission lines. Qin et al. [32] successfully implemented left-handed transmission lines operating at the V band by using the negative epoxy photoresist SU8 as a dielectric. Tong et al. [33,34] successfully implemented left-handed metamaterial coplanar waveguide components and circuits based on the GaAs and high resistivity silicon MMIC technology. However, the use of GaAs and high resistivity silicon substrates, combined with oxygen plasma etching during the fabrication process, increase 26 B

27 fabrication costs. On the other hand, previous work has demonstrated that the use of thin and thick polymer dielectric materials such as SU8, BCB and Polyimide, allows the implementation of micromachined transmission lines and passive devices on CMOS grade low resistivity silicon substrates. This approach enables significant size reduction, decreases the losses associated with the low resistivity silicon substrate, avoids the use of expensive high resistivity silicon wafers, and creates a way to implement CMOS compatible devices in the microwave and millimeter wave range [35-40]. Since the printed circuit board (PCB) and LTCC processes are well established technologies while there are still many challenges in 3-D organic multilayer fabrication approaches, the microfabrication approach based on low resistivity silicon, glass or other low cost substrates with photopatternable low curing temperature dielectric interface layers have not been broadly explored and used for implementing advanced metamaterial applications for microwave range. Meantime, the advantages of this approach include the low substrate cost, the compact device size, the process compatibility with CMOS-MEMS processes, and the integrability with other components. In addition, the process can be easily scaled to the millimeter wave/thz range Motivation The end of the previous section suggests an opportunity to explore the implementation of compact metamaterial applications in the microwave and millimeter wave range. Metamaterial concepts are used for implementing of a broad variety of applications by using conventional PCB and LTCC processes. However, not so many applications have been demonstrated by using micromachining techniques on low cost substrates such as CMOS grade low resistivity silicon, glass and the flexible liquid crystal polymer (LCP). The motivation of our study is to explore the use of low resistivity silicon, conventional printed circuit board, flexible dielectric materials and organic substrates for achieving compact metamaterial applications that make use of advanced 27

28 new concepts, architectures, and fabrication processes. The design, modeling, simulation and fabrication process of highly compact metamaterial devices is addressed in our work. In addition to the conventional fabrication techniques based on the printed circuit board technology, we want to implement single layer and multilayer broadband and multiband metamaterial applications operating at microwave and millimeter wave frequencies by combining the original metamaterial concepts, namely the CRLH approach and the complementary split ring resonators (CSRR), with a surface micromachined fabrication process that utilizes different dielectric materials such as the negative tone photopatternable resin Benzoclyclobutene (BCB), the negative tone photosensitive epoxy SU8 and the flexible liquid crystal polymer LCP. On the other hand, we want to extend our study of metamaterial concepts to devising new compact resonators for wideband filter applications that make use of reduced mode versions of the substrate integrated waveguide (SIW) transmission line structures [41,43] loaded with complementary split ring resonator (CSRR). Conventional cavity resonators, implemented with either the metallic waveguide or the substrate integrated waveguide, are useful for narrow band filter applications due to their high achievable external quality factors. However, when wideband filters are required, i.e. filters with more than 5% fractional bandwidth, these cavities do not offer the required specifications for broadband operations, such as a low external quality factor and a high internal coupling coefficient between coupled cavities. To overcome these limitations, we propose in our work new in-substrate cavity resonators that are useful for wideband filter designs. The working principle, design, simulation and implementation of single, dual band, and tunable resonators and bandpass filters, operating under the principle of evanescent wave amplification, are demonstrated. Moreover, by using the same multilayer surface micromachined fabrication process in combination with the in-substrate waveguide concept, micromachined evanescent 28

29 mode filters are presented on SU8, LCP and BCB substrates. It is believed that the in-substrate nature of the resonators and filters can be extended to embedded implementations that allow conventional handling and packaging of 3D passive microstructures without additional mechanical consideration, which otherwise would require a very delicate and expensive vacuum packaging process [44]. Finally, photopatternable SU8 epoxy and BCB have been selected as the dielectrics for implementing embedded passive devices due to their optical, electrical and mechanical properties that offer great optical transparency, low curing temperature, the capability for high aspect ratio vertical interconnection useful for multilayer implementations, the compatibility and integrability with CMOS/MEMS processes, and the batch processability for multiple devices. It is expected that since the conventional printed circuit board (PCB), glass, low resistivity silicon and flexible LCP substrates have been selected as the supporting materials for the fabricated CRLH devices and filters, the compatibility with conventional PCB implementations and CMOS integrated circuits is maintained Research Objectives and Contributions The main objective of our study is the design and implementation of highly compact advanced metamaterial engineering applications for microwave and millimeter wave applications, which can be easily scaled in frequency. To achieve compact devices, the CRLH architecture and the complementary split ring resonator loaded on in-substrate waveguide structures are combined with a surface micromachined fabrication process that makes use of different organic substrates as dielectric layers. Multilayer Grounded Coplanar Waveguide (GCPW) and Finite Ground Coplanar Waveguide (FGC) composite right/left-handed balanced transmission lines are demonstrated for the design of multiband RF applications up to 40GHz. Compact photolithographically defined multilayer metal-insulator-metal (MIM) capacitors and polymer 29

30 embedded meander inductors are used to implement the left-handed transmission line contribution. The behavior of the CRLH transmission lines is evaluated by both 3D full structure electromagnetic simulations and experimental measurements. The electrical equivalent circuits are provided and the comparison with electromagnetic simulation and measurements is discussed. Further, the use of microfabrication techniques eliminates the necessity of using surface mounting device (SMD) based lumped components and makes the CRLH structures compatible and integrable with CMOS/MEMS processes. The optimized fabrication process is able to provide an easy way to use micromachining techniques for the implementation of miniaturized microwave and millimeter wave components containing metamaterial transmission lines and concepts. On the other hand, the study of devices that use metamaterial concepts is extended to the implementation of novel 3D integrable, compact CSRR loaded reduced mode substrate integrated waveguides structures for single, dual band and wideband bandpass filters by using the same dielectric materials used for the CRLH implementations. The combination of metamaterial concepts with the reduced mode version of the substrate integrated waveguide [41-43], allows the implementation of resonators with a resonance frequency below the characteristic waveguide cutoff frequency or the original cavity resonance frequency due to evanescent wave amplification [45-46], which offers a great size reduction since the resonator can be smaller than the quarter wavelength at the resonance frequency. Important contributions of our study include implementation of grounded coplanar waveguide (GCPW) and finite ground coplanar waveguide (FGC) balanced CRLH transmission lines on organic polymer substrates for broadband and multiband applications; the use of integrated and embedded passive components such as metal-insulator-metal (MIM) capacitors, meander line inductors and complementary split ring resonators (CSRR) for implementing multilayer metamaterial applications up to 40GHz; 30

31 the modeling of the CRLH transmission lines and metamaterial filters including dielectric and conductor losses, as well as the electrical equivalent circuit extraction procedure; the development of an optimized micromachining procedure based on SU8, BCB and LCP for implementing the proposed devices; the introduction of new in-substrate waveguide cavity metamaterial resonators for the design of wideband bandpass filters in the microwave and millimeter wave range; the introduction of a new CRLH architecture with all-pass behavior for future projects Dissertation Organization The dissertation is organized in six chapters as follows: Chapter 1 introduces the general literature review of the application of electromagnetic metamaterials to microwave and millimeter wave engineering. It also presents the motivation and research objectives. Chapter 2 provides the theory background on electromagnetic metamaterials, the transmission line approach for implementing planar metamaterial and the review of some pioneer work on CRLH transmission lines. Chapter 3 discusses previous work focusing on the technical challenges and the implementation of micromachined RF devices by using both conventional and CRLH transmission lines. Also, the proposed general multilayer architectures for CRLH transmission lines and substrate integrated waveguide devices are presented. Chapter 4 presents single band and multiband metamaterial applications that use conventional printed circuit board and micromachining techniques for the implementation. A dual band three way Bagley polygon power divider is implemented with CRLH transmission lines. Also, SU8 is used for implementing multilayer micromachined CRLH transmission lines working at microwave frequencies. At the end, a new CRLH architecture useful for all-pass operation is introduced. The new structure is studied both theoretically and experimentally in order to be used as a future work in our study. Chapter 5 explores the implementation of CRLH transmission lines by using BCB as a dielectric interface layer on a low resistivity silicon substrate. The design procedure 31

32 and the electrical, simulations and measurement results for a dual band micromachined CRLH unit cell are presented. Chapter 6 introduces the CSRR loaded Half Mode Substrate Integrated Waveguide (HMSIW) resonator for the implementing single and dual band evanescent mode coupled resonator filters. A tunable resonator, a wireless sensing application and a dual band filter that uses the evanescent mode resonator are also presented. Chapter 7 demonstrates a 3D SU8 embedded evanescent mode half mode substrate integrated waveguide resonators and filters fabricated by using the same micromachined process. Chapter 8 presents the novel proposed CSRR loaded reduced mode substrate integrated waveguide cavities for wideband filter applications, namely the Quarter Mode and the Eighth Mode SIW. A wideband filter is implemented on a conventional PCB by using the CSRR loaded Eighth Mode Substrate Integrated Waveguide (EMSIW). In addition, the design and simulation of wideband bandpass filters on LCP and BCB are presented for microwave and millimeter wave applications. Chapter 9 concludes the dissertation with a summary of the major results and contributions. Figure 1-5 summarizes the organization of this document. 32

33 Introduction Metamaterials concepts Compact metamaterials Motivation Objectives Organization Electromagnetic Metamaterials The transmission line approach Metamaterial Applications Fabrication Technology Micromachined Transmission Lines Micromachined Metamaterial Transmission Lines Proposed metamaterial architectures. Dielectrics choice Printed Circuit Board Metal Insulator Metal capacitors Embedded Inductors Multiple dielectric layers Electroplated vertical interconnections Flexible substrates Applications Dual band CRLH applications Multilayer CRLH Transmission Lines The Bridged-CRLH concept. Evanescent mode HMSIW filters. 3D embedded HMSIW filter. EMSIW and QMSIW micromachined filters. Conclusions Summary Future Work Micromachined Bridged- CRLH 3D embedded single and dual band evanescent mode filters. Figure 1-5. Overview of the dissertation structure 33

34 CHAPTER 2 ELECTROMAGNETIC METAMATERIALS Electromagnetic metamaterials are artificial effective homogeneous structures, not found in nature, with unusual and interesting electromagnetic responses [1]. Artificial complex materials with unusual properties have been an active research area since the first experiments were performed in the 19 th century [5]. It is believed in 1898 Jagadis Chunder Bose conducted the first microwave experiment with twisted structures that created what are called today artificial chiral elements [47]. Further in 1914, Lindel et al. worked on an artificial chiral media composed of randomly oriented small wire helices embedded in a host medium [48]. Kock in 1948 [49] performed experiments on compact microwave lenses that use periodically embedded metallic strips, wires and disks in order to control the refractive index of the artificial media. Modern experiments, based on new concepts, novel fabrication techniques and the periodic inclusion of novel miniaturized geometric shapes and forms in a host media, have developed new artificial electromagnetic structures and composite materials with similar properties to those of their known bulk counterpart, or in some cases, with new properties not readily available in nature. New concepts such as double-negative (DNG) materials, chiral metamaterials, split ring resonators and complementary split ring resonators, omega media, bianisotropic media, among others, have been the subject of research of many leading groups around the world. In a bulk composite host medium with inclusions, the electromagnetic waves induce electric and magnetic moments, which affect the transmission capabilities of the material and its constitutive parameters such as permeability and permittivity [5]. The artificial medium should be homogeneous in order to be characterized by the electrical permittivity () and the magnetic permeability (). In order to ensure that the medium is electromagnetically uniform along the direction of propagation of an electromagnetic wave, and thus, allowing refractive phenomena to 34

35 dominate over scattering/diffraction phenomena, the structural average cell size p, which is the size of the unit cell containing one inclusion, should be much smaller than a quarter of the incident guided wavelength g, p < g /4 [6]. This condition is called as the effective homogeneity limit or the effective homogeneity condition, and defines the medium as an effective homogeneous medium that behaves as a real material, in which the electromagnetic waves are essentially unaware of the lattice structure and the macroscopic parameters of the medium depend on the nature of the unit cell. This homogeneity relation is a rule of thumb effectiveness condition. The relation is often used in microwave engineering to distinguish lumped components (p < g /4) from quasi-lumped components, ( g /4 < p < g /2) and distributed components (p > g /2) Theoretical Prediction of Metamaterials The first theoretical work speculating the existence of substances with negative values of permittivity,, and permeability,, (left-handed materials), was introduced by Veselago [1] in In his paper, Veselago predicted a medium which would allow the propagation of electromagnetic waves with the electric field, E, the magnetic field, H, and the wave vector, k, building a left-handed orthogonal set or triad, in contrast with the well known right-handed triad in conventional materials. Left-handed materials (LHMs), also known as negative index materials (NIMs), or double negative materials (DNG), have simultaneous negative permittivity,, negative permeability,, and negative refractive index, n, over a specific frequency range. This new kind of speculated material would show interesting new electromagnetic properties not readily available in conventional naturally occurring materials, which are currently used for the creation of new applications in physics and engineering. Some of those properties include 35

36 frequency dispersion of the constitutive parameters and, which means the propagation constant is a nonlinear function of frequency; reversal refraction (Snell s law can be still applied); anti-parallel relation of the group and phase velocity; subwavelength focusing and imaging. In order to investigate the electromagnetic properties of left-handed materials, Veselago first studied how the electromagnetic waves propagate in a medium with both negative permittivity,, and permeability,. For wave propagation problems, the source-free Maxwell equations are given by B E t, (2-1) D H, t (2-2) D 0, (2-3) and, B 0, (2-4) where E (V/m) is the electric field intensity, H (A/m) is the magnetic field intensity, B (W/m 2 ) is the magnetic flux density and D (C/m 2 ) is the electric field density. In a linear ( and not depending on E or H) and non dispersive ( and not depending on frequency ), such as simple homogeneous isotropic dielectric and magnetic materials, the constitutive relations are as in D=E, (2-5) and B=H. (2-6) 36

37 Now in order to get valuable information on the fundamental response of the medium, a planar wave is considered [11], then the electric and magnetic field are given by E E e jr 0, (2-7) and H E0 e jr, (2-8) where = E/H represents the wave impedance. Now replacing previous Equations 2-7 and 2-8 in the Maxwell equations as in E=+H, (2-9) and H=-E, (2-10) the conventional solution for right-handed materials is obtained, where the vectors E, H and build the well known right-handed triad, as shown in Figure 1-1A. If a left-handed medium is considered ( < 0, < 0), then E= -H, (2-11) H= +E, (2-12) which creates the left-handed triad as previously shown in Figure 1-1B. It is known that the propagation constant is positive for a right handed medium (outward propagation from the source), which is not true for a left-handed medium, in which, according to the previous equations and the previously shown in Figure 1-1B, the propagation constant is negative (inward propagation to the source) and hence, the phase velocity v p is opposite to the phase velocity of a right handed medium as in RH medium : >0, v p > 0, (2-13) and LH medium : <0, v p < 0. (2-14) 37

38 The direction of the energy flow is given by the Poynting vector as in S=EH. (2-15) The non-zero Poynting vector, always forms a right-handed coordinate system with E and H, independent on the signs of and. Therefore, in a left-handed material, the wave vector,, is in the opposite direction of the energy density flow, S. Such a wave is called backward wave. In contrast, in a normal right-handed material (RHM), the wave vector,, and the energy flow, S, are in the same direction and the wave is a forward wave Negative Refraction The reflection and refraction of an incident plane wave on the boundary between two homogeneous media of different dielectric properties are well known classical problems in electromagnetism. As illustrated in Figure 2-1, an incident wave at the interface of two different media will generate a reflected wave and a refracted (transmitted). The refractive angle is determined by the Snell's Law of refraction: n. (2-16) 1 sin1 n2 sin In a left-handed medium, since the constitutive parameters and are negative, a negative refractive index, n < 0, is obtained: n= -n. (2-17) When considering an incident wave at the interface between a right-handed (RH) medium and a left-handed (LH) medium, a negative sign in the refractive index of the LH medium appears, which means a negative refraction angle is obtained, as illustrated in Figure 2-1. A more general form of the Snell s law is given by 1 n1 sin1 s2 n2 sin 2 s, (2-18) 2 38

39 where s is the sign of the refractive index: negative for a LH medium and positive for a RH medium. If the two media are left-handed, the Snell s law does not change due to the cancelation of the two minus signs of the refractive indices, which means that an incident wave on the boundary between two media with same handedness properties has the conventional positive refraction with positive refraction angle. Incident wave k Reflected wave S ' S i i k ' LH medium k ''' RH medium Transmitted wave in LH medium t t k '' Transmitted wave in RH medium S ''' S '' Figure 2-1. Reflection and refraction of EM waves in the RH and LH media. (Adaptation of Figure 3 from V. G. Veselago, The electrodynamics of substances with simultaneously negative values of, and, Sov. Phys. Usp.,vol. 10, Copyright 1968, with permission from IOPSCIENCE). It is observed in Figure 2-1 that the wave vector of the refractive wave in the LH medium, k ''', heads toward the interface, indicating that the refractive wave is a backward wave traveling towards the source. On the other hand, the Poynting vector, S ''', which represents the energy, heads away from the interface, which means that the group velocity v g, is antiparallel with the phase velocity, v p, as it was stated for the propagating waves in the left-handed medium. 39

40 2.3. Phase Velocity and Group Velocity It was previously mentioned that the phase velocity, v p, and the group velocity, v g, are in the opposite direction in left-handed materials. The phase velocity is defined by v p, (2-19) where is the unit vector along the propagation direction. Since the frequency,, is always a positive quantity, the relations in Equations 2-13 and 2-14 are valid for the RH and LH medium, respectively. The group velocity, v g, which is related to the Poynting vector, is given by v. (2-20) g Since the Poynting vector S depends only on the electric field vector E and magnetic field vector H, but not on the constitutive parameters and as the propagation constant does, it is oriented toward the direction of the energy over time and hence, is parallel to the group velocity. Then it is shown that the phase and group velocities are anti-parallel in the LH medium. As a summary, the following relations are valid : RH medium : >0, v p > 0, v g > 0 (2-21) and LH medium : <0, v p < 0, v g > 0. (2-22) 2.4. Experimental Demonstration of Metamaterials Although Veselago theoretically proposed the existence of left-handed materials in 1967, it was only recently after more than 30 years that such materials were implemented and demonstrated experimentally. The first experimental demonstration of left-handedness was based on an artificial effectively homogeneous material, instead of a naturally occurring substance, as expected by Veselago. The pioneering work of Pendry and his colleagues at the Imperial College, 40

41 London, UK [3], on microwave plasmonic structures showing negative-/positive- (metal thin wires) and positive-/negative- (split ring resonators) was the foundation for implementing artificial left-handed materials. Both of these plasmonic structures are considered effectively homogenous structures since they feature a unit cell size p much smaller than the guided wavelength (p << g ). z p A p w/wpe B Figure 2-2. Negative permittivity medium. A) The metallic thin wire (TW) array. B) The real part of the permittivity,, as a function of the frequency ratio / pe. (Adaptation of Figure 2 from J. B. Pendry, A. J. Holden, D. J. Robbins and W. J. Stewart, Magnetism from conductors and enhanced nonlinear phenomena, IEEE Trans. Microw. Theory Tech., vol. 47, no. 11, pp , Copyright 1999, with permission from IEEE) Negative Permittivity Medium The metal thin wire (TW) periodic array shown in Figure 2-2 is known as a negative- /positive- plasmonic structure [3]. This structure can be designed to have its plasma frequency in the microwave range. Although negative permittivity materials are found in nature, such as gas and metal plasmas, their plasma frequency is far above the microwave range and their 41

42 permittivity is too large to be considered practical. By using Pendry s metal thin wire array, negative permittivity values on the order of -1 can be achieved at microwave frequencies. When the wires are excited with an electric field E parallel to their axis, a current is induced along the wires and equivalent electrical dipoles moments are generated. The permittivity of the excited metal wires is a function of the frequency given by 2 2 pe pe r ( ) j 2 pe j 2 2, (2-23) where pe is the electric plasma frequency, and is the damping factor due to metal losses. These parameters are defined in terms of the geometry of the periodic wire array as in 2 2c pe 2, (2-24) p ln( p / a) and 2 0 ( p pe / a), (2-25) where c is the speed of the light, a is the radius of the wires, p is the lattice constant or separation between wires, and is the conductivity of the metal. The negative permittivity is achieved if Re( r )<0, for pe, (2-26) which is reduced to r < 0, for < pe, (2-27) when losses are not considered (=0). Then the array shows negative permittivity below the plasma frequency Negative Permeability Medium In order to achieve negative permittivity over a frequency range, Pendry proposed the positive-/negative- metal split ring resonator (SRR) design [3], illustrated in Figure 2-3. When an incident magnetic field H is perpendicular to the plane of the rings, resonating currents will be induced on both the inner and outer rings, charges will accumulate between the gaps of both 42

43 rings, and equivalent magnetic dipole moments are created. Due to the artificial magnetic dipoles, the SRR has a magnetic response despite not being constructed with any magnetic material. y a (a) p A Figure 2-3. Negative permeability medium. A) Split ring resonator (SRRs) structure with lattice size, p, B) Single SRR configuration. (Adaptation of Figures 12(a) and 13 from J. B. Pendry, A. J. Holden, D. J. Robbins and W. J. Stewart, Magnetism from conductors and enhanced nonlinear phenomena, IEEE Trans. Microw. Theory Tech., vol. 47, no. 11, pp , Copyright 1999, with permission from IEEE). The split ring resonator exhibits a frequency dependent permeability, given by [3] B 2 F r ( ) m F ( 0m ) j ( ) ( ) 2 0m 2 2 F j 2 2 ( ) ( ) 2 0m 2. (2-28) The factor F is given by F 2 ( a / p), (2-29) where a is the inner radius of the smaller ring and p is the lattice constant or the center-to center distance between the periodically arranged split rings resonators. The magnetic resonance frequency, 0m, is given by 3p 0m c, (2-30) 3 ln(2a / ) where is the width of the rings, is the space between the inner and outer rings, and c is the speed of the light. The damping factor due to metal losses,, is given by 43

44 ' 2 pr, (2-31) a 0 with R as the metal resistance per unit length Real Imaginary w/w0m Figure 2-4. Variation of the permeability of the SRR with the frequency. The real part in blue, the imaginary part in red. The magnetic resonance frequency is 0m. From Equation 2-28, and as a general case when losses are not considered (=0), it is observed that negative permeability, Re( r ) < 0, exists within the frequency range defined by 0m 0 m pm, (2-32) 1 F with pm as the magnetic plasma frequency. Due to the resonant behavior of the split ring resonator, the nature of the expression for the permeability is resonant ( = 0m = ), which means that the negative permeability is provided within a narrow band of frequencies. Figure 2-4 illustrates the frequency dependent permeability of the SRR. In the SRR, each individual ring behaves as a series LRC circuit with a resonant frequency 1/ 0 LC, where the inductance of the ring is modeled by L, C represents the capacitance due to the gap and R models the conductor losses [11]. Figure 2-5 shows the 44

45 electrical equivalent circuit for the double ring configuration, where the capacitance C m models the capacitive coupling between the rings and the transformer with ratio n models the magnetic coupling. If the mutual coupling between rings is weak, then the circuit parameters of each ring are very close in value because the dimensions of both rings are similar, then L 1 L 2 L, C 1 C 2 C, which gives a combined resonance frequency close to that of a single SRR with the same dimensions. However, due to the higher current density in the double ring SRR, the magnetic moment is larger [11]. L 1 C 1 R 2 C 2 L 2 R 1 C m A n B Figure 2-5. Split ring resonator. A) Double ring split ring resonator. B) Electrical Equivalent Circuit Left-Handed Material Demonstration In naturally occurring bulk materials, the permittivity,, and permeability,, are used to present a homogeneous view of the electromagnetic properties, such as polarization and magnetization. The macroscopic electric and magnetic properties of a medium can be considered as the average behaviors of the electrons and atoms when interacting with the electric and magnetic fields of electromagnetic waves. Therefore, conventional materials are considered composites, where the atoms and molecules are the individual unit cells with sizes much smaller than the wavelength of the EM waves in the medium. In this sense, artificial composite periodic structures, defined by a unit cell with dimension a, can be considered as an electromagnetic 45

46 homogenous medium if the size of the unit cell is much smaller than the wavelength of the EM waves (a << ). Then, the macroscopic electromagnetic properties of the unit cell are considered as the effective EM properties of the medium, allowing refraction phenomena to dominate over scattering and diffraction when EM waves propagate inside the medium. Following the previous reasoning, in [2] Smith et al. created a composite structure by combining the thin metal wire (TW) and the metal split ring resonator (SRR) structures developed by Pendry et al. [3] with the expectation to demonstrate the first artificial left-handed (LH) material. In order to reduce the coupling interactions between the two structures and to cancel mutual interference, the wires were located in the symmetry center of the rings, which creates currents with opposite signs in the SRRs and the TWs [2]. Then, since there is no mutual interference, it is supposed the composite structure preserves both the negative permittivity from the thin metal wires and the negative permeability from the split ring resonators. The TWs and SRRs are designed with overlapping frequency ranges of negative permittivity and permeability respectively. The experiment is performed by exciting the three structures, the SRRs array, the TWs array, and the composite structure with an electromagnetic wave e -jr in a waveguide environment. As shown in Figure 2-4, a frequency band with negative permeability is obtained for the SRRs structure, which causes a stopband when excited due to the magnetic resonance [2]. When the combined structure is excited, the previously observed stopband disappears due to the insertion of the thin wires with negative permittivity. Instead, a passband is observed in the frequency range of interest, which proves that both the permittivity and permeability of the combined structure are negative, therefore a propagation constant with a real value is observed. A similar experiment was later demonstrated by Osbay et al. [4] in a free space environment, in 46

47 which intercalated planar arrays of SRRs and TWs were used to implement a resonant bulk LH material, as previously illustrated in Figure 1-2 and recalled here in Figure 2-6. A x y z B Figure 2-6. Left-handed materials in free space. (Recalled of Figure 1-2). A) The left-handed structure. B) The transmission and reflection responses. (Adaptation and reprint of Figures 1 and 4 from E. Ozbay, K. Aydin, E. Cubukcu and M. Bayindir, M, Transmission and reflection properties of composite double negative metamaterials in free space, IEEE Trans. Antennas and Propagat., vol.51, no.10, pp , Copyright 2003, with permission from IEEE). The two experiments were the beginning of the demonstration of left-handed materials and led to a large number of both theoretical and experimental work confirming the existence and electromagnetic properties of the left-handed materials. In [50], Shelby et al. implemented a two dimensional bulk TW-SRR structure with the shape of a prism. An electromagnetic wave is launched into the structure and the transmission coefficient of the wave refracted by the structure is measured at different angles. The left-handed behavior of the structure is confirmed by observing that the maximum of the transmission coefficient of the refracted wave is obtained in the negative angle with respect to the interface of the prism ( wrong direction with respect to conventional materials). When the structure is replaced by a conventional material (Teflon) with the same shape, the maximum of the transmission coefficient is obtained in the positive angle 47

48 with respect to the interface of the prism. The measured refractive angle was consistent with the Snell s law when the refractive index, n, is negative. Further theoretical and experimental work was performed by different groups [14, 51-56], which confirmed the left-handed properties of the composite metamaterial designs and proposed practical engineering applications Transmission Line Approach of Metamaterials The first experimental demonstration of the composite left-handed (LH) material constituted by the combination of thin wires (TW) and split ring resonators (SRR), opened a new exciting research area in physics and engineering, namely the metamaterials (MTMs). Although the TW-SRR structure shows the unusual properties described by the theory of left-handed materials, it is not practical for engineering applications due to its resonant behavior, narrow bandwidth and high loss. A resonant structure is not a good propagation medium for a modulated signal because the quality factor of the resonator will affect the propagation of the signals, so these cannot be transmitted without distortion. In contrast, nonresonant media can offer a broad bandwidth and low losses, so the modulated signal can be transmitted efficiently. In order to offer practical implementations of metamaterials for engineering applications, the nonresonant transmission line approach for the design of planar metamaterials was proposed, almost simultaneously, by three different groups [7-10] in 2002, leading to the conception of the generalized planar negative-refractive index transmission lines or the composite right/left-handed (CRLH) transmission lines. In contrast with the resonant approach, planar metamaterial transmission lines exhibit broad bandwidth, low loss and can be easily integrated with other microwave components and systems. The CRLH transmission lines (TL) are a mix of both right handed (RH) and left-handed (LH), which creates unique properties widely used for practical engineering applications. 48

49 The Left-handed Transmission Lines The incremental circuit model of a uniform right handed transmission line consists of a series inductor and a shunt capacitor, as shown in Figure 2-7 [57]. The transmission line theory is derived from this model, which offers the well known telegrapher equations that define the most important transmission characteristics of the transmission line, such as the characteristic impedance, propagation constant, phase and group velocities [57]. L R (H.m), Z c Uniform RH TL Z Y C R (F.m) z = p A z = p B Figure 2-7. Modeling of a uniform right handed transmission line. A) Transmission line section of size p. B) Incremental circuit model. C L (F.m), Z 0 Uniform LH TL Z Y L L (H.m) z = p A z = p B Figure 2-8. Modeling of a uniform left-handed transmission line. A) Transmission line section of size p, B) Incremental circuit model. (Adaptation of Figure 1(b) from Lai, A.; Itoh, T.; Caloz, C.;, "Composite right/left-handed transmission line metamaterials," Microwave Mag., IEEE, vol.5, no.3, pp , Copyright 2004, with permission from IEEE). 49

50 A left-handed (LH) transmission line is considered as the dual of a right handed (RH) transmission line, with its incremental model consisting of a series capacitance per unit length (C L ) and a shunt inductance per unit length (L L ), as shown in Figure 2-8 [6]. By using the telegrapher equations [11], and considering the lossless case, the transmission characteristics of this kind of transmission lines can be defined as in 1 j Z' Y' j, (2-33) L ' C ' L L 1 0, (2-34) L ' ' L L ' Z' LL Z c 0, (2-35) ' Y' C L v p 2 L ' L C ' L 0, (2-36) 1 2 ' ' and vg LLCL 0, (2-37) where is the propagation constant, Z c is the characteristic impedance, v p and v g are the phase and group velocities, respectively. It is found that the propagation constant is negative, which means backward waves can be propagated in the structure. Also, Equations 2-36 and 2-37 show that the phase and group velocities are antiparalell. In conclusion, this structure shows the previously described left-handed behavior, and hence can be used as the constitutive unit cell for engineering periodic structures that behave as a left-handed transmission medium if the effective homogeneous medium condition that the size p of the unit cell would be much smaller than the guided wavelength g (p<< g ) is accomplished. 50

51 The Composite Right/Left-handed Transmission Lines A planar microstrip implementation of a left-handed transmission line is shown in Figure 2-9A (recalled from Figure 1-4) [6]. Figure 2-9B shows the unit cell. The series capacitance and shunt inductance of the unit cell are implemented with the interdigital capacitor and the shorted stub inductor, respectively. Although this implementation can show a left-handed behavior, it also has right handed contribution due to the parasitic series inductance provided by the fingers of the interdigital capacitors and the parasitic trace to ground capacitance of the shorted stub inductor. Then, a practical realization of a left-handed transmission line is in reality a combination of right and left-handed contributions, which constitutes the composite right/lefthanded (CRLH) transmission line, as introduced in [6]. A Figure 2-9. Planar composite right/left-handed transmission line. A) Microstrip implementation of a 5 unit cells CRLH Transmission line. (Adaptation of Figure 8 from A. Lai, T. Itoh and C.Caloz, Composite right/left-handed transmission line metamaterials, IEEE Microwave Mag., vol.5, no.3, pp , Copyright 2004, with permission from IEEE). B) Unit cell (Reprint of Figure 5 from C. Caloz and T. Itoh. Transmission line approach of left-handed (LH) structures and microstrip realization of a low-loss broadband LH filter, IEEE Trans. Antennas Propagat., vol. 52, no. 5, Copyright 2004, with permission from IEEE). The equivalent incremental circuit model of a CRLH transmission line is shown in Figure 2-10A. At lower frequencies (0), the model shows a left-handed behavior, since only the left- B 51

52 handed contributions C L and L L are taken into account because L R and C R behave as short and open circuits respectively, as illustrated in the dispersion diagram in Figure 2-10B. At higher frequencies (), the model shows a right handed behavior because L L and C L tend to be open and short circuit, respectively, as illustrated in the dispersion diagram in Figure 2-10B. It is observed that the CRLH unit cell presents a non-linear dispersion diagram, in contrast to the linear phase presented by conventional RH transmission lines. The non-linear dispersion makes the CRLH approach useful for implementing planar multiband applications [16, 21]. L R Z A C L Y z = p C R L L Unbalanced PLH Line - B Figure CRLH transmission line. A) Incremental unit cell model. B) Dispersion diagram showing the three different kind of lines: Pure RH (green), pure LH (blue) and balanced and unbalanced cases for CRLH line (red). (Adaptation of Figure 4 from Caloz, C., Itoh, T., "Metamaterials for High-Frequency Electronics," Proceedings of the IEEE, vol.93, no.10, pp , Copyright 2005, with permission from IEEE). The transmission characteristics of the CRLH TL are obtained by using a similar analysis to that used for RH transmission lines. The series impedance and shunt admittance are given by w sh LH high pass band CRLH w 0 w se Bandgap PRH Line p Z' ' 1 j L R, (2-38) ' CL and Y' ' 1 j C R. (2-39) ' LL 52

53 Then, the complex propagation constant is given by j Z'Y'. (2-40) The series and shunt resonances are given by 1 se, and L C ' R ' L 1 se. (2-41) L C ' L ' R Then two frequency ranges for the LH and RH behaviors are observed. Due to the different values of the series and shunt resonances, a gap exists between the LH and RH ranges, where the group velocity is zero (v g =0) and the maximum attenuation occurs. It is found that the resonance frequency of the CRLH unit cell, which is the frequency of maximum attenuation, is given by 0 L C 1 L C 4 ' ' ' ' R R L L se sh. (2-42) In addition, the characteristic impedance is given by: Z C Z L 2 / se 1, with 2 ( / ) 1 sh Z L L C ' L ' L, and Z R L C ' R ' R, (2-43) where Z L and Z R are the characteristic impedances of the left and right handed contributions, respectively. The case when the series and shunt resonance are equal ( se = sh ) is called balanced, where there is no gap and the three resonance frequency, 0, se and sh are the same. Therefore, the propagation constant and the characteristic impedance for the balanced case are given by ' ' 1 ( bal) LRCR, (2-44) w L C ' L ' L and Z Z Z C( bal) L R. (2-45) 53

54 It is observed that the characteristic impedance is frequency independent, which means broadband matching is possible. On the other hand, it is observed in the diagram for the balanced case that the propagation constant,, is zero at the transition frequency 0, which means that an infinite wavelength regime with non zero group velocity is achieved at a non zero frequency and the phase of the propagating wave is uniform along the CRLH transmission line. This property is very unique and useful on the design of zeroth order resonators. Finally, the effective constitutive parameters can be obtained as in ( ) L, (2-46) C ' 1 R 2 ' L and ( ) C, (2-47) L ' 1 R 2 ' L while the refraction index is given by n, which shows negative values for frequencies r r below the transition frequency o [12] Periodic Structure Implementation It was demonstrated that the incremental equivalent circuit of the CRLH transmission line has both right handed and left-handed properties. As seen in Figure 2-9, a generalized implementation of CRLH metamaterial transmission lines for microwave applications is realized by periodically arranged series capacitors and shorted shunt inductors. A variety of conventional substrates and planar technologies can be used for practical engineering implementations. As previously discussed, if the size, p, of the unit cell is much smaller than the guided wavelength (p<< g ) then the periodic structure is considered an effective homogeneous medium. Periodic structures can be described by using the propagation constant ( = + j) and the Bloch impedance (Z B ), which are obtained by applying periodic boundary conditions to the unit cell represented by its ABCD matrix [57]. Practical CRLH transmission lines are realized by 54

55 using real LC components (C L, L L,C R,L R ) that implement a determined amount, N, of unit cells with small electrical length ( < /2) or small size ( p < g /4). Then the idealized CRLH transmission line can be considered as a practical transmission line if the following relations are applied: Z Z', p Y Y ', (2-48) p ' L R LR p, C ' R CR p, (2-49) ' L L ' L p, C C p. (2-50) L After applying periodic boundary conditions based on the ABCD matrix, the propagation constant,, and the Bloch impedance, Z B, are obtained [11]: L L L R CR ZY L RCR, (2-51) 2 p p LLCL LL CL Z B Z 1 ZY, or from the ABCD parameters Y 4 B Z B. (2-52) A 2 1 Moreover, since the transmission line is a microwave network, the S parameters can be used for characterizing the unit cell. As shown in the dispersion diagram, a CRLH transmission line has a passband behavior where the Bloch impedance is real and the structure can support traveling waves [11]. The lower and higher cutoff frequencies are given by: L cl R 1 1, (2-53) R and L cr R 1 1, (2-54) R 55

56 with the variables w R and w L introduced for convenience as in 1 R, (2-55) L C R R and 1 R. (2-56) L C L L In addition, since the Bloch impedance is frequency dependent and not constant in the passband, CRLH transmission line based designs do not offer optimum solutions in terms of the passband insertion loss. However, the unique properties they offer make them attractive for a great variety of microwave engineering applications. On the other hand, because of the non linear behavior of the dispersion diagram (Figure 2-10B), the group velocity, v g d d 1, and the phase velocity, v p of a propagating wave along the structure can be parallel (RH range ) or antiparallel (LH range). During the LH range ( cl < < 0 and < 0), the group velocity is positive while the phase velocity is negative, which means that a backward wave is propagated where the energy flows from source to load and the phase advances toward the source [11]. During the RH range ( 0 < < cr and > 0) both the group and phase velocities are positive, which means that a conventional forward wave propagation is obtained where the energy flows from source to load and the phase advances toward the load [11]. A special and interesting case is presented when the frequency of the propagating signal is equal to the transition frequency ( = 0 ). At this point, the propagation constant is zero ( = 0), which means that the phase velocity is infinite (v p =) and the group velocity is positive. Therefore, an infinite wavelength propagation regime is presented where the energy flows toward the load but the phase is constant along the CRLH transmission line (as a DC behavior). 56

57 2.6. Review of CRLH Transmission Line Applications After the introduction of the transmission line approach for implementing planar metamaterials, many microwave and millimeter wave guiding and radiating applications have been demonstrated. The unusual properties of the proposed CRLH transmission lines such as non linear dispersion, bandpass behavior, infinite-wavelength regime, backward wave propagation and leaky-wave radiation, are exploited for implementing creative new applications. In this session, some examples of the CRLH transmission lines are discussed and their operating principle is explained Metamaterial Couplers and Filters The conventional broadband microstrip directional couplers have weak coupling levels of 10 db or less [57]. Tight coupling levels around -3 db are normally achieved over a narrow bandwidth (<10%) by using non-coupled line couplers, such as the branch-line and ring couplers [57]. However, if tight coupling over a broad bandwidth is desired, the use of complicated coupler configurations, such as the Lange coupler, is necessary. As an alternative, PCB based symmetric (two CRLH TLs) and asymmetric (only one line is CRLH TL) broadband (50%) arbitrary coupling level (even 0 db) directional couplers [12,13] were demonstrated by using CRLH transmission lines. Arbitrary tight coupling achieved in the CRLH TL couplers is the result of imaginary even and odd mode impedances within the coupling range, which offers a broad matching frequency range. On the other hand, the broadband passband behavior of CRLH transmission lines make them useful for the implementation of low loss broadband filter applications [14, 15] Multiband Components The unique non linear dispersion characteristic of the CRLH transmission lines makes them useful for the design of dual band components, since the specific values of phase shifts can 57

58 be obtained at arbitrary frequencies [16]. Moreover, the combination of conventional CRLH transmission lines with the dual-crlh transmission line concept [17] is used for the design of triple-band and quad-band miniaturized applications [18, 19]. Branch line and rat race ring couplers [16], quadrature mixers [20], Bagley polygon power divider [21] (part of our study) and dual band antennas [22] were demonstrated by using dispersion engineering of the CRLH and dual-crlh transmission lines. Figure 2-11shows one of the dual band applications of the CRLH transmission lines. A B Figure CRLH dual band components. A) Dual band CRLH branch line coupler. B) Frequency response of the dual band coupler. (Reprint of Figures 10 and 11 from I.H. Lin, M. De Vincentis, C. Caloz and T. Itoh. Arbitrary dual-band componentsusing composite right/left-handed transmission lines, IEEE Trans. Microw. Theory Tech., vol. 52, no. 4, pp , Copyright 2004, with permission from IEEE) Metamaterial Resonators The CRLH transmission lines can also be used as short or open ended resonators if their length is a multiple of half of the guided wavelength [23]. In addition to the conventional positive order resonances (in the RH band of the resonator), CRLH resonators exhibit negative order resonances in the LH band, and an unusual zero-order resonance at the transition frequency 58

59 o. Moreover, since the propagation constant β= 0 at o, CRLH TL zeroth order resonators are independent of the physical length, which allows them to be arbitrarily small. This kind of resonators has been used for the implementation of small zeroth order resonant antennas and tunable applications [24, 25] Compact Multilayer Components Although the CRLH approach is a good way to implement planar metamaterials, some of the demonstrated applications are implemented by using surface mount technology (SMT) discrete chip components (capacitors and inductors), which are available only in discrete values and have a limited frequency operation range due to the natural self resonance frequency associated with the chips. On the other hand, chip components cannot be easily integrated in flexible, multilayer or 3D structures that may be used for miniaturization purposes or reconfigurable radiated wave applications. The above considerations, which represent the need for additional size reduction, operation at higher frequencies and 3D integration, extended the transmission line approach to the implementation of super compact multilayer vertical CRLH transmission lines on conventional printed circuit board processes and ceramic LTCC processes combined with MIM capacitors and meander lines [26-27]. This multilayer approach offers an alternative to the conventional implementations with printed planar passive components (interdigital capacitors, stub inductors) and allows 3D structure implementation. The LTCC technology is an example of 3D approaches for miniaturized CRLH devices for the microwave range. Since the CRLH concept does not depend on the fabrication technology used, the multilayer vertical CRLH transmission line has all the unusual properties previously described and thus, can be used for the implementation of new metamaterial applications. 59

60 CRLH Substrate Integrated Waveguide Components Recently, the substrate integrated waveguide (SIW) [41] and the half mode substrate integrated waveguide (HMSIW) [42] technologies, have been integrated with interdigital capacitors in order to achieve SIW and HMSIW CRLH transmission lines that can be used for implementing microwave applications [28], with all the well known advantages offered by the SIW and HMSIW, such as high Q factor and low radiation losses. Broadband and dual band couplers [28,29], leaky wave antennas [30] and negative order resonator antennas and filters [31] ( Figure 2-12 ) have been demonstrated by using the concept, which places the recently introduced SIW and HMSIW technologies as good architectures for the implementing CRLH devices for microwave and millimeter wave applications. Figure CRLH SIW slot antennas. A) One and two stages CRLH SIW antennas. B) Frequency response of the two stages open ended antenna. (Reprint of Figures 10 and 11 from Y. Dong, T. Itoh, Miniaturized Substrate Integrated Waveguide Slot Antennas Based on Negative Order Resonance, IEEE Trans. Antennas and Propagat., vol.58, no.12, pp , Copyright 2010, with permission from IEEE) Micromachined CRLH Applications A As the frequency increases, the demand for miniaturized devices is stronger. This seems to be the motivation for using microfabrication techniques, CMOS and MMIC processes in the implementation of left-handed transmission lines. Qin et al. [32] successfully demonstrated left- 60 B

61 handed transmission lines operating at the V band by using the negative epoxy photoresist SU8 as a dielectric. Also, CRLH transmission lines have been implemented in MMIC GaAs and Si processes [33, 34]. Previous work has demonstrated that the use of thin and thick polymer dielectric materials wit low loss such as SU8, BCB and Polyimide, allows the implementation of micromachined transmission lines and passive devices on a CMOS grade low resistivity silicon substrate. The dielectric on top of the silicon serves as a passivation layer that decreases the losses due to the silicon and avoids the use of the expensive high resistivity silicon wafers, offering at the same time a way to implement CMOS compatible devices in the microwave and millimeter wave range [35-40]. However, due to the availability of low cost printed circuit board (PCB) and LTCC processes, and the technical challenges in multilayer fabrication, organic based microfabrication processes have not been broadly used for implementing multilayer metamaterial devices in the microwave range and millimeter wave range. 61

62 CHAPTER 3 MICROMACHINED TRANSMISSION LINES During the past decade, advanced 3D integration microelectromechanical systems (MEMS) fabrication technologies have been used to build RF components and devices with similar or superior performance to that of conventional counterparts, while simultaneously achieving more compact and less expensive products. MEMS fabrication techniques for 3D structures, such as surface micromachining, bulk micromachining, LIGA processes, wafer bonding, polymer based processes and lamination, have revolutionized the development of RF and wireless communication systems. The new applications and fabrication technologies have created what is commonly known today as RF-MEMS, which includes not only RF devices with movable microelectromechanical parts, but also passive guiding and radiating wave RF components fabricated by using MEMS technologies in order to reduce size, enhance performance parameters and achieve 3D integrability. These microfabrication technologies have been beneficial for the development of new devices for microwave and millimeter wave applications, such as RF micro-switches [58], miniaturized micromachined filters [59] and antennas [60,61], miniaturized embedded high Q passive components [44] and micromachined transmission lines [35-40]. It is well known that at high frequencies the signal to be transmitted is considered to be an electromagnetic wave, whose wavelength is normally comparable with the size of the passive and active components in the system. That is the reason to use different kinds of distributed wave guiding structures, such as transmission lines, which can effectively propagate the signal with low loss and distortion. The characteristic impedance of the transmission line, the propagation constant and the wave propagation mode are important parameters taken into account during the design process. According to the wave polarization properties, different modes of propagation 62

63 are possible in a wave guiding structure, such as Transversal-Electro-Magnetic (TEM), Transversal Electric (TE) and Transversal Magnatic (TM). Most practical applications work with a single mode of propagation, which is defined by the geometry of the structure and the wave polarization properties. Figure 3-1 shows some of the planar transmission lines that are widely used for microwave and millimeter wave applications. W W S Ground S Ground S H Substrate Ground plane Ground A B C W W g S W g W S S S Vias D E F Ground Figure 3-1. Cross section of some transmission lines used in microwave and millimeter wave circuits. A) Microstrip. B) Coplanar Waveguide (CPW). C) Grounded Coplanar Waveguide (GCPW). D) Finite Ground CPW (FGC). E) Coplanar Strip Line (CPS). B) Substrate Integrated Waveguide (SIW). One of the most popular planar guiding structures is the microstrip line in Figure 3-1A, which has a metallic signal line on top of a ground plane. The fundamental propagation mode in the microstrip line is known as quasi-tem, because of its similarity to the pure TEM mode. However, because of the needs for vias to the ground, some dispersion in the parameters of the line (frequency dependence), high radiation loss, and cross talk between adjacent lines, the microstrip line is often rejected to be used in some microwave and millimeter wave applications. As alternatives, the coplanar waveguide (CPW) in Figure 3-1B and its variants such as the 63

64 Grounded Coplanar Waveguide (GCPW) in Figure 3-1C and the Finite Ground Coplanar Waveguide (FGC) in Figure 3-1D, are used in a variety of applications. The uniplanar implementation of the CPW allows only CPW modes to be transmitted when all the ground terminals are kept at the same potential. However, in the majority of the applications air bridges are necessary to connect the ground terminals and avoid slotline modes. One of the most interesting variants of the CPW is the Finite Coplanar Waveguide (FGC) [62], which has finite size narrow ground planes in order to avoid parallel plate modes and spurious resonances normally presented in the CPW or GCPW configurations, while at the same time reducing mutual coupling between adjacent lines and radiation losses. Another popular waveguiding structure is the Coplanar Strips Line (CPS), shown in Figure 3-1E, which has a balanced structure that is useful for RFIC applications with reduced cross talk, high common rejection ratio and high quality signal integrity [40]. Recently, the Substrate Integrated Waveguide (SIW) [41], shown in Figure 3-1F, has been widely used for microwave and millimeter wave applications. The SIW and their different variants, such as the Half Mode SIW (HMSIW), offer a planar in-substrate implementation of a rectangular waveguide-like transmission line. A row of metalized vias at each side connecting top and bottom conductors behave as a metallic wall, and so a waveguide mode is propagated. This in-substrate implementation offers all the advantages of a conventional cavity waveguide, such as high Q factor, low radiation losses and low cross talk between adjacent lines, but in a planar fashion that is compatible with conventional microstrip and CPW implementations. However, the necessity of an array of via holes makes its implementation more complex than that of CPW or Microstrip. 64

65 It is observed that the availability of MEMS micromachining techniques has made it possible to fabricate miniaturized transmission lines for microwave, millimeter wave and terahertz applications. It is the purpose of this chapter to present a literature review of the most important and interesting works on micromachined conventional and left-handed transmission lines Bulk Micromachined Transmission Lines In selected microwave and millimeter wave applications, different micromachining techniques are used for fabricating transmission lines with reduced dielectric and radiation losses, in part due to the use of low resistivity substrates. This is the reason why micromachining techniques have been used more on both conventional low resistivity and high resistivity silicon than on GaAs substrates. By removing part of the lossy dielectric substrate, where the electromagnetic field is confined, the line is basically implemented on air dielectric and thus the dielectric losses and the dispersive characteristics of the parameters of the line are significantly reduced. In fact, having air rather than the lossy substrate as the dielectric, the attenuation of the CPW can be reduced considerably. On the other hand, the use of microfabricated shields can reduce the radiation losses. Bulk micromachining is the selected technique to remove the substrate between metalized signal lines or ground planes. One of the pioneer works is presented in [63] by Herrick et al., in which trenched finite ground coplanar waveguide (FGC) lines are implemented with a removed substrate between terminals by using ethylenediamene pyrocatechol (EDP) wet etching. Figure 3-2A shows the final structure. The measured attenuation of the micromachined FGC line is db/mm at 60 GHz. The fabricated line has lower dielectric, ohmic and radiation losses in addition to lower parasitic and coupling effects between adjacent lines. 65

66 Open areas SiO 2 Ground Signal Ground Ground Signal Ground Ground Signal Ground Si Si Dielectric membrane Si Si Air Si A B C SiO 2 Ground Signal Ground Ground Si Signal Air cavity Ground Si Air Membrane Si D E Figure 3-2. Bulk micromachined transmission lines. A) Trenched finite ground coplanar (FGC) micromachined transmission line. B) Dielectric membrane supported coplanar waveguide transmission line (CPW) (Adaptation of Figures 1(c) and 2(c) from K. J. Herrick, T. A. Schwarz and L. P. B. Katehi, Si-micromachined coplanar waveguides for use in high-frequency circuits, IEEE Trans. Microw. Theory Tech., vol. 46, pp , Copyright 1998, with permission from IEEE). C) Partially membrane supported CPW line with SiO 2 membrane (Adaptation of Figure 2 from V. Milanovic, M. Gaitan, E. D. Bowen and M. E. Zaghloul, Micromachined microwave transmission lines in CMOS technology, IEEE Trans. Microw. Theory Tech, vol. 45, pp , Copyright 1997, with permission from IEEE). D) High aspect ratio CPW transmission lines (hicoplanar) (Adaptation of Figure 5 from S. T. Todd, X. T. Huang, J. E. Bowers and N. C. MacDonald, Fabrication, modeling, and characterization of high aspect ratio coplanar waveguide," IEEE Trans. Microw. Theory Tech, vol. 58, no. 12, part 1, pp , Copyright 2010, with permission from IEEE). E) Membrane supported cavity backed grounded CPW transmission line (GCPW) (Adaptation of Figure 1(b) from Y. Yoshida, T. Nishino, J. Jiao, S. Lee, Y. Suehiro, K. Miyaguchi, T. Fukami, M. Kimata, O. Ishida, A novel grounded coplanar waveguide with cavity structure," in Proc. Micro Electro Mechanical Systems Conf., Kyoto, vol., no., pp , Copyright 2003, with permission from IEEE). A different approach reduces the dielectric losses by partially or completely removing the substrate underneath the signal trace, which creates a suspended thin dielectric membrane backed by an air cavity. In [64] anisotropic wet etching is used to remove part of the silicon substrate in order to create a thin membrane. Then, CPW transmission lines are fabricated on a thin dielectric membrane made of SiO 2 /Si 3 N 4 /SiO 2, which is grown on the surface of the substrate. The lines 66

67 are characterized up to 40 GHz and their performance is compared with that of CPW lines fabricated on bulk high resistivity silicon wafers, featuring lower dielectric loss. Similar work has used different dielectric materials for implementing the thin membrane, such as SiO 2 [65] and polyimide [66], with similar results in term of attenuation, ranging from 0.06dB/mm at 6 GHz to 0.4 db/mm at 40 GHz. Some membrane based structures are shown in Figure 3-2B and Figure 3-2C. High aspect ratio coplanar waveguide transmission lines (hicoplanar) have been implemented by using a combination of micromachining techniques. In [67] Todd et al. have used deep reactive ion etching (DRIE), thermal oxidation, anisotropic oxide etching, electroplating and chemical mechanical planarization for the fabrication of a thick membrane (38 m) that implements a high performance cavity backed hicoplanar waveguide transmission line with low conductor loss. An attenuation constant of 2.4 db/cm at 30 GHz has been obtained. The fabricated structure is shown in Figure 3-2D. Membrane supported grounded coplanar waveguide (GCPW) transmission lines have been also implemented by using bulk micromachining. In [68] Yoshida et al. present a GCPW transmission line fabricated by using a combination of alkaline etching, thick photoresist patterning and chemical mechanical planarization (CMP). Silicon nitride is used as the dielectric material for the membrane. The ground plane is metalized on the cavity after the alkaline etching is performed. A maximum attenuation of 0.08 db/mm is obtained at 20 GHz. The fabricated structure is shown in Figure 3-2E. The work mentioned above has offered new ways to implement miniaturized micromachined transmission lines for MMIC applications and opened a new area for the development of wireless communication devices and RF MEMS. However, bulk micromachining is not the only technique that has been used for these new implementations. 67

68 3.2. Surface Micromachined Transmission Lines A different group of transmission lines for RFIC applications has been fabricated by using surface micromachining techniques, which instead of removing part of the substrate incorporate the use of different non-photopatternable and photopatternable organic materials on top of the substrate as dielectric membranes or dielectric interface layers in/on which the transmission lines are implemented. Further, this approach allows the fabrication of multilayer structures with vertical interconnections for microwave and millimeter wave applications. In [35] Ponchack et al. characterized thin film microstrip lines (TFMS) on polyimide. A thin polyimide layer is deposited onto a previously metal coated silicon wafer used as a carrier substrate. Thicknesses from 2.45 m to 7.45 m are used. Standard metal evaporation, polymer spinning, lithography and via hole formation by using deep reactive ion etching (DRIE) are used for the fabrication of the transmission lines. Characterization is performed from 1 to 110 GHz, featuring a maximum attenuation constant around 5 db/cm at 60 GHz for a TFMS line implemented on a 7.45 m thick polyimide. The cross section of the fabricated structure is shown in Figure 3-3A. Following his own work, in [36] Ponchack et al. implemented low loss CPW transmission lines on low resistivity silicon substrates with a micromachined polyimide interface layer. A 20 m thick polyimide is deposited as an interface layer on top of a low resistivity silicon substrate. The thickness of the polyimide is optimized in order to minimize the interaction of the electromagnetic field with the lossy silicon substrate. Further steps of lithography, metallization and DRIE are followed to create a trenched CPW line in which the polyimide between conductors has been removed in order to reduce the attenuation constant. The cross section of the structure is shown in Figure 3-3B. The etching of the polyimide lowers the 68

69 effective dielectric constant, line capacity and current density, thus, decreasing conductor and dielectric losses. A maximum attenuation of 2.75 db/cm is obtained at 40 GHz. W H Polyimide t H W S W t Si Ground Si Dielectric Layer A B H W S SU8 W H S W SU8 S Carrier Substrate Si C D Figure 3-3. Surface micromachined transmission lines. A) Thin film microstrip line (TFMS) on polyimide (Adaptation of Figure 1from G. E. Ponchak and A. N. Downey, Characterization of thin film microstrip lines on polyimide, IEEE Trans. Comp., Packag., Manufact. Technol. B, vol. 21, pp , Copyright 1998, with permission from IEEE). B) Trenched CPW line on a micromachined dielectric layer (Adaptation of Figure 1from G. E. Ponchak, A. Margomenos and L. P. B. Katehi, "Low-loss CPW on low-resistivity Si substrates with a micromachined polyimide interface layer for RFIC interconnects," IEEE Trans. Microw. Theory Tech., vol. 49, pp , Copyright 2001, with permission from IEEE). C) Grounded CPW line on SU8 (Adaptation of Figure 1from F. D. Mbairi and H. Hesselbom, High frecuency design and characterization of SU-8 based conductor backed coplanar waveguide transmission lines, in IEEE Int. Symposium on Advanced Packaging Materials, Copyright 2005). D) Coplanar strips line on SU8 (Adaptation of Figure 1from M. S. Arif and D. Peroulis, Loss optimization of coplanar strips for CMOS RFICs, in Proc. Asia-Pacific Microw. Conf., Copyright 2009, with permission from IEEE). Similar work is presented in [37], in which a thick SU8 layer is used as a dielectric interface on top of a low resistivity silicon wafer. The use of SU8, which is a negative tone photoresist widely used in MEMS fabrication, offers various thicknesses, high aspect ratio imaging, low curing temperature, high transparency and easy fabrication of multiple layers, in contrast with the use of polyimide. The SU8 is patterned by using standard UV lithography, 69

70 followed by metallization and lift-off techniques to define the trenched CPW lines. Maximum attenuation of 0.6 db/mm at 60 GHz is achieved. Although the attenuation is higher than in previous work with polyimide, the use of SU8 offers advantages in the fabrication process. In a similar way, Benzocyclobutene (BCB) has also been used as an interface layer for implementing low loss CPW transmission lines on low resistivity silicon, although with increased fabrication costs [38]. Grounded coplanar waveguide (GCPW) and coplanar strip (CPS) transmission lines have been also implemented by using SU8 as a dielectric. In [39] GCPW transmission lines are fabricated and characterized up to 50 GHz. A 35 m thick SU8 layer deposited on top of a metalized substrate, in this case glass, is used as the dielectric material. Maximum attenuation of around 3.5 db/cm is obtained at 50 GHz. The work also provided the loss tangent and dielectric constant of the SU8. The cross section of the structure is shown in Figure 3-3C. On the other hand, in [40] loss optimization of CPS lines on low resistivity silicon is presented, as shown in Figure 3-3D. SU8 thicknesses of 10 m and 15 m are selected for the implementation. Conventional UV lithography, metallization and lift-off techniques are used for fabrication. A maximum attenuation constant of 0.9 db/mm at 40 GHz is obtained, which is comparable with that obtained in previous work reporting CPW transmission lines. Further, the use of a less lossy substrate such as polyimide or BCB can decrease the loss, at the expense of higher fabrication complexity and costs. Finally, a complicated dielectric post elevated structure with air gap as dielectric, which basically eliminates dielectric losses, has been demonstrated for the implementation of microstrip lines for millimeter wave applications by using surface micromachining techniques [69]. By elevating the microstrip line, the loss can be reduced to 70

71 1.1 db/cm at 50 GHz, in comparison with the 10dB/cm achieved by a conventional microstrip line implemented for comparison. It is clear that micromachining techniques have been used for the implementation of transmission lines and passive components at microwave and millimeter wave frequencies. These techniques offer great flexibility during the fabrication and allow the implementation of vertical 3D interconnected multilayer devices compatible with the CMOS MEMS technology. On the other hand, it is worth to mention that multilayer lamination technologies, such as the low temperature cofire ceramic (LTCC) [70-72] and the liquid crystal polymer (LCP) [73-75], have been used, in combination with MEMS fabrication techniques, for the implementation of compact 3D integrable passive microwave and millimeter wave devices. In the same way, some work has explored the microfabrication of metamaterial transmission lines. In the next session, a review of left-handed transmission lines fabricated by using MEMS or lamination processes is provided Multilayer and Microfabricated CRLH Transmission Lines Applications Multilayer metamaterial transmission lines and passive devices are implemented in a variety of technologies. The need for size reduction, high frequency operation and the 3D vertical integration with multilayer devices extended the transmission line approach to the implementation of compact multilayer CRLH transmission lines on conventional printed circuit board processes and ceramic LTCC processes. In [26] Horii et al. introduced an unbalanced super-compact multi-layered left-handed transmission line with a narrow left-handed passband. The implementation is done based on the conventional printed circuit board technology with multiple layers containing parallel plate capacitors and grounded meander line inductors for providing the left-handed contributions. Since the direction of propagation is on the vertical direction, perpendicular to the ground plane, large electrical length can be achieved over a short 71

72 footprint of the transmission line. A diplexer for 1 GHz / 2 GHz operation is implemented by using the compact multilayer transmission line. Further, in [76] the same author proposed the balanced version of the transmission line with a wide bandwidth from 1.8 GHz to 8.7 GHz, featuring at the same time less inner coupling between capacitors and lack of in-band transmission zeros. Following the same trend in the implementation of miniaturized CRLH transmission lines, in [77] Nguyen et al. presented a 6 GHz 3dB coupled line coupler implemented with CRLH transmission lines that incorporate metal-insulator-metal (MIM) capacitors for providing the lefthanded contributions. Conventional printed circuit board was used for the implementation. A 30% bandwidth around the design frequency is obtained. The coupler features compact size, tight coupling and spurious free response. The schematic of the proposed structured is shown in Figure 3-4A. A B Figure 3-4. Unit cells of multilayer CRLH transmission lines. A) MIM CRLH unit cell used in [77] (Reprint of Figure 1(a) from H.V. Nguyen and C. Caloz, Simple-Design and Compact MIM CRLH Microstrip 3-dB Coupled-Line Coupler, in IEEE MTT-S Int. Microw. Symp. Dig., Copyright 2006, with permission from IEEE). B) A similar MIM CRLH unit cell implemented on LTCC [27] (Reprint of Figure 2 from A. Rennings, T. Liebig, C. Caloz and P. Waldow, CRLH series mode zeroth order resonant antenna (ZORA) implemented in LTCC technology, in Proc. Asia-Pacific Microw. Conf., Bangkok, Thailand, Copyright 2007, with permission from IEEE). 72

73 Although the previous mentioned works used the conventional printed circuit board technology, the architecture is useful for implementation on advanced lamination processes such as low temperature cofire ceramic (LTCC) or liquid crystal polymer (LCP). In [70] Piatnitsa et al. demonstrated fully integrated multilayer LTCC 3 db and 10 db directional, as well as dual band rat race couplers, using a combination of right-handed and left-handed transmission lines. The structure features an area four times smaller than conventional couplers implemented for comparison. Further, in [27] a CRLH zeroth order resonant antenna (ZORA) is implemented in the LTCC technology. The schematic of the unit cell is presented in Figure 3-4B. Metalinsulator-metal (MIM) capacitors and grounded stub inductors are used for the left-handed contributions. The antenna exhibits an excellent efficiency of 71% and a high gain of 10 db around 11.5 GHz. The high gain is provided by the large electrical size of the antenna (2 0 ), which is a result of the series implementation. Leaky wave antennas and dual band couplers designs by using multilayer CRLH transmission lines on the LTCC technology have been also implemented [71-72], demonstrating high performance and a reduced size. Although flexible substrates, such as liquid crystal polymer (LCP), have been explored for the implementation of transmission lines and passive devices at microwave and millimeter wave frequencies, no significant work has been presented so far on metamaterial devices in the LCP technology [73-75]. The implementation of devices on liquid crystal polymer have used MEMS fabrication techniques, such as conventional UV lithography, metallization, lamination and via formation trough laser or mechanical machining, and can be a good alternative for implementing metamaterial devices on low cost organic substrates. On the other hand, the demand for 3D CMOS-MEMS integrable miniaturized devices, MMIC applications and higher operating frequencies seems to be the motivation for using 73

74 microfabrication techniques in the implementation of left-handed transmission lines and applications. Qin et al. [32] successfully implemented left-handed transmission lines operating at the V band by using the negative epoxy photoresist SU8 as a dielectric. The mushroom structure [11] is selected as the unit cell for 1-D and 2-D CRLH transmission lines. Wide left-handed passband between 42 GHz and 73 GHz is achieved for 5, 7 and 9 unit cells CRLH transmission lines. The structure of the implemented trasnsmission lines and the measurement results are shown in Figure 3-5D. Despite of the high insertion loss, the structure demonstrated to be useful for implementing miniaturized devices for millimeter wave applications. A similar approach is used by Tong et al. [33] for implementing left-handed metamaterial coplanar waveguide components (transmission line, short and open stub) on the GaAs MMIC technology with application to a filter and a power divider. A miniaturized left-handed bandpass filter working from 1.02 GHz to 1.42 GHz and a power divider working from 2.8 GHz to 3.72 GHz were demonstrated. In similar work, the same authors presented a 3D multilayered left-handed bandpass filter on high resistivity silicon, featuring a total area of 2.6 mm 2 [34]. Wide band operation from 2 GHz to 7.5 GHz is achieved. However, the use of GaAs and high resistivity silicon substrates, combined with oxygen plasma to etch thin polyimide layers during the fabrication process, increases fabrication costs and complexity. In addition, full planar implementation of CRLH transmission lines on high resistivity silicon has also been presented. In [78] interdigital capacitors and grounded stub inductors are used for the implementation of a CRLH based directional coupler working from 10 GHz to 14 GHz. Standard UV lithography and metallization are used for the fabrication of the coupler. However, it should be noticed that interdigital capacitors present high parasitic inductance, which causes low self resonance frequency, resulting in limited operating bands. 74

75 C A D B Figure 3-5. Implemented CRLH transmission line on SU8. A) 5 cell transmission line structure. B) Side view of the CRLH TL. C) Measured insertion loss. D) Measured phase highlighting the left-handed region. (Reprint of Figures 1 and 6 from C. Qin, A. B. Kozyrev, A. Karbassi and D. W. van derweide, Microfabricated v-band left-handed transmission lines, Metamaterials, vol. 2, Copyright 2007, with permission from Elsevier). Recent work has integrated MEMS devices with microfabricated CRLH transmission lines in order to create switchable devices. In [79] MEMS switches are integrated with a CPW CRLH transmission line implemented on the Silicon on Glass (SOG) technology. Left-handed behavior from 7.5 GHz to 10.4 GHz with an insertion loss of 3.4 db 0.9 db is achieved. Moreover, switchable negative to positive phase response or vice versa is obtained by activating the MEMS switches that connect or disconnect MIM capacitors in the structure. Using a similar approach, in [80] Ouagague et al. implement reconfigurable CRLH cells with RF MEMS switches on the high resistivity (HR) silicon technology. MEMS switches and shunt stub capacitors are used to modify the frequency range of the left-handed and right-handed behaviors. Measurement results from 2 GHz to 30 GHz are provided, showing 15 % of frequency tunability by adding the shunt stub capacitor and up to 150 % by changing the MEMS capacitor state. 75

76 Recently, a new work demonstrating a different configuration of a CRLH unit cell with an all-pass behavior up to 35 GHz has used high resistivity silicon and air bridges [81]. The previously mentioned research works have demonstrated the feasibility of the integration of CRLH transmission lines and RF MEMS devices and fabrication technologies to achieve flexible and reconfigurable applications. As a summary, it is observed that a great variety of fabrication technologies are used for implementing compact multilayer metamaterial applications in the microwave and millimeter wave range. However, most of the work uses a high resistivity silicon or GaAs substrate, which increases fabrication costs and is not compatible with the conventional CMOS technology. As previously discussed, some work has demonstrated the possibility of implementing miniaturized conventional transmission lines on low resistivity silicon by using dielectric interface layers. However, due to the availability of printed circuit board (PCB) and LTCC processes, and the technical challenges in multilayer fabrication, organic based microfabrication processes based on low resistivity silicon have not been broadly used for implementing multilayer metamaterial devices for microwave range. It is the aim of our study to develop single and multiband multilayer micromachined metamaterial applications for microwave range up to 40GHz. In the next session, the proposed unit cell for the implementation of CRLH metamaterial applications in our study is introduced Proposed Multilayer Unit Cell for the Implementation of CRLH Metamaterial Applications The main focus of our research is the design and implementation of highly compact metamaterial engineered components and devices for microwave and millimeter wave applications. At first, the previously discussed CRLH architecture is combined with a multilayer surface micromachined fabrication process based on the negative tone photopatternable epoxy 76

77 SU8 and the negative tone photopatternable resin Benzoclyclobutene (BCB) as dielectric interface layers implemented on low cost substrates such as low resistivity silicon, glass, and conventional printed circuit board substrates. Multilayer grounded coplanar waveguide (GCPW) and finite ground coplanar waveguide (FGC) composite right/left-handed balanced transmission lines are demonstrated for the design of multiband microwave applications up to 40 GHz. Moreover, with minor modifications, the unit cell can be used for implementing a different version of the CRLH architectures, such as the dual-crlh [17] or the new CRLH unit cell proposed in our study. The schematic and simplified equivalent circuit of the proposed structure for a multilayer micromachined composite right/left-handed unit cell is presented in Figure 3-6. Note the structure is patterned in a dielectric embedded fashion, which can be SU8 or BCB, while the dielectric and upper ground plane is not drawn for structural clarity in Figure 3-6A. The via connects the inductor to the ground plane. The structure allows coplanar waveguide (CPW), grounded coplanar waveguide (GCPW), finite ground coplanar waveguide (FGC) or microstrip (MS) implementations. In this work, FGC and GCPW implementations are explored. L line W cap MIM capacitor 2C L 2C L W line h To carrier substrate g i l i L cap d w i L R /2 L R /2 L L C R Via-Hole A SU8 or BCB embedded inductor B Figure 3-6. Multilayer CRLH unit cell on SU8 or BCB. A) Schematic. B) Simplified equivalent circuit. 77

78 The left-handed inductor L L is implemented using a dielectric embedded meander line inductor. The metal patches on the lower and upper layers create MIM capacitors, which make the left-handed capacitance contribution C L. The right-handed contribution is provided by the feeding transmission lines at both sides, modeled by the right-handed inductor and capacitor L R and C R, respectively. The left-handed inductance value can be controlled by adjusting the width w i of the inductor trace, the number of turns and the total length L i. Although the inductance of an embedded single layer meander line inductor is smaller than that of the conventional solenoid or spiral inductor, it facilitates the fabrication process. Neglecting fringing effects, the lefthanded capacitance value is controlled by the area of the metallic patches and the separation distance d between the upper and lower layers. The total height of the structure is represented by h. The inter-capacitor gap g i is selected to be large enough to be negligible in the left-handed capacitor design. From the previous description it is observed that the combination of the right-handed and left-handed components L R, C R, L L, and C L creates a composite right/left-handed artificial transmission line, which presents a frequency range with backward wave propagation and positive phase. The design, simulation, fabrication and testing of this unit cell for single and dual band applications at microwave and millimeter wave frequencies vs presented in our study Proposed Multilayer Embedded Substrate Integrated Waveguide Filter Architecture The study of devices that use metamaterial concepts is not limited to the composite right/left-handed approach for planar metamaterial transmission lines. A different metamaterial concepts is used for the implementation of 3D integrable, compact SU8 embedded evanescent mode resonators and bandpass filters by using the same multilayer surface micromachined fabrication process in combination with the in-substrate waveguide. The complementary split 78

79 ring resonator (CSRR) [82], which is considered a metamaterial particle, is loaded in a transmission line implemented with the half mode substrate integrated waveguide (HMSIW), which produces a resonance frequency below the characteristic waveguide cutoff frequency due to evanescent wave amplification [45-46]. The evanescent wave amplification concept offers a great size reduction since the resonator can be smaller than the quarter wavelength at the resonance frequency. Photopatternable SU8 epoxy is selected as the dielectric for implementing the embedded passive devices on low cost carrier substrates. Conventional printed circuit board (PCB) or glass are selected as the supporting substrates for the micromachined filters, which keeps the compatibility with conventional microwave PCB implementations and CMOS integrated circuits. Although the complete implementation of these filters requires further study, the cross section of the multilayer embedded filters is shown in Figure 3-7 [83]. SU8 or BCB Metal Metal Figure 3-7. Cross section of the proposed dielectric embedded resonators and filters. In Figure 3-7 it is observed that multilayer fabrication is possible by adding multiple dielectric layers and vertical interconnections. BCB can also be used as a dielectric for the filters. A long metalized via wall is used for the substrate integrated waveguide architecture. Since the 79

80 process uses surface micromachining, the supporting substrate can be either silicon, glass, or organic materials such as ones for printed circuit board. The resonator is implemented on top of the first dielectric layer. The ground plane is implemented on the second dielectric layer, and the electroplated vertical interconnects are used for the signal line and the vias to ground, as proposed in [44]. The upper layer of dielectric allows the implementation of CPW or microstrip lines for feeding as well as different components such as wideband antennas. More detail on the implementation of the multilayer filters is further provided in Chapter Proposed Multilayer Architecture for Micromachined Wideband Bandpass Filters In addition to the previously described embedded filter architecture, we also study the design and implementation of surface micromachined wideband cavity filters. New in-soubstrate waveguide cavity resonators are proposed by using reduced mode versions of the original substrate integrated waveguide cavity. Conventional printed circuit board, liquid crystal polymer (LCP) and BCB are selected as dielectric materials for the proposed filters. At first, a wideband bandpass filter is demonstrated on a printed circuit board and fabricated by using a conventional CNC (computerized numeric control) milling machine. Then, LCP is used for the implementation of two pole and three pole filters for 25 GHz. The fabrication process of the filters on LCP is proposed and uses a combination of mechanical drilling of the via holes and surface micromachining techniques for metal patterning. Finally, millimeter wave filters for 60GHz are also designed. Benzocyclobutene (BCB) is selected as the dielectric substrate for the implementation. The same micromachining process used for implementing the CRLH devices is used for this purpose. Two and four pole wideband filters are designed and simulated. The fabrication of the filters is left as a future work of our research, hence, only simulated results and analysis are presented. Figure 3-8 illustrates the 3D 80

81 structure used for the filter implementation on BCB. A single coating of 21m BCB is used. However, multiple coatings are possible in order to increase the substrate thickness SU8 and BCB as Dielectric Materials for RF Circuits Although specific details of the fabrication process is further provided, it is the purpose of this section to discuss the properties of SU8 and BCB (Benzocyclobutene) as dielectric materials for RF applications. For micromachined RF devices, the dielectric materials to be used have two basic requirements: to have appropriate RF electromagnetic properties (dielectric constant and loss tangent) and to be easily micromachinable. SU8 and BCB can be easily spin-coated onto any carrier substrate, such as silicon, glass or printed circuit board organic material. The two dielectric materials are photpatternable, with good optical, electrical and mechanical properties that offer great optical transparency, low curing temperature, the capability for high aspect ratio vertical interconnection useful for multilayer implementations, the compatibility and integrability with CMOS/MEMS processes, and the batch processability for multiple devices. Moreover, the negative tone SU8 resist and the BCB resin are permanent, which is a requirement since the RF devices are to be fabricated on top of the dielectric. Since they offer a permanent dielectric, photolithography, metallization, electroplating and deposition of multiple dielectric layers are possible. SU TM series from Microchem TM and Cyclotene 4026 resin from Dow TM are selected in our study. Table 3-1 summarizes the electrical and mechanical properties of SU8 and BCB. 81

82 Figure 3-8. Cross section of the proposed micromachined cavity resonators and filters. Table 3-1. Properties of the dielectric materials Property SU series BCB Cyclotene 4026 Dielectric Constant 10GHz (2-20GHz) Loss tangent 30GHz 10GHz Thermal stability 5% wt. 315C 1.7% wt. 350C Thermal conductivity 0.3 W/m K 0.29 W/m K at 24 C Coeff. of thermal expansion CTE (ppm) ppm at 25 C Tensile Strength (MPa) Young s modulus (GPa) GPa Curing temperature 150C to 250C 250C UV processing Near UV nm Near UV nm 82

83 CHAPTER 4 COMPOSITE RIGHT/LEFT-HANDED (CRLH) METAMATERIAL APPLICATIONS This chapter introduces some work on single band and multiband composite right/lefthanded (CRLH) metamaterial applications. A dual band three way Bagley polygon power divider is implemented with CRLH transmission lines [21]. Lumped elements are used to realize the left-handed (LH) contribution of the unit cell, while conventional microstrip transmission lines provide the right-handed (RH) contribution. The dual band behavior of the CRLH transmission lines is discussed theoretically. The procedure for a dual band design using the CRLH TL is also provided. Measurement and simulations are compared. In the same way, as an advanced implementation of the CRLH approach, the second part discusses the design, fabrication, and test of surface micromachined compact CRLH unit cells for broadband and dual band applications using the photosensitive SU8 as dielectric. Grounded Coplanar Waveguide (GCPW) CRLH unit cells working up to 8 GHz are implemented. The fabrication process on SU8 serves as the initial stage for implementing CRLH unit cells using BCB as a dielectric, which is studied in Chapter 5. Measurement and simulation results are compared for both cases. At the end, a new CRLH structure with an all-pass behavior and triband response is fully demonstrated and proposed as a unit cell for future work in this area Compact Dual Band Three Way Bagley Polygon Power Divider Using CRLH Transmission Lines A compact dual band three way power divider based on the Bagley polygon is implemented using composite right/left-handed (CRLH) transmission lines consisting of microstrip lines and lumped elements for the GSM frequencies of 860 MHz and 1.92 GHz [21]. Also, a dual band power divider consisting of conventional quarter wavelength (λ/4) transmission lines with shunt connections of open and short stubs has been implemented for comparison. An advantage of using the Bagley polygon for three way power divider is that it 83

84 allows an arbitrary phase selection at the third port by using single or dual band, CRLH or conventional transmission lines. The CRLH based power divider shows an area of 7.95 cm 2 and a fractional 3 db fractional bandwidth of approximately 8% at both bands while the comparison structure shows an area of cm 2, a bandwidth of 5.8% at 860 MHz, and 2.6% at 1.92 GHz. Less than 5.5 db insertion loss is achieved in both cases. Also, full wave structure simulations are performed and the results agree well with those of measurement Compact Bagley Polygon Power Divider The odd N way Bagley polygon power divider is a symmetric structure using λ/4 and λ/2 transmission lines. Figure 4-1 shows the original three way Bagley polygon and its equivalent circuit [57]. The characteristic impedances of the λ/4, λ/2 transmission lines, and the ports are Z q, Z h and Z o, respectively. Due to the symmetry of the circuit, the impedance connected at port 2 is Z o /2 and the input impedance at port 2 is Z o /3. The impedance seen at the matched port 1 should be Z o, therefore the characteristic impedance for the quarter wavelength input transformer is given as Z q = 2Z o /3. The value of Z h does not affect the matching condition, but usually is taken as the same value of Z q. In [86] the very long λ/2 transmission line is replaced by an arbitrary flexible electrical length line, L a, providing phase control at the third port. The matching condition is achieved by specifying the characteristic impedances Z a for the new line and Z q for the quarter wavelength transformer at the input. According to the analysis performed in [86] for a 3 way divider, the characteristic impedance of the lines connecting port 3 to ports 2 and 4, Z a, is given by 2Z o. The characteristic impedance of the λ/4 transformer at the input does not change. In this work dual band CRLH and conventional powers dividers of this type are implemented. Results from measurements and full-wave simulation are compared and analyzed. The dual band CRLH theory is first studied in the next section, and then the design, simulation and fabrication of the power dividers are addressed. 84

85 A Figure 4-1. Conventional Bagley polygon three way power divider. A) Layout schematic. B) Equivalent circuit Dual Band CRLH Transmission Line Theory Review Some work describing dual band quarter wavelength transformers uses a combination of conventional transmission lines with open stubs and short stubs [87]. By using these structures dual band operation is achieved over a moderate range of frequency ratios that depend on the limitation of the fabrication process. The use of very low and very high characteristic impedance for implementing the lines and stubs, in combination with the electrical length depending on the frequency ratio, does not allow significant size reduction and frequency ratios greater than 4.9 and smaller than By using the non linear dispersion property of the CRLH transmission lines [16], the compact Bagley Polygon power divider can be implemented for dual band operation at two arbitrary frequencies f 1 and f 2 with a broader frequency ratio. The quarter wavelength transmission lines connecting ports 1 and 2, and 1 and 4 are replaced by dual band CRLH transmission lines with a phase delay φ = +90 at f 1 and φ = -90 at f 2. The lines connecting ports 3 and 2 or 3 and 4 can be designed by using single or dual band, conventional or CRLH transmission lines depending on the application. One single band conventional transmission line is used in this work for comparison purposes. The theoretical analysis of dual B 85

86 band CRLH transmission lines is presented below. The CRLH unit cell consists of a righthanded (RH) series inductance L R and a shunt capacitance C R which create a conventional transmission line, and a left-handed (LH) series capacitance C L and a shunt inductance L L which create the artificial transmission line with left-handed properties. The characteristic impedance for the right and left-handed contributions, respectively, are given by L Z R or C and R L Z L ol C. (4-1) L unit cell by Series and shunt resonant frequencies, se and sh, respectively, can be defined for the 1 se and L C R L 1 sh. (4-2) L C L R When the unit cell is balanced, the resonant frequencies are equal which means that L R C L =L L C R and Z or = Z ol, then the characteristic impedance Z o of the CRLH unit cell is given by Z o L L R L CR C. (4-3) L By periodically cascading N balanced unit cells a CRLH transmission line can be implemented with a positive phase in the left-handed region. The phase of a balanced CRLH transmission line with N unit cells can be approximated by using the following equation [16]: 1 N LRCR. (4-4) w LLCL Figure 4-2 shows the phase response of the LH, RH and CRLH transmission lines. The non linear behavior of the LH TL in combination with the linear phase of the RH TL creates a controllable non linear phase for the CRLH TL. A positive phase + and a negative phase - can be achieved at two different arbitrary frequencies 1 = 2f 1 and 2 = 2f 2 harmonically not related. 86

87 87 Therefore, /4 transmission lines for dual band operation can be implemented and size reduction is achieved. Figure 4-2. CRLH phase response. A) Phase responses of LH, RH and CRLH unit cell. B) Schematic of a CRLH unit cell. For the design, the two targeting frequencies are selected and the phases + and - required at each frequency are specified. The equation system created by Equations 4-3 and 4-4, evaluated at each frequency, are solved for L R, C R, L L and C L. Equations 4-5 to 4-8 show the expressions for calculating the lumped element values of the CRLH transmission line. These equations are in the function of the frequency ratio 2 / o L Z N C (4-5) N Z L o L (4-6) N Z C o R (4-7) N Z L o R (4-8) B A

88 The number of unit cells, N, is selected according to the requirements. The negative phase contributions, which are the electrical length of the RH TL needed to achieve the phase specifications at f 1 and f 2, respectively, are given by N L C 1 1 R R, and N L C 2 2 R R. (4-9) Design of the CRLH and Conventional Dual Band Quarter Wavelength Transmission Lines. The quarter wavelength transmission lines used in the Bagley polygon have a characteristic impedance Z q = 57.74, with Z o =50. The lines L a have a characteristic impedance Z a = 2Z o = 100. By following the procedure outlined in [87], conventional /4 dual band TLs are designed for operating at GSM frequencies of 860 MHz and 1.92 GHz. Figure 4-3 shows the design schematic. The frequency ratio is 2.23, which gives an electrical length for the lines and stubs of = The characteristic impedance of the lines is Z line = 39.41, and the characteristic impedances for the open and short stubs are selected to be Z open = and Z short = 20, respectively. Meander lines allow an additional size reduction in microstrip. An Arlon Diclad 880 (Arlon Materials for Electronics) substrate with r = 2.2 and a thickness h =31 mil is used. The design is optimized in order to achieve the specifications. By following the CRLH dual band design procedure explained in section 4.1.2, quarter wavelength transmission lines are designed. Phases 1 = +90 at f 1 =860 MHz and 2 = -90 at f 2 = 1.92 GHz are desired for dual band operation. Frequency ratios higher than 5 and smaller than 2 are also achievable, limited by the value of the lumped elements and their self resonance frequency (SRF), especially L L, which is kept below 20 nh with SRF = 6 GHz for GSM frequency designs. Three unit cells with a phase of each cell 1 of 30 at f 1 are selected. Table 4-1 summarizes the calculated and selected commercial values for the lumped elements and the 88

89 phases of the LH, RH and CRLH lines. Surface mounting device inductors and capacitors (Taiyo Yuden, Inc.) are employed for the implementation, and the Arlon Diclad 880 substrate is used. Due to parasitic effects, the lumped elements are selected based on their manufacturer provided frequency response to have effective component values and behavior close to those of calculation at the operating frequencies. Figure 4-4 shows the layout of the dual band CRLH /4 transmission line. A B Figure 4-3. Dual band quarter wavelength transmission line with shunt connections of open and short stubs. A) Schematic (Adaptation of Figure 2(a) from H. Zhang and H. Xin, Designs of Dual Band Wilkinson Power Dividers with Flexible Frequency Ratios, in IEEE MTT-S Int. Microw. Symp. Dig, pp , Copyright 2008, with permission from IEEE). B) Implementation in microstrip. W 3 =3.6 mm, W 1 =0.66 mm, W 2 = 8.04 mm, L 3 =47 mm, L 1 =41 mm, L 2 = 42 mm. Table 4-1. Parameters of the unit cell. Parameter Calculated Real Implementation C L 3.379pF Taiyo Yuden 2.7pF EVK105CH2R7 L L nH Taiyo Yuden 15nH HK1005_15N C R 1.362pF Zo=57.74 Transmission Line L R 4.538nH LH at f 1, at f at f 1, at f 2 RH at f 1, at f RH TL at f 1, RH TL at f 2 CRLH 30at f 1, -30 at f at f 1, at f 2 89

90 Figure 4-4. Layout of the dual band CRLH /4 transformer. 2C L is implemented with pf Implementation. Two Bagley polygons are implemented in microstrip using the design procedures described in the previous section. Full wave structure simulations and circuital co-simulations using High Frequency Structure Simulator (HFSS v. 10, Ansoft Inc.) and Ansoft Designer v. 2.0 (Ansoft Inc.) are performed in order to evaluate the performance of the power dividers. Figure 4-5 shows the implemented dual band Bagley polygon power dividers. The fabrication is carried out by using a CNC milling machine. For the conventional dual band architecture in Figure 4-5A, the total lengths and widths of the /4 TL are L 3 = 47 mm, L 1 = 41 mm, L 2 = 42 mm, W 2 = 3.6 mm, W 1 = 0.66 mm, W 2 = 8.04 mm, as previously shown in Figure 4-3. Conventional 50 lines are used for the ports of the power divider with W= 2.40mm. The total area is 51.98cm 2. For the CRLH dual band divider, L Li = 27 mm and W Li = 1.74mm as previously shown in Figure 4-4. The meander lines allow additional size reduction. The total area is 7.95cm 2.Figure 4-6 and Figure 4-7 show the simulation and measurement results of the return loss (S11) and the insertion loss (S21 or S41) for the two architectures. Dual band operation with center frequencies of 860 MHz and 1.92 GHz is clearly observed. The simulation and measurement results show good agreement. A frequency shift of 25 MHz around 1.92 GHz associated with the tolerance of 90

91 the lumped elements and the fabrication process is presented in the CRLH device. Table 4-2 summarizes the test results for both dividers. A maximum insertion loss of 5.5 db within the bandwidth has been observed infigure 4-7. A B Figure 4-5. Implemented dual band Bagley polygon power dividers. A) Conventional transmission lines with shunt connections of open and short stubs. B) CRLH transmission line approach. A B Figure 4-6. Return loss (S11) and insertion loss (S21 or S41) for two implemented Bagley polygon power dividers. A) CRLH TL power divider. B) Comparison of CRLH and conventional dividers 91

92 Figure 4-7. Insertion loss at each port for the two implemented Bagley polygon power dividers. Table 4-2. Summary of power divider measurements. CRLH Conventional Frequency (GHz) Insertion Loss (S 21, S 41 ) Insertion Loss (S 31 ) f o measured dB Bandwidth 7.93% 8.10 % 5.80 % 2.83 % Phase (S 21, S 41 ) Magnitude imbalance 0.02 db 0.06 db 0.10 db 0.21 db (S 21 to S 31 ) Area occupied 7.95 cm cm Summary This section presented a compact dual band three way CRLH Bagley polygon power divider for 860 MHz and 1.92 GHz, and a fully right-handed dual band power divider with shunt connections of open and short stubs for comparison. The power divider offers a degree of freedom for the phase at the third port, since the half wavelength transmission lines used in the original Bagley polygon design are not used here. The dual band properties of artificial transmission lines allow the selection of two arbitrary frequencies with achievable frequency 92

93 ratio higher than 5 and smaller than 2, limited only by the value of the lumped elements and their resonant frequencies instead of the length and impedance of the transmission lines, which is an advantage compared to the conventional dual band /4 transformer approach. CRLH TL based implementation shows an 84% size reduction and an enhanced bandwidth of 8% for both bands compared to the conventional approach. Full wave simulations confirming the dual band operation are in good agreement with measured characteristics. The insertion loss in both cases is kept lower than 5.5dB, which is acceptable for three way power dividers. This work is developed as a preliminary step towards the implementation of multilayer micromachined dual band metamaterial applications Surface Micromachined CRLH Unit Cell on SU8 for Microwave Applications In this section a highly compact composite right/left-handed (CRLH) unit cell is designed, simulated and implemented at microwave frequencies using a multilayer surface micromachined fabrication process with the negative tone photopatternable epoxy SU8 as a dielectric layer. Grounded coplanar waveguide (GCPW) CRLH unit cells for broadband operation up to 8 GHz are implemented as a first step towards the implementation of transmission lines working at higher frequencies. Metal-insulator-metal capacitors and an SU8 embedded meander inductor are used for a left-handed unit cell. The microfabrication process and measurement test results are discussed in detail. The dispersion diagram of a unit cell designed for 2.4 GHz operation is extracted from the S-parameters, showing broad left-handed behavior from 2 to 5.5GHz. Full wave structure and circuital simulations are compared with measurement results. For higher frequencies of operation, BCB is used in our work to implement finite ground coplanar waveguide (FGC) CRLH transmission lines. The design, simulation, modeling and testing of the fabricated unit cell on SU8 are provided. 93

94 Unit Cell Structure and Modelling Figure 4-8 presents the general structure, simplified equivalent circuit and cross section view of the proposed microfabricated composite right/left-handed unit cell, which has been previously introduced in Chapter 3. Note the structure is patterned in a dielectric embedded fashion, while the dielectric layer and upper ground plane is not drawn for structural clarity in Figure 4-8A.. It is worth to mention that the unit cell can be implemented on a variety of dielectric materials, and for our study, SU8 and BCB have been selected. The via connects the inductor to either the upper or lower ground plane of a CPW, GCPW or Microstrip implementations. The left-handed inductor L L is implemented using a meander line inductor embedded in the dielectric. The metal patches on the lower and upper layers create MIM capacitors, which make the left-handed capacitance contribution C L. The right-handed inductive contribution, L R, is mainly provided by the combination of small pieces of the feeding transmission lines at both sides and the MIM capacitor pads. The right-handed capacitive contribution C R is provided by the combination of the parasitic capacitance between the transmission lines and the ground, and the parasitic capacitance between the meander line and the ground. The left-handed inductance value can be controlled by adjusting the width w i of the inductor trace, the number of turns and the total length L i. Although the inductance of an embedded single layer meander line inductor is smaller than that of the conventional solenoid or spiral inductor, its single layer architecture facilitates the fabrication process. Neglecting fringing effects, the left-handed capacitance value is controlled by the area of the metallic patches and the separation distance d between the upper and lower layers. The total height of the structure is represented by h. The inter-capacitor gap g i is selected to be large enough to be negligible in the left-handed capacitor design. 94

95 W line L line h To carrier substrate Via-Hole A g i l i W cap L cap MIM capacitor d w i SU8 or BCB embedded inductor Via interconnection 2C L L L B 2C L L R /2 L R /2 C R d h 1 BCB or SU8 BCB or SU8 h Copper Low Res Si or Glass Ground plane only for CBCPW and Microstrip C Figure 4-8. The CRLH unit cell. A) Structure of the proposed micromachined CRLH unit cell. B) Simplified electrical equivalent circuit. C) Cross section view of the 3D structure. From the previous description it is observed that the combination of the right-handed and left-handed components L R, C R, L L, and C L creates a composite right/left-handed artificial transmission line, which contains a frequency range with backward wave propagation and positive phase. Equations 4-1 to 4-9, presented in the previous section, are used in this section Implementation and modeling of MIM capacitors using SU8 as dielectric layer In this section, the design, simulation, modeling and parameter extraction of the integrated MIM capacitors on SU8 are discussed. MIM capacitors ranging from 0.3 pf to 4.2 pf are designed and simulated. Simulations are performed on the 3D full wave structure simulator HFSS TM. The electrical equivalent circuit models for the integrated components and the parameter extraction based on the simulated S and ABCD parameters are discussed in detail. The 95

96 design of the CRLH unit cells on SU8 uses grounded coplanar waveguide implementation (GCPW), and hence, bottom and top ground planes are used for this case. Since the CRLH unit cells to be implemented are symmetrical, the design and simulation of the MIM capacitor is done by using a configuration of two capacitors in cascade, as shown in Figure 4-9. In Figure 4-9A, the 3D view of the two capacitors details the geometrical parameters. The cross section of the structure is shown in Figure 4-9B. The top conductor in each side with the dielectric embedded bottom conductor is used to create the MIM parallel plate capacitor, whose geometry and basic formula is shown in Figure 4-9C. It is observed that the two MIM capacitors are connected by the small transmission line gap section on the bottom conductor. The fringing effects and parasitic capacitances to the ground planes are not taken into account for the capacitance formula, however, they are considered in the electromagnetic simulation and modeling of the MIM capacitors. The complete electrical equivalent circuit model of the two capacitors in cascade is shown in Figure The inductor L S models the parasitic inductance of the MIM capacitor due to the metallic trace. The capacitor C S models the parallel plate capacitance of the MIM capacitor, which is basically due to the capacitor created by the top and bottom conductors and the fringing effects. The shunt capacitors C P model the overall parasitic capacitance of the metal traces to the ground plane, which also takes into account fringing effects. The inductor L gap models the small inductance due to the gap and the capacitor C gap models the small capacitance due to the gap. The equivalent circuit is reduced to two capacitors directly connected in cascade if the gap is selected to be long enough to make the gap capacitance C gap neglectable at the frequency range of interest, so it will behave as an open circuit. In the same way, since the gap is a short piece of transmission line, its inductance is very small so the parasitic inductor L gap can be considered a 96

97 short circuit at the frequency range of interest. With these simplifications, the frequency dependent ABCB parameters of the two cascaded parameters can be expressed as in A C B A D C 1 1 B1 A D 1C 2 2 B2, (4-10) D2 where [ABCD] 1 and [ABCD] 2 define the ABCD matrix of capacitors 1 and 2, respectively. In the same way, each capacitor can be define by its frequency dependent admittance matrix (Y) as in Y Y Y Y YCp 1 - ZCs 1 Z Cs -Y - Cp 1 Z Cs 1 Z Cs, (4-11) with Y Cp and Z Cs as the shunt admittance and the series impedance of the capacitor equivalent circuit model, given by Y Cp 1 jc, (4-12) p and Z Cs 1 jls. (4-13) jc p By using the previous equations and the simulation values of the frequency response of the two capacitors in cascade, the extraction of the parameters of the equivalent circuit model for each capacitor can be performed. In the 3D structure simulator the ABCD matrix is obtained for the two cascaded capacitors, and neglecting the gap capacitance and inductance, the [ABCD] 1,2 matrix for each individual capacitor is obtained as ABCD ABCD 1,2.Then an extraction procedure, based on the conversion to Y parameters and the solution of linear system equations, is performed. It is not the aim of the section to show the detailed extraction procedure, but the extracted values of capacitance that are useful for our study are presented in Figure Full 97

98 wave 3D structure simulations are performed from 1 GHz to 8 GHz. Table 4-3 shows the geometrical parameters for the implementation of the MIM capacitors using SU8 as a dielectric. The width W is selected based on the design of a 50 GCPW transmission line on a 100 m thick SU8 layer, which is further explained. d W Gap L A Conductive plates B C Area Dielectric d C 0 r Area d Figure 4-9. MIM capacitors. A) Structure of two cascaded MIM capacitors. B) Cross section view. C) Parallel plate capacitor geometry and formula. Capacitor 1 Capacitor 2 Figure Equivalent electrical circuit model of the two cascaded MIM capacitors. 98

99 Inductance Ls (nh) Capacitance (pf) Capacitance Cp (pf) Cs Cth L(um) A L(um) B Figure Simulation of extracted parameters for the MIM capacitors. A) Extracted series capacitance C S and calculated theoretical value C th. B) Extracted parasitic capacitance C P. C) Extracted parasitic inductance L S. Table 4-3. Design parameters of the MIM capacitors Parameter W d L Gap L(um) Value 200 m 2 m 100 m 750 m 100 m It is observed that by varying the length L of the MIM capacitor, capacitance values ranging from 0.38 pf up to 2.7 pf are obtained. The parasitic parameters are kept very low due to the small size of the capacitors and the relatively thick SU8 dielectric layer between the electrodes and the ground plane. With these values of capacitors, it is possible to implement CRLH transmission lines working in the frequency range of interest. C 99

100 Implementation of meander line SU8 embedded inductors Meander line inductors are selected for implementing the inductive left-handed contribution of the CRLH transmission lines. Due to the multilayer implementation of the unit cell, the inductors are embedded with a distance of 3m from the surface in the selected dielectric (SU8 in this case) and connected to the bottom conductor of the capacitors, which facilitates the fabrication procedure and protects the inductor from oxidation. The geometry of the inductor is presented in Figure 4-12A. A three-turn inductor is used. The width of the trace is w i = 20 m with a separation distance of d = 80 m. The inductance is then controlled by the length L i Figure 4-12B depicts the cross section of the inductor in order to highlight its embedded nature. Table 4-4 summarizes the geometrical parameters. w i L i Bottom Conductor d Via to ground B A Figure Embedded meander line inductor. A) Geometry. B) Cross section. Table 4-4. Design parameters of the inductor Parameter w i D L i Via to ground Embedded distance Thickness trace Value 20 m 80 m 400 m to 1000 m 100 m 100 m 3 m in SU8 1.5 m Copper 100

101 The equivalent electrical circuit of the inductor is shown in Figure 4-13A. The inductor L s models the connection to the bottom conductor, and the inductor L P models the inductance value of the meander line inductor. The parasitic capacitance of the trace to the ground is modeled by the capacitor C P. The conductor loss due to the trace is modeled by the series resistor R S. The extraction procedure is based on the simulation of the frequency response of a transmission line loaded with the inductor, as shown in Figure 4-13B. Neglecting the inductance L s, which is used only for simulation purposes, the admittance of the inductor is defined by the Equation Finally, the parameters C P, L P and R S are extracted from the simulated frequency response of the inductor. Y i jc P 1 jl R P S. (4-14) A Figure Inductor modeling. A) Electrical equivalent circuit. B) Simulation setup in 3D structure simulator HFSS TM. Figure 4-14 shows the extracted values for a variation in the inductor length, L i, from 400 m to 1000 m. It is observed that the parasitic capacitance is kept with a low value, mainly due to the thin 20 m inductor trace and the relatively thick SU8 dielectric layer of 100 m. Inductance values from 1.35 nh up to 2.95 nh are obtained with this variation. The resistor B 101

102 Value Capacitance Cp (pf) values, from 1.2 to 2.8, mainly due to the skin depth in Copper, can be reduced if the thickness of the inductor trace is increased. Full wave structure simulations from 1 GHz to 8 GHz are performed Rs (Ohms) Li(um) Figure Extracted parameters for the inductor. A) Inductance L P and series resistance R S. B) Parasitic capacitance C P Implementation of CRLH Unit Cells. A Lp (nh) As previously studied, the physical implementations of MIM capacitors and meander line inductors have additional parasitic capacitances and inductances, which depend on the dimensions of these components and can change the behavior of the unit cell if they are not compensated. Figure 4-15 shows the full equivalent circuit of the unit cell taking into consideration the parasitic effects. The complete equivalent circuits of the MIM capacitors and meander line inductors are included. The small inductance of the gap is split into two in order to be used for a symmetrical unit cell. The right-handed contribution due to the conventional transmission lines are modeled by the capacitor C R and the inductor L R, which are split into two because a symmetrical representation of the unit cell is used. Conductor and dielectric losses are modeled by the resistor R C. The resistor R S models the intrinsic resistance of the inductor trace, which takes into account the metal resistivity and the skin depth effect Li(um) B 102

103 C gap R C /2 L R /2 L CP 2C L L gap /2 L gap /2 2C L L R /2 R C /2 L CP C R /2 C CP C CP C CP C CP C R /2 L L C L R S Figure Complete equivalent electrical circuit of the CRLH unit cell including parasitic contributions. A B Figure General representation of the unit cell structure. A) Layout of top microstrip or coplanar waveguide line and the embedded inductor. B) Cross section view. Transitions are used for measurement purposes. CPW top ground planes are not shown for clarity. The via can be connected to either upper or lower ground planes in GCPW implementations. Two different unit cells are implemented in this work: one unit cell with a 90 electrical length at 2.4 GHz, and one unit cell for dual band operation with 45 electrical length at 2.4 GHz and -45 at 5.8 GHz. The dual band unit cell is designed to be used in the design of a dual band CRLH quarter wavelength transmission line. The implementation of the unit cell is done on SU from Microchem TM. Figure 4-16 presents the layout of the top transmission line, the 103

104 SU8 embedded inductor, and the cross section view of the unit cell. For this application, the height of SU8 substrate is selected to be 100 m, which is achieved by double coating of SU on a carrier substrate. SU series are well known to have good thermal stability, good adhesion, single coating thick films and high resistance to solvents and electrochemical processes. The inductor is embedded in SU8 3 m below the top layer, with a trace thickness of 1 m, which is close to a skin depth of copper at 2.4 GHz. The width of the implemented inductors is 20 m, which is selected in order to decrease any parasitic capacitance. As previously discussed, the inductors are implemented with three turns with 80 m spacing and variable length according to the required inductance value. The MIM capacitors are implemented with the bottom and top metallic layers. The interlayer distance d is 2 m in order to achieve high values of capacitance while keeping the overall area small. The width of the capacitors is selected to be 10 m less than the width of the conventional top transmission line, giving a good tolerance range to the fabrication process. The theoretical capacitance is calculated based on the formula for the parallel plate capacitor. Further optimization processes are performed in order to achieve design specifications. Table 4-5. Parameters of the unit cells Unit Cell L R (nh) C R (pf) L L (nh) C L (pf) Broadband 90 at 2.4GHz Dual band 45 at 2.4GHz and 5.8GHz Grounded coplanar waveguide implementations have been selected for this work. The characteristic impedance of the top GCPW line is 50, which allows the calculation of its width. The length of the top transmission line depends on the phase requirements of the unit cell. The design is based on Equations 4-1 to 4-3 and the dual band design procedure described in section 104

105 Table 4-5 summarizes the calculated parameters of the two designed unit cells. In the same way, Table 4-6 summarizes the dimensions of the two implemented unit cells in m. Table 4-6. Dimensions of the unit cells in m Unit Cell w cap h h1 d L cap L ind W line w ind Broadband 90 at 2.4 GHz Dual band at 2.4 GHz and 5.8 GHz Figure 4-17 compares the electromagnetic and circuital simulation results of a balanced quarter-wavelength unit cell for 2.4 GHz operation. A loss tangent of 0.02 is used for the SU8. Less than 0.9 db insertion loss at 2.4 GHz is expected. The circuital simulation is performed with the extracted values. It is observed that the full wave electromagnetic simulations show the expected results for the designed unit cells. The next section describes the fabrication process and measurement results Fabrication Process In this section the procedure used for the fabrication of the unit cells is described. This fabrication procedure will be used for the micromachining of all the multilayer CRLH devices to be implemented in this research. For demonstration purposes, a printed circuit board (PCB) FR-4 substrate has been selected for the fabrication, although the fabrication can also be done on silicon and glass wafers as carrier substrates. The fabrication procedure is based on micromachining techniques such as metallization using DC sputtering and electroplating, multilayer coating of SU8-2025, conventional UV lithography of SU8 and negative photoresist, and cleaning procedures [85]. Figure 4-18 illustrates the fabrication procedure. First, substrate cleaning is performed with TCE (trichloroethylene), followed by a rinse with Isopropanol, DI water and dehydration in a vacuum oven at 120 C for 10 minutes. Further, a seed layer of 30 nm / 300 nm of Titanium/Copper is sputtered to create the ground plane, followed by electroplating 105

106 S11 and S21 (db) 5 m Copper. Finally, a thin layer of 30 nm Titanium is sputtered as an adhesion promoter for the SU layer to be coated Frequency (GHz) Figure Comparison of the electromagnetic and circuital simulation for the broadband CRLH unit cell with a 90 phase at 2.4 GHz. Extracted values are C L = 0.65 pf, C CP = pf, L CP = nh, L L =2.6 nh, R S = 2.8, C LP = pf. The total length of the transmission line is 2.3 mm. Inter-capacitor gap is 100 m. Following the formation of the ground plane, SU is coated in a two-step process in order to achieve better uniformity. Both coating steps are performed at 1500 RPM for an approximate thickness of 55 m. Edge bead is removed. Soft bake is performed on a hot plate with the temperature ramped up to 65C at a rate of 250C/hour and kept for 30 min. Temperature is ramped up to 95C at the same rate and kept for 20 minutes. At the end of the soft baking process the samples are allowed to cool down to room temperature on the hot plate. The second coating step is then performed and the soft baking time at 95C is increased to 30 minutes. Lithography with conventional UV light (365nm) is performed with with a dose of 240 mj/cm 2. Post-bake is realized performed on a hot plate with the temperature ramped up to 65C at a rate of 250C/hour and kept for 5 minutes. Temperature is then ramped up to 95C at 106 S11 EM sim S21 EM Sim S11 Circuit Sim S21 Circuit Sim

107 the same rate and kept for 10 minutes. The samples are developed in SU8 developer for 60 minutes and finally rinsed with Isopropanol and blow dry with nitrogen gun. Figure Fabrication process for the multilayer CRLH devices. Further, a metallic seed layer of Ti/Cu/Ti (30 nm/300 nm/30 nm) is sputtered on top of SU8 followed by conventional negative photoresist patterning of the embedded inductor and bottom patch of the capacitor by using negative resist NR Electroplating a 1.5m Copper layer is performed with a prior etching of the top Titanium layer used for protecting Copper from oxidation. After electroplating, NR is removed and also the seed layer of Ti/Cu/Ti is time-etched. This step creates the bottom conductor of the MIM capacitors and the meander line inductor, along with the vias connecting the ground planes. The next step consists of the coating of a 4 m think layer of SU to create the interlayer dielectric for the MIM capacitor. Soft bakes is performed on a leveled hot plate with the temperature ramped up to 95C at a rate of 107

108 250C/hour and kept for 5 minutes. Exposure is performed with a dose of 150mJ/cm 2. Post bake uses the same temperature profile used for soft bake. A near 50% planarized SU layer is obtained with the coating of 4 m of SU8 on top of a 1.5 m thick and 200 m width metal layer, which gives 2 m SU8 layer needed for the capacitors. Finally, a layer of Ti/Cu/Ti (30nm/300nm/30nm) is again sputtered, followed by 10 m thick negative resist NR-9800 coating. Lithography is performed with a dose of 300 mj/cm 2 and the top layer pattern is created. Titanium etching is performed with diluted hydrofluoric acid (HF) in de-ionized water (DI) with a 1:10 ratio. Electroplating of a 5 um thick Copper layer is performed to create the top line and capacitor top conductors. At the end, the seed layer of Ti/Cu is time-etched. Figure 4-19 shows scanning electron microscope (SEM) photographs of the fabricated unit cell Measurement Results Measurements are performed by using a Cascade Microtech Probe Station connected with an Agilent E8361A Vector Network Analyzer. Short-Open-Load-Thru (SOLT) Calibration is done from 1 GHz to 8 GHz. Figure 4-20shows the measurement setup. Figure 4-21shows the measured return and insertion loss for the 50 ohms broadband CRLH unit cell. Less than 2 db insertion loss is achieved in the left-handed range with around 3 db at 2.4 GHz, mainly due to the tolerance in the fabrication process and difference in the dielectric constant and loss tangent of the SU8. Figure 4-22 shows a broadened left-handed behavior ranging from 2 GHz to 5.5 GHz with 90 phase at around 2.4 GHz. On the other hand, since the area of the unit cell is very compact, around 5.6 mm 2, an area reduction close to 90% is achieved when compared with conventional quarter wavelength transmission line based PCB implementation on a substrate with the same dielectric constant. 108

109 A C B Figure Photographs of the microfabricated broadband CRLH unit cell. A) SEM picture of the unit cell, B) Close look of the embedded inductor, C) Cross section view of the unit cell. Total area is around 5.6mm 2. Figure Measurement setup consisting of a Cascade Microtech probe station and an Agilent E8361A VNA. 109

110 Frequency (GHz) Figure Insertion and return loss for a balanced microfabricated CRLH unit cell Bp Meas Bp Sim Figure Dispersion relation for the microfabricated CRLH unit cell. The electrical length around 2.4GHz is Summary Bp (deg) This section has presented the design, simulation and fabrication of a super compact multilayer micromachined CRLH unit cell by using SU8 as a dielectric layer. Circuital and 110

111 electromagnetic simulations show good agreement with measured data. Almost 90% size reduction, in comparison with one fabricated by conventional PCB processes is achieved for the unit cell. Although insertion loss seems to be large in comparison with that of conventional implementations, mainly due to the relatively large loss tangent of SU8 and tolerances in the fabrication process, the surface micromachined process on SU8 still can be used for achieving compact circuits at microwave frequencies. This preliminary step has developed the design procedure and fabrication process to be used for the fabrication of CRLH devices working at higher frequencies Bridged Composite Right/Left-Handed Unit Cell with All-Pass Behavior This section studies theoretically and experimentally a modified design of the composite right/left-handed (CRLH) unit cell showing all-pass behavior and triband response [88]. By using an additional inductance which cross-couples the input and output ports of the conventional CRLH, a bridged-crlh unit cell (B-CRLH) is created. This allows additional right-handed (RH) wave propagation below the cutoff frequency of the conventional CRLH to DC, forming a new low-frequency-rh band with a non-linear dispersion relationship. Meantime, a mid-frequency-lh band and a high-frequency-rh band, which are inherited from the conventional CRLH, are still present, resulting in three-band configuration. The dispersion relation and Bloch impedance of the B-CRLH are investigated by standard periodic analysis. Balanced conditions to close transitions between the three bands, to realize all-pass behavior, are derived. The proposed structure is fully investigated with analytical calculations, circuital and full-wave simulations, and physical implementation. Furthermore, the triband operation is demonstrated with an open-stub configuration. For demonstration purposes, this B-CRLH is implemented on PCB technology by using a double layer fashion that combines metal-insulatormetal (MIM) capacitors, meander line and stub inductors, and a patterned ground plane. 111

112 L B L R /2 2C L 2C L L R /2 Z 1 Z 2 Z 2 L L C R Z 3 p A B Figure Proposed topology. A) Circuit model of the bridged-crlh (B-CRLH). B) Bridged-T topology for analysis Proposed Bridged-CRLH Figure 4-23A shows the circuit model of the proposed bridged-crlh unit cell in the case its size p is much smaller than the guided wavelength ( p << g ). The inductor L B cross-couples the input and output ports of the B-CRLH unit cell. The two series LC resonators defined by Z se jlr 2 1 j2c L, and the shunt LC resonator Ysh jc R 1 jll are inherited from the original CRLH structure. In order to facilitate the analysis, we start with the balanced case where the series and shunt resonators have the same resonance frequency se = sh = o. The left-handed and right-handed characteristic impedances of the CRLH unit cell are given by Z ol LL CL and or LR CR Z, respectively [6]. In the low frequency range, below the lefthanded cutoff frequency cl of the original CRLH unit cell ( < cl ), the series LC resonator is mainly capacitive, while the shunt LC resonator is mainly inductive. Meantime, the inductance L B, with a low impedance value, cross-couples the input and output ports and the B-CRLH structure is considered a bridged-t left-handed unit cell with a right-handed wave propagation at low frequencies i.e. below the low cutoff frequency. At frequencies higher than the low cutoff frequency and smaller than the resonance frequency ( cl < < o ), the series and shunt resonators keep their capacitive and inductive behavior, respectively, while the inductance L B 112

113 becomes a high impedance value. Then, the behavior of the B-CRLH unit cell is that of an LH transmission line with backward wave propagation. It is noticed that the operation of the unit cell has a transition from RH to LH operation in the low frequency band, as that of the dual-crlh (D-CRLH) structure [17], with a possibility of a seamless all-pass behavior by choosing an appropriate L B value. At higher frequencies, ( 0 < < cr ), where cr is the high cutoff frequency of the CRLH unit cell, the behavior of the B-CRLH unit cell is that of an RH transmission line. Therefore, the proposed topology exhibits two RH and one LH frequency bands with non-linear dispersion behavior and with the possibility of all-pass behavior. The conditions for seamless transitions from alternating RH-LH-RH bands are analyzed Analysis Bloch analysis is used to investigate the operation of the unit cell. The Bloch complex propagation constant and impedance of the unit cell are given by p jp cosh 1 ( A), (4-15) and B Z B, (4-16) A 2 1 where p and p are the attenuation and phase-shift per unit cell of size p, respectively. The bridged-t configuration in Figure 4-23B is used in the analysis. The A parameter of the ABCD matrix of the network is given by where, Z A 1 1, (4-17) Z 2 Z 1 jlb, Z2 jlr 2 1 j2c L 3 Z 2 Z 1 Z 2, and Z jl L C 2 [57]. Replacing, 3 L L C B L sh se A 1 2, (4-18) se sh L B 113 L R

114 where se 1 LRCL, sh 1 LLCR are the series and shunt resonance frequencies, respectively, L 1 L L C L is related to the cutoff frequency of a pure left-handed (PLH) transmission line unit cell, and B 1 L R L B C L is the resonance frequency of the LC resonator that would be created by the bridge inductor L B and the series branches of the B-CRLH unit cell. It is observed that at the shunt and series resonance frequencies the A parameter is 1, which means that the complex propagation constant is zero, = 0 and = 0, as in the original CRLH. For the balanced case it is assumed se = sh = 0, and Z or = Z ol = Z o. When the frequency is = B, the A parameter is reduced to 2 L C A 1, (4-19) 1 2 B B 2 B L 2 se where after replacing expressions, A = -1, which means that the complex propagation constant is + j = 0 + j. It is worth to notice that the point = B does not necessarily represent a seamless transition from the right-handed (RH) to the left-handed (LH) behavior of the B-CRLH. From Equations 4-18 and 4-19 it is observed that a seamless transition ( = 0) is possible if an appropriate condition is presented. In order to obtain the all-pass behavior of the network, where the propagation constant is purely imaginary, one more condition is evaluated at the left-handed cutoff frequency cl of the original CRLH unit cell, forcing = cl = B, as given in cl 1 1, (4-20) R L R where R 1 L R C R. After some mathematical steps, L B is calculated as in L B cl 1 se, (4-21) C L which defines the optimum value of the inductor L B needed in the B-CRLH unit cell to achieve 114

115 the all-pass behavior with = 0, = and a seamless transition from RH to LH operation at = B = cl. Meantime, when the frequency L << R, the LH cutoff frequency of the original CRLH unit cell is approximately cl L /2 [11]. By replacing terms in Equation 4-21, the inductor L B can be approximated to L B = 4L L L R, which offers a simpler expression and an initial value that can be further optimized. w b g g u l c l c g w i l i Ground A B C Figure Physical configuration of the B-CRLH unit cell. A) Top layer view of the geometry. B) Bottom layer showing the stub inductor and patterned ground plane with g u = 2 mm. C) Top layer of the implemented unit cell. D) Bottom layer Physical Implementation To demonstrate the design concept, the B-CRLH unit cell is implemented on a double side fashion. The substrate Arlon DiClad 880 with a thickness of mm and a dielectric constant of r = 2.2 is used. Figure 4-24A and Figure 4-24B show the geometrical configurations for the top and bottom layers, respectively. Figure 4-24C and Figure 4-24D show the top and bottom views of the physical implementation. Square MIM capacitors with a side length of l c = 5.4 mm 115 D

116 are used to provide the LH capacitive contribution, C L. A stub inductor with a width w i = 0.25 mm and a length l i = 3.35 mm is patterned on the ground plane to implement the LH inductor L L. The RH capacitor, C R, and inductor, L R, are provided by the parasitic effects of the MIM capacitors and the stub inductor. The bridged inductance L B is implemented on the top layer using a meander line structure with a total length of 17.3 mm and a width of w b = 0.25 mm, while the ground plane underneath is removed in order to reduce its parasitic capacitance. Feeding lines at both ends of the unit cell, with a 50 characteristic impedance, are used for testing purposes, which are de-embedded from simulations and measurements. The unit cell is designed based on the previous theoretical analysis. Figure 4-25A shows the dispersion diagrams for the circuital simulation from 300MHz to 8GHz of balanced and unbalanced unit cells, and for the full-wave simulation of a balanced unit cell using Ansys Designer V.6.1. It is observed that the circuital and full-wave simulations show good agreement, with differences at higher frequencies due to additional parasitic effects not taking into account in the circuit model. In Figure 4-25A, seamless transitions from RH-LH-RH bands are observed, as previously predicted, where the B-CRLH unit cell shows all balanced conditions. The first RH region is observed from 300 MHz to 1.8 GHz. The LH band is from 1.8 GHz to 2.8 GHz. The second RH region is observed up to 6.5 GHz. It is also observed that the unbalanced case has stop bands between the RH-LH-RH regions. Simulated and measured frequency responses, presented in Figure 4-25B, show good agreement. In addition, in Figure 4-26A the implementation of a 50 /4 open stub made of two B-CRLH unit cells is presented, while the simulated and measured results are shown in Figure 4-26B. The inset of Figure 4-26B shows the open stub schematics. In Figure 4-26B the triband behavior of the B-CRLH structure is highlighted. Due to the continuous operation, phase angles per unit cell of 45 and 135 are achievable, which gives 90,

117 S 21 (db) Frequency (GHz) S 11, S 21 (db) for a two-unit cell TL. Because of this, two additional peaks appear between the frequencies f 1 and f 2 of interest, as observed in Figure 4-26B. However, these peaks can be eliminated if the allpass condition is relaxed and an unbalanced implementation is used. Complete details of the multiband behavior are left as a future work. EM Sim. Circuit Sim. Bal. Circuit Sim. Unbal S 11 S 21 unbal f sh unbal bal -20 f o f B f cl Frequency (GHz) x 10 9 p A B Figure Simulated and measured results of the B-CRLH unit cell. A) dispersion diagrams for balanced and unbalanced cases. B) frequency response of the balanced unit cell. Unit cells parameters are L R = 2.23 nh, C R = 0.89 pf, L L = 2.27 nh, C L = 0.91 pf, and L B = nh for the balanced case. C R = 0.59 pf and L B = 5 nh for the unbalanced case Measurement Simulation A f 2 f 3-25 Measurement f 1 Simulation Frequency (GHz) 6 8 Figure Triband two B-CRLH unit cells /4 open-stub. A) Photograph of the fabricated stub. B) Insertion loss. Operation at f 1 = 0.61 GHz, f 2 = 2.4 GHz and f 3 = 3.5 GHz are highlighted. Additional peaks are due to the continuous mode of operation. B Z o, /4 117

118 Summary A bridged CRLH (B-CRLH) metamaterial unit cell design with an all-pass behavior consisting of two RH bands and one LH band was proposed and analyzed. The conditions for achieving seamless transitions between RH-LH-RH bands were provided. The B-CRLH topology also enables multiband operation as the CRLH does, while the B-CRLH offers the possibility of more band selections due to the additional low-frequency-rh band. Also, an achievable low frequency band could be much lower than that of the original CRLH, which allows implementing devices with much greater size reduction. The B-CRLH working principle was validated both numerically and experimentally. Also, a triband operation was fully demonstrated. 118

119 CHAPTER 5 MICROMACHINED METAMATERIAL UNIT CELLS ON BCB This chapter explores the implementation of highly compact three dimensional (3D) integrable metamaterial based transmission lines on a low resistivity (10-20.cm) CMOS grade silicon substrate for microwave and millimeter wave applications. The composite right/lefthanded (CRLH) architecture is able to be integrated with an integrated circuit (IC) using a multilayer surface micromachined fabrication process as a post-cmos process. The fabrication process employs the negative tone photo sensitive Benzocyclobutene (BCB) as a low-loss dielectric interlayer material allowing packaging compatible high performance RF circuits. Since the low temperature and multilayer fabrication is compatible with CMOS/MEMS processes, it allows the batch fabrication of multiple devices and the easy implementation of 3D vertical interconnects. Finite ground coplanar waveguide (FGC), widely used in CMOS and MMIC design [36-37], has been selected for the implementation of multilayer CRLH unit cells and transmission lines suitable for broadband and multiband microwave and millimeter wave applications. The design, modeling, fabrication and on-wafer characterization are presented for 50 compact multilayer finite ground coplanar waveguide (FGC) CRLH unit cells and transmission lines for broadband and multiband operation at Ku and Ka frequencies of 14 GHz and 35 GHz, respectively. Also, the comparison between the simulation and measurement results up to 40 GHz on the aforementioned 3D electromagnetic structures is provided. The left-handed capacitance and inductance components of the CRLH structures are implemented with photolithographically defined Metal-Insulator-Metal (MIM) capacitors and BCB embedded meander inductors, respectively, which allows the fabrication of very compact CRLH devices. The fabricated dual band unit cell features a size of 0 /30 at 14 GHz and an insertion loss of less than 2 db within the passband [89]. 119

120 5.1. Analysis of Finite Ground Plane Coplanar Waveguide Transmission Lines on BCB A CMOS grade silicon wafer with a thickness of 280 m and a resistivity of 10.cm is used in this work. Prior to the design of CRLH unit cell, a finite ground coplanar waveguide transmission line on BCB is designed, implemented and tested in order to analyze the attenuation offered by this type of line on BCB. Figure 5-1A presents the cross section view of a finite ground plane CPW line implemented on the low resistivity silicon with a Benzocyclobutene (Cyclotene from Dow Chemical, r = 2.65, tan = at 10 GHz ) interface layer. Figure 5-1B shows the side view of the unit cell to be implemented. For the implementation, the BCB thicknesses are h 1 = 14 m, h = 21 m and the inter-capacitor distance, d, to create the MIM capacitors is 5 m. The selected CPW gap G, width W, and ground plane length W G for a 50 feeding line are 20 m, 55 m and 440 m, respectively. Sputtered Ti/Cu/Ti (30 nm / 300 nm/ 30 nm) and subsequent electroplated Cu with thicknesses of 2 m and 5 m are used for the embedded and top metal layers, respectively. Different values of the BCB thickness h, are used for the analysis of loss.va loss analysis is performed based on simulation results. Figure 5-2A shows the simulated attenuation of a 2 mm long CPW line with different values of the BCB thickness, h. It is observed that the thicker the BCB layer is, the smaller the loss is, mainly due to the minimization of the interaction of the electromagnetic field with the lossy silicon substrate. Figure 5-2B shows the measured performance when a 21 m thick BCB layer is used, featuring an attenuation smaller than 0.3 db/mm at 40 GHz. In our study, the first layer of BCB is 14 m, and the second layer is 7 m thick in order to achieve a total thickness of 21m. A similar procedure can be used to achieved thicker layers of BCB, however, 21 m is selected in our work. 120

121 Attenuation (db/mm) S 11 (db) S 21 (db) t G W G Ground Benzocyclobutene (BCB) h Low Resistivity Si (10.cm) d h 1 L cap BCB Top metal Embedded Metal h BCB Low Resistivity Si (10.cm) A Figure 5-1. Cross section view of the CPW structures. A) CPW on low resistivity silicon with a Benzocyclobutene (BCB) interface layer. B) Multilayer architecture for the CRLH implementation. B h=28um Frequency (GHz) Figure 5-2. Insertion loss of a CPW line on a low resistivity silicon substrate with a BCB interface layer: W = 55 m, G = 20 m. A) Simulated attenuation, B) Measured and simulated return loss and insertion loss for a 2 mm long CPW line with BCB thickness h = 21m. Figure 5-3 shows the electric field in the cross section of the CPW when the BCB thickness h is 21 m, which minimizes the interaction of the electromagnetic field with the silicon substrate. The interaction of the electromagnetic field with the silicon substrate results in an eddy current loss, which is significant when low resistivity silicon is used. For this reason, this interaction needs to be minimized. The BCB with a thickness of 21 m, which offers a modest insertion loss and can be achieved by two coating steps, is selected in our study. However, the use of thicker BCB and less lossy substrates (such as glass) will decrease the loss even further. A h=14um h=21um S 21 S 11-3 S 11 Sim -30 S 11 Meas S 21 Meas S 21 Sim Frequency (GHz) B

122 Figure 5-3. Electric field in the cross-section for a CPW line with W = 55 m, G = 20m Design and Implementation of the Dual Band CRLH Unit Cell and Transmission Line By following the dual band CRLH design procedure described in section 4.1.2, a CRLH unit cell is designed to implement a quarter-lambda two unit cell transmission line with dual band operation at Ku and Ka band frequencies of 14 GHz and 35 GHz, respectively. The same unit cell structure previously shown in section 4.2.1, recalled here for convenience in Figure 5-4, is used for the implementation. Calculated parameters are C R = 0.12 pf, C L = pf, L R = nh and L L = nh. Table 5-1 shows the geometrical parameters of the CRLH unit cell implemented by using the structure introduced in Figure 5-4A. The first and second layers of BCB are 14 m and 7 m thick, respectively. The inter-capacitor distance is 5m, which is achieved as the result of a planarization of 71% for a BCB layer over a 2 m thick and 250 m wide copper trace. Small pieces of CPW line with a width W line of 55 m and a length L line of 40 m are used at both feeding sides to compensate for the right-handed contribution. Table 5-1. Dimensions in m of the CRLH unit cell on BCB. W Cap h h 1 d L cap L ind W line w ind

123 L line W cap MIM capacitor Via interconnection W line h To carrier substrate Via-Hole A g i l i L cap d BCB embedded inductor w i d h 1 BCB BCB Low Res Si or Glass B h Figure 5-4. Multilayer CRLH unit cell on BCB. A) 3D schematic of conductors: note the ground and BCB layers are not seen for clarity. B) Cross section view of the via interconnection The meander inductor is embedded in BCB 5 m below the top layer, with a trace thickness of 2 m, which is close to 4 times the skin depth of copper at 14 GHz. The width of the implemented inductors is 20 m, which is selected in order to achieve high values of inductance. In addition, each turn has a turn spacing of 40m and variable lengths according to the required inductor value. The MIM capacitors are implemented with the bottom and top metallic layers. The interlayer distance d is 5 m. The width of the capacitors is selected to be 250 m and the length is variable according to the needed capacitance value. The theoretical capacitance is calculated based on the formula for the parallel plate capacitor. Further optimization processes are performed in order to achieve design specifications. Figure 5-5 shows the extracted parameters of the equivalent circuit of the MIM capacitors with variable length L Ca. The equivalent circuit presented in section has been used. Losses are not taken into account for the simple capacitor modeling, but they are considered in the complete model of the unit cell. Full wave structure simulation of the unit cell is performed in 3D high frequency structure simulator (HFSS, ANSYS Inc.). Figure 5-6A compares the electromagnetic and circuital simulation results of the frequency response for a balanced unit cell. Less than 1.27 db insertion 123

124 S 11 and S 21 (db) Frequency (GHz) Capacitance (pf) Inductance (nh) loss at 35 GHz and a return loss better than 15 db are expected, which are comparable with the loss values per unit cell obtained in other work [32]-[34], [89]. Figure 5-6B shows the simulated results for the propagation constant p. It is observed that there is a dual band operation at 14 GHz and 35 GHz where the phase constant of the unit cell is approximately 45 and -45, respectively, as designed L cp C s C cp C Th Lcap (um) Figure 5-5. Extracted parameters for the MIM capacitors on BCB. Extracted series capacitance C S (2C L ), parasitic capacitance C CP and calculated theoretical value C TH. The Extracted parasitic inductance L CP is also shown S 11 S 11 EM S 21 EM S 11 Circuit S 21 Circuit Frequency (GHz) A Propagation constant (deg) Figure 5-6. Simulated performance of the unit cell. A)Insertion and return loss. B) Phase constant. The extracted values are: 2C L = 0.33 pf, C CP = pf, L CP = 0.11 nh, L L = 0.43 nh, C LP = 0.01 pf, L R = nh B EM simulation Circuit simulation 124

125 Loss factor Loss Analysis In order to analyze the loss factor of different elements in the unit cell, simulations are performed in various conditions that all the parameters are assumed lossless except one parameter, whose loss contribution needs to be identified. Figure 5-7 shows the loss factor as a function of frequency. It is observed that the major loss is caused by the silicon wafer due to its low resistivity. Radiation loss is higher in the high frequency band where the size of the unit cell becomes comparable with a quarter guided wavelength. Figure 5-8A and Figure 5-8B show the current distribution at 14 GHz and 35 GHz, respectively, where it is concentrated on the lefthanded components, especially in the meander inductor. It is speculated that the left-handed components have a great contribution to the loss in the unit cell, which is not avoidable BCB loss Conductor loss Silicon loss Radiation loss Total loss Loss Factor=1- S S Frequency (GHz) Figure 5-7. Simulated loss factor of the unit cell for ideal dielectric, ideal conductor, lossless silicon, radiation loss and total loss. A Figure 5-8. Current distribution in the CRLH unit cell. A) Current distribution at 14 GHz. B) Current distribution at 35 GHz. 125 B

126 5.3. Fabrication In our study, a silicon wafer with a thickness of 280 m and a resistivity of 10.cm is used as the carrier substrate, which maintains compatibility with the standard CMOS/MEMS processes. Cyclotene from Dow Chemical, has been selected as the BCB dielectric interlayer due to its low loss properties (tan =0.002 at 10 GHz). Metallization is performed via DC sputtering of Ti/Cu seed layer and Copper electroplating. Figure 5-9 illustrates the fabrication procedure. First, substrate cleaning is performed with aggressive Piranha etching (H 2 SO 4 : H = 7:3 ratio), followed by rinse in deionized (DI) water and dehydration on a hot plate at 120 C for 15 minutes. Further, a layer of BCB with a thickness of 14 m is coated on the silicon substrate by following the recommended procedure by Dow Chemical [84], which can be found in the Appendix. Soft bake is performed on a hot plate at 100 C for 90 seconds. At the end of the soft baking process the samples are allowed to cool down to room temperature. Lithography with conventional UV light (i-line: 365 nm) is performed with the recommended values for exposure, developing and post-baking times. The samples are developed for 3 minutes in DS3000 developer at 35C, followed by a second developing step at room temperature to stop the developing process, and finally rinsed with DI water and blow dried with a nitrogen gun. Curing is performed in a vacuum oven at 150C for 15 minutes and 210C for 40 minutes. Descum is performed for 60 seconds using reactive ion etching (RIE) with an O 2 :SF 6 gas composition of 90:10 at 10mTorr. A metallic layer of Ti/Cu (30 nm/300 nm) is sputtered on top of the BCB layer followed by negative photoresist patterning with NR (Futurrex, Inc.) for the embedded inductor and the bottom conductor of the capacitor. Electroplating of a 2 m Cu layer and etching of the 126

127 Ti/Cu seed layer are performed. A thin 7 m layer of BCB is further coated to create the interlayer dielectric that forms the MIM capacitor. Soft bake, exposure, developing and curing are repeated as the Dow Chemical recommended procedure. LR Silicon LR Silicon Spin coating of BCB 14m of BCB layer on Silicon substrate. Cu DC sputtering of Ti seed layer. LR Silicon LR Silicon Cu electroplating of 2m layer and etching of seed layer Coating of the second layer of BCB and lithographically pattern of the via interconnection. LR Silicon NR9- Pattern of embedded conductor LR Silicon Sputtering of seed layer, pattern of top conductor and electroplating of 5m Cu layer. Figure 5-9. Fabrication process of the unit cell. BCB is used for achieving the interlayer distance d = 5m. Finally, a layer of Ti/Cu/Ti (30 nm /300 nm /30 nm) is sputtered, followed by negative resist NR-9800 coating to a thickness of 10 m, and the lithographical patterning of the top layer. After removing the upper Titanium layer, a 5 m thick Cu layer is electroplated to create the top metal trace and the capacitor top conductors. At the end, the seed layer of Ti/Cu is etched away. Figure 5-10 shows the SEM image of the fabricated CRLH unit cell, where the shadow of the embedded inductor is observed. The total area of the unit cell, including just the ground planes but not the launching feeding lines, is mm 0.67 mm, which gives at 14 GHz and at 35 GHz, confirming the compact fashion of the implementation. Figure 5-11 shows the microphotographs in color of the fabricated devices, where the embedded inductor can be appreciated. The unit cell, the two unit cell transmission line and the CPW feeding line used in the initial analysis are presented. 127

128 Figure SEM Image of the CRLH unit cell. A B Figure Photomicrographs of the microfabricated devices. A) Unit cell. B) CPW transmission line. C) Dual band transmission line. C 128

129 S 11 and S 21 (db) S 11 and S 21 (db) 5.4. Measurement Results Figure 5-12 shows the measured return and insertion loss of the 50 ohms unit cell. Conventional on-wafer SOLT (short, open, load and thru) calibration is used from 5 GHz to 40 GHz. Measurements are performed with a vector network analyzer Agilent E8361A and a Cascade Microtech probe station with a probe tip pitch of 150m. Measurement results show good agreement with simulated results. An insertion loss less than 2 db and a return loss better than 10 db are achieved within the passband of the CRLH unit cell. As observed, the fabricated unit cell is not completely balanced and the measurement results show additional loss, mainly due to the tolerance in the fabrication process, metal roughness, variation in the metal thickness, resistivity and loss tangent of the silicon, and variation in the loss tangent and dielectric constant of the BCB. The tolerance in the fabrication process and the resolution of the photomask, as well as mis-aligning between multiple layers, can cause variation in the width of the inductors and capacitors, which results in the degradation of the performance. However, the results are very promising and a much more optimized fabrication can alleviate the fabrication errors and improve the performance S 11 Meas S 21 Meas -30 S 11 EM sim S 21 EM sim Frequency (GHz) S Frequency (GHz) A B Figure Measured performance. A) Unit cell. B) Two unit cell dual band CRLH transmission line S

130 Phase angle (deg) Figure 5-12B shows the measured performance of the two unit cell transmission line, which features a maximum insertion loss of 6 db at 14 GHz and 5 db at 35 GHz. The implementation shows higher losses than expected that can be attributed to tolerances in the dimensions of the capacitors and inductors, as well as the CPW gap, resulting in a slightly unbalanced frequency response as observed in Figure 5-12B. However, the fabrication process can be optimized in order to reduce loss Measured Simulated Frequency (GHz) Figure Measured phase constant of the two unit cells transmission line showing a dual band behavior around 12.8 GHz and 36 GHz. Figure 5-13 shows the measured phase angle of the two unit cell transmission line. A slight difference in the low frequency is observed, in which the /4 behavior is presented around 12.8 GHz instead of 14 GHz. These results are attributed to the differences in the final dimensions of the capacitors, inductors and the CPW gap during the fabrication process. However, an optimized fabrication process would alleviate these differences. Nevertheless, the measured results are in good agreement with the expected results, demonstrating that an optimized multilayer process by using low resistivity silicon and BCB interlayer is suitable for 130

131 the implementation of compact multilayer CRLH devices. Additional steps can be undertaken in order to minimize loss, such as a thicker BCB interlayer that reduces even further the interaction of the electromagnetic field with the silicon substrate, a thinner inter-capacitor BCB layer (which is currently 5 m) to reduce the size of the MIM capacitors, conductor backed coplanar (CBCPW) launching feeding lines (although not preferred), or micromachined CPW gap by using reactive ion etching as featured in [35] Summary This section has presented the design, simulation and fabrication of super compact multilayer micromachined CRLH transmission lines by using BCB as a dielectric interface layer on a CMOS grade low resistivity silicon substrate. The selection of BCB is based on its mechanical and electrical properties. Circuital and electromagnetic simulations show good agreement with preliminary measured data. The insertion loss seems to be still high mainly due to cross talk with the lossy silicon substrate as identified from the loss analysis, while the usage of a thicker BCB layer is expected to reduce the loss factor significantly. The surface micromachined process on BCB has demonstrated to be useful for achieving compact circuits at microwave and millimeter wave frequencies. A compact dual band unit cell and a two unit cell CRLH transmission line have been designed, fabricated and tested. The fabrication process is being optimized in order to reduce differences between the simulated and measured results. This process can be extended to the fabrication of a great variety of metamaterial circuits. 131

132 CHAPTER 6 METAMATERIAL LOADED THE HALF MODE SUBSTRATE INTEGRATED WAVEGUIDE High performance microwave bandpass filters with low insertion loss, high selectivity, compact size and multiple bands are widely used for wireless and satellite communication systems [90]. Conventionally, high performance filters are implemented with the conventional bulky waveguide technology, which, however, is not readily compatible with other planar or multilayer integrated circuits technologies such as printed circuit board (PCB), monolithic microwave integrated circuits (MMIC) and complementary metal oxide semiconductor (CMOS) processes. In order to reduce the size and improve the performance, during the years, bandpass waveguide filters have used all kinds of metallic and non-metallic insertions [91]. Since the first experimental demonstrations of metamaterial particles exhibiting either negative permeability such as the split ring resonators (SRR), or negative permittivity such as the complementary split ring resonators (CSRR), different implementations that combine the rectangular bulky waveguide with such structures [92-93] have also been investigated. Such combination has been mainly motivated by their extraordinary property of generating backward wave transmission below the waveguide cutoff frequency. However, modern technologies demand more and more System on Package (SoP) or System on Substrate (SoS) approaches to achieve the compactness of the devices and systems. Planar Microstrip and Coplanar Waveguide bandpass filters are good alternatives for hybrid or on-chip implementations, while their performance is usually inferior to that of the bulky waveguide filters in terms of radiation leakage and Q factor [90, 94]. The need for new applications and integration with digital circuitry is the motivation for proposing and implementing planar microwave filters with performances similar to those provided by the bulky waveguide filters. Recently, the substrate integrated waveguide (SIW) and the half mode substrate integrated waveguide concepts (HMSIW) [41-42] have demonstrated to 132

133 be key wave guiding structures for the implementation of low loss, high quality factor and improved selectivity waveguide bandpass filters on the printed circuit board (PCB) technology. In addition, taking into account the possibility of having forward wave propagation below the waveguide cutoff frequency, the substrate integrated waveguide has been combined with complementary split ring resonators (CSRR) for the implementation of compact size and high selectivity bandpass filters [45]. In this chapter, the half mode substrate integrated waveguide (HMSIW) concept and the complementary split ring resonators (CSRR) are used to implement miniaturized bandpass filters working with forward wave propagation below the waveguide cutoff frequency. The half mode substrate integrated waveguide allows additional size reduction compared to that provided by the substrate integrated waveguide (SIW) and more flexibility in the control of the quality factor, since no complicated microstrip to HMSIW transition is needed. At first, a single band bandpass filter working at 5.25 GHz and a dual band filter working at 3.4 GHz and 5.85 GHz are presented. The filters make use of CSRR loaded HMSIW resonators working under the principle of evanescent wave amplification. The second part shows how an electrically tunable resonator can be implemented by using the proposed CSRR loaded HMSIW resonator. The third part presents how the proposed evanescent mode CSRR loaded HMSIW resonator can be used for wireless sensing purposes. Finally, the design of dual band filters by using CSRR loaded HMSIW resonators is presented Single and Dual Band Bandpass Filters Using CSRR Loaded Half Mode Substrate Integrated Waveguide Theoretical Backround Before studying the proposed resonators, the technology used to implement them is studied. For that purpose, the conventional rectangular waveguide is first analyzed. The equations that 133

134 define the propagation of its dominant TE 10 mode are visited. In the same way, in order to illustrate its working principle, the simulation of the electric field distribution of the dominant TE 10 mode is provided. In addition, the planar version of the rectangular waveguide, the Substrate Integrated Waveguide (SIW) and its reduced mode version, the Half Mode Substrate Integrated Waveguide (HMSIW) are also studied. The geometries, design equations and the simulation of the distribution of the electric field for the propagation of the quasi-te 10 mode in both structures are provided. At the end, the proposed evanescent mode HMSIW resonators are introduced. The single band and dual band resonators are analyzed in detail, emphasizing on the working principle, the electrical equivalent circuit and the parameters to be used for bandpass filter design Rectangular waveguide In Figure 6-1 the geometry of the conventional rectangular waveguide is presented, where it is assumed a material with an electric permittivity and a magnetic permeability is filling the waveguide. In a conventional waveguide the side a is longer than the side b (a > b) [57]. It is not the aim of this section to give a complete mathematical treatment of the rectangular waveguide, but to offer a highlight of the equations that define the propagation of the dominant transverse TE 10 mode. Propagating transverse modes in the waveguide have null longitudinal component of the electric field, E z = 0 [57]. For the case of the TE 10 propagating mode, the field components can be reduced to [57] H z A x cos e a jz 10, (6-1) E y ja A x sin e a jz 10, (6-2) 134

135 H x ja A x sin e a jz 10, (6-3) and E x = E z = H y = 0, (6-4) where A 10 is the amplitude. Additionally, the TE 10 mode waveguide cutoff frequency, propagation constant and phase constant in the medium are defined by f 1 C 10 2a, (6-5) 2 k 2 a, (6-6) k. (6-7) y Metallic surface b, a Metallic wall x z Figure 6-1. Geometry of the rectangular waveguide. The effective dielectric constant of a rectangular waveguide is obtained as in 2 f 10 1 c eff r, (6-8) f from which is observed the rectangular waveguide presents negative effective electric permittivity below the waveguide cutoff frequency. Equation 6-5 shows the high pass behavior 135

136 of the rectangular waveguide. In order to illustrate the propagation of the TE 10 mode in a rectangular waveguide, Figure 6-2 shows the simulated distribution of the electric field. It is observed that the field is concentrated in the center of the waveguide. y z x Figure 6-2. Simulated electric field distribution for the TE 10 mode in a rectangular waveguide The substrate integrated waveguide Although the rectangular waveguide is a useful guiding structure for filter design [91], due to its bulky configuration the integration with conventional planar technologies is challenging. To overcome this issue, the substrate integrated waveguide (SIW) concept was introduced in [41], which preserves the wave guiding properties of the conventional rectangular waveguide and allows the implementation of high performance bandpass filters and components using the wellknown standard rectangular waveguide techniques. Figure 6-3A shows the SIW basic geometry, where the waveguide is implemented in a substrate of height h and relative dielectric constant r, as shown in Figure 6-3B. Its simplest configuration consists of feeding microstrip lines, tapered transitions at both ends, and the waveguide section of width W and length p. A row of metalized via-holes replace the metallic sidewalls of the waveguide. The transitions are necessary to convert the propagating quasi-tem mode of the microstrip line into a propagating quasi-te 10 mode of the SIW. 136

137 Microstrip Line D s Transition Metallic via wall Top metal W r Ground plane h p A B Figure 6-3. The substrate integrated waveguide. A) Top view with geometrical parameters. B) Cross section view depicting the metallic via walls. The SIW behaves as a rectangular waveguide if the separation of the metalized via-holes is much smaller than the guided wavelength, so the via wall can be considered a metallic wall and the radiation loss is minimized [41]. For design purposes [41] : and D g 5, s 2D. (6-9) (6-10) Equations 6-9 and 6-10 offer good design guidelines, but they are not considered to be completely necessary. On the other hand, the TE 10 cutoff frequency can be approximated by : f c10 c, 2 W r eff _ SIW (6-11) which is similar to equation (6-5) taking into consideration an effective width W eff_siw for the SIW, given as in [95] W 2 2 D D 2W (6-12) s W eff _ SIW 2 Before illustrating the propagation of the quasi-te 10 mode in the SIW, the next section studies its reduced mode version, the Half Mode Substrate Integrated Waveguide (HMSIW) [42]. 137

138 The half mode substrate integrated waveguide A different version of the in-substrate rectangular waveguide uses only half of the conventional SIW [42], the so called half mode substrate integrated waveguide (HMSIW). Its configuration is illustrated in Figure 6-4. Its working principle is derived from Equations 6-1 to 6-4, where it is observed that the tangential component of the magnetic field along the direction of propagation H z is zero at the center of the waveguide ( x = a/2 ), hence, if the SIW is cut into half along this direction (E-plane), the open end will behave as a perfect magnetic wall under the condition that the width a is much larger than the thickness of the dielectric h ( a >>h ) [42, 95]. Because of the magnetic wall, the propagating mode in the HMSIW keeps half of the field distribution of the quasi-te 10 mode in the conventional SIW, therefore the TE dominant propagation mode can be denominated a quasi-te 0.5,0 mode [95]. Since the propagation characteristics of the HMSIW are similar to those of the SIW, its design equations can be derived from those of the SIW. The cutoff frequency of the fundamental quasi-te 0.5,0 mode of the HMSIW is calculated as in [95] f cte 0.5,0 c, (6-13) 4 W r eff _ HMSIW where the W eff_hmsiw is the effective width of the HMSIW, calculated by the empirical formulas [95]: and W eff _ HMSIW Weff _ SIW / 2 W, (6-14) W h 0.3 W.05 ln 0.79 r 104W '2 ' eff _ HMSIW eff _ HMSIW h h , (6-15) h ' where W W / 2. eff _ HMSIW eff _ SIW 138

139 Microstrip Line W HMSIW s A p D Transition Metallic via wall Top metal Ground plane Figure 6-4. The half mode substrate integrated waveguide (HMSIW). A) Top view with geometrical parameters. B) Cross section view depicting the metallic via wall. r B h The advantages of using the HMSIW is that the size is reduced almost 50% in comparison with the original SIW. However, due to its reduced size, the unloaded quality factor is also reduced to almost half of that of the SIW [41]. To illustrate the wave propagation of both SIW and HMSIW, Figure 6-5 shows the electric field distribution of the dominant quasi-te modes on these two structures. It is observed that the HMSIW propagates a mode that resembles the half portion of the TE mode in the SIW, hence, the propagation characteristics of the SIW are preserved by the HMSIW. In addition, Figure 6-6 shows a typical frequency response for an SIW or an HMSIW designed for an 8.2 GHz quasi-te 10 mode cutoff frequency. It is observed that the frequency response is also similar to that of the dominant TE 10 mode in a conventional rectangular waveguide [57]. Therefore, the propagation characteristics of the SIW and HMSIW make them useful for implementing microwave and millimeter wave applications with similar performance to those of their rectangular waveguide counterparts [41, 42]. Since the performance and geometry of the SIW and the HMSIW are similar to that of the dielectric filled rectangular waveguide, the well-known rectangular waveguide circuits such as filters, power dividers and couplers can be implemented with similar design procedures [41,42]. The next section discusses HMSIW evanescent mode resonators proposed in our work. 139

140 S 11, S 21 (db) y z x A B Figure 6-5. Electric field distribution of the dominant TE mode. A) The substrate integrated waveguide (SIW). B) The half mode substrate integrated waveguide (HMSIW) S S Frequency (GHz) Figure 6-6. A typical frequency response for the SIW and HMSIW structures. The cutoff frequency is 8.2 GHz. 140

141 Proposed CSRR Loaded Half Mode Substrate Integrated Waveguide Evanescent Mode Resonators The proposed CSRR loaded HMSIW resonators for single and dual band operation are presented in Figure 6-7 [46]. All designs are implemented on the substrate Arlon Diclad 880 with a thickness of mm, a dielectric constant of 2.2 and a loss tangent of Each resonator has a row of metalized vias with a diameter d of 0.7 mm and a separation from center to center s of 1.4 mm are used to provide the electric sidewall of the waveguide. The waveguide cutoff frequency is selected to be 8.2 GHz, which is achieved with an optimized width w of 6 mm for HMSIW implementations. The complementary split ring resonator is etched on the metallic surface of the waveguide. A 50 microstrip line with a width w 1 of mm is connected directly to the waveguide with no transition. A Figure 6-7. Proposed (HMSIW) resonator with a series of vias for electric walls and complementary split ring resonator (CSRR) on the top surface. A) Single band. B) Dual band Working principle The working principle is based on the epsilon negative behavior of a rectangular waveguide under the cutoff frequency, whose effective dielectric constant, eff, for the propagating TE 10 mode in a substrate with relative permittivity r, was previously given by Equation 6-8. The complementary split ring resonator is a resonant metamaterial particle that B 141

142 S 11, S 21 (db) provides a negative permittivity at its particular resonance frequency [93]. When the CSRR is loaded on the surface of a waveguide, the interaction with the HMSIW structure causes the resonance frequency to be different to that of the original CSRR and generates a positive effective dielectric constant eff, which provides a frequency band with forward wave propagation below the waveguide cutoff frequency [45]. In order to further understand its working principle, at first a conventional complementary split ring resonator is simulated and its electrical permittivity is retrieved by using the method described in [96]. It is not the aim of our study to show the complete retrieval procedure, but to proof the metamaterial behavior. The frequency response of the conventional CSRR is shown in Figure 6-8A, where it is observed that the CSRR presents a bandstop behavior at 6.1 GHz. In addition, Figure 6-8B confirms the negative electrical permittivity of the CSRR at its resonance frequency S 11 S Frequency (GHz) A Frequency (GHz) B Figure 6-8. Simulated results for a conventional split ring resonator. A) Frequency response. B) extracted electrical permittivity and magnetic permeability. Geometrical parameters are a = b = g =0.25 mm, c 1 =c 2 = 3.85 mm, p = 0.77 mm, l = 1.4mm. Next, the simulated results of the CSRR-loaded HMSIW resonator for single band operation at 5.25 GHz are presented in Figure 6-9. All simulations have been performed by using 142

143 Ansys High Frequency Structure Simulator (HFSS). Figure 6-9A shows that the behavior of the HMSIW resonator is bandpass, in contrast to the bandstop behavior of the original CSRR. Moreover, due to the interaction with the waveguide, the resonance frequency the HMSIW is shifted up. On the other hand, the effective permittivity of the HMSIW resonator has been retrieved from the S parameters and it is presented in Figure 6-9B. As observed, the effective permittivity is positive at the resonance frequency, which indicates that a frequency band with forward wave propagation is created below the original waveguide cutoff frequency. To further confirm the forward wave propagation behavior, the propagation constant p of the HMSIW resonator is shown in Figure 6-9C. A passband with a positive slope is obtained around 5.25 GHz, which confirms the forward propagation behavior. Moreover, the attenuation constant, presented in the same plot, shows a near-zero value in the passband, indicating a low attenuation to the propagating wave. It is also confirm that the waveguide cutoff frequency is shifted up to a higher value. In addition Figure 6-9D shows the electric field distribution at resonance, indicating that the propagating mode is mainly concentrated in the CSRR and is completely different from the original quasi-te 0.5,0 dominant mode of the HMSIW. This also is an indication that a tapered microstrip to waveguide transition is not strictly necessary. The same analysis can also be done for the dual band resonator of Figure 6-7B with similar results. In this case, the inner ring of the CSRR uses a meander line configuration in order to lower the second intrinsic resonance frequency and achieve a double forward wave propagation below the characteristic waveguide cutoff frequency. Next section studies the resonator characterization for bandpass filter design. 143

144 S 11, S 21 (db) S 11 S Frequency (GHz) A Frequency (GHz) B Frequency (GHz) C Figure 6-9. Simulated results for a CSRR loaded HMSIW resonator. A) Frequency response. B) Extracted electrical permittivity and magnetic permeability. C) Simulated dispersion and attenuation diagram. D) Electric field distribution at resonance. Geometrical parameters are a = b = g =0.25 mm, c 1 =c 2 = 3.85 mm, p = 0.77 mm, l = 1.4 mm, w = 6 mm, t = 0 mm Bandpass filter design The classic methodology for the design of coupled resonator banpdasss filters is followed [90]. For bandpass filter design, the loaded or external quality factor Q e of the resonator and the internal coupling coefficients M ij between resonators play important roles in the design procedure [90]. According to the filter design specifications, the circuit for a low pass prototype of the filter is first synthesized. After the low pass filter design, the required external quality factor and internal coupling coefficients are determined. The next step is to match the physical D 144

145 design with the synthesized circuit design. This section briefly discusses the basic design guidelines without giving a complete treatment of the filter theory available in [90]. Resonator characterization : In the proposed resonators, the position t of the microstrip feeding line and the offset location l of the complementary split ring resonator in the unit cell control the external quality factor (Q e ). For simplicity, all designs use t = 0. On the other hand, the distance p from the top of the waveguide also affects the external quality factor of the resonator, therefore, it is selected to be fixed in the design. Thus, the external quality factor is entirely controlled by the offset distance l. Out of four possible CSRR orientations in the waveguide, the proposed configurations in Figure 6-7 show the best transmission responses since the electric field in the HMSIW structure is mainly concentrated at the center. In Figure 6-7B, the inner ring of the CSRR is modified with a meander line structure in order to reduce the second intrinsic resonance frequency of the CSRR and achieve dual band operation below the waveguide cutoff frequency. Thus, the resonance frequencies can be arbitrarily selected by modifying the dimensions of the CSRR. Selected resonance frequencies and the dimensions of the unit cells are summarized in Table 6-1. Full wave structure simulations are performed using High Frequency Structure Simulator (HFSS, Ansys Inc.). Table 6-1. Dimensions of the proposed HMSIW-CSRR resonators Resonator g=a=b=c c 1 c 2 p Single band at 5.25 GHz 0.25mm 3.85mm 3.85mm 0.77mm Dual band at 3.5 and 5.85 GHz 0.25mm 6mm 4.5mm 0.57mm Initially, the external quality factors (Q e ) of the single and dual band resonators are obtained. For that purpose, a double loaded resonator is used in order to get the frequency response in terms of the S parameters. Then, the external quality factor is obtained as in Q 2 f o e, (6-16) BW3dB 145

146 where f o is the resonance frequency and BW 3dB is the 3dB bandwidth for the response of S 21, as illustrated in Figure 6-10A. Figure 6-10B shows the range of Q e for the single band resonator when varying the offset distance l. Figure 6-10C shows the external quality factor of the dual band resonator at both frequency bands. It is observed that the Q e of the first band achieve higher values than those of the second band, mainly due to weaker second order resonance of the CSRR. S 21 BW 3dB f L f o f H A f External Quality Factor Q e l (mm) B Lower Band Higer Band Q e l (mm) Figure External quality factor Q e of the CSRR loaded HMSIW resonator. A) Illustration of the method used to obtain Q e for a doubly loaded resonator. B) Extracted Q e from simulations for the single band resonator. C) Extracted quality factors for the dual band resonator at both bands. 146 C

147 Internal coupling coefficient : When two resonators or stages are close each other, the electromagnetic coupling between them causes the original resonance frequency of the resonator to be split into two different resonance modes, as illustrated in Figure Conventionally, the inter-resonator coupling coefficient between the stage i and the stage j, M ij is extracted from those two new modes as in M ij f f, (6-17) f f where f 1 and f 2 are the resonance frequencies of the low and high new generated modes, respectively. Generally, the sign of the coupling coefficient does not affect the design procedure, while it is only important when designing cross-coupled bandpass filters [90]. Different methods can be use to get the internal coupling coefficient [45, 90]. Figure 6-11 shows one way to extract the internal coupling coefficient, in which the resonators are fed with a weak external coupling which causes a high external quality factor [90]. For this case, a weak capacitive external coupling is used for that purpose, while the internal coupling is controlled by the inter-resonator distance l r Figure 6-12A and Figure 6-12B show the extracted internal coupling coefficient of the two coupled resonator for single and dual band cases. f 1 f 2 f Weak external coupling L I S 21 A B Figure Internal coupling coefficient. A) Split resonance frequency. B) Extraction of the coupling coefficient. 147

148 Internal Coupling Coefficient M Internal coupling coefficient (M) First Band Second band L IR (mm) A Figure Extracted internal coupling coefficients. A) Single band case. B) Dual band case Two Pole Filter Implementation and Measurement Results L IR (mm) In order to demonstrate the design concept of HMSIW filters, a set of two pole Chevishev filters have been designed by using the coupled resonator design procedure [90]. A two pole single band Chebyshev filter for 5.25 GHz with a fractional bandwidth of 3.8% and a 0.02dB passband ripple, and a two pole dual band Chebyshev filter for 3.5GHz and 5.85 GHz with a 20dB return loss and a fractional bandwidth of 5.7% and 7.5%, respectively, are designed and implemented. Table 6-2 summarizes the calculated parameters and optimized dimensions. B A Figure Two pole bandpass filters. A) Single band. B) Dual band. Table 6-2. Calculated parameters and dimensions of the two pole HMSIW-CSRR filters Filter Q M l l r Size Single band at 5.25GHz mm 9 mm g g Dual band at 3.5GHz Dual band at 5.85GHz mm 8.6 mm g g B 148

149 S11 and S21 (db) S11 and S21 (db) S21 and S11 (db) S11 and S21 (db) Figure 6-14 shows the measurement and simulation results of the fabricated filters. The fabricated HMSIW-CSRR unit cells and filters are shown in Figure The conventional printed circuit board (PCB) fabrication process with a CNC milling machine has been used. Measurements are performed using an HP8510C vector network analyzer after standard shortopen-load-through (SOLT) calibration in the frequency range of 2 GHz to 10GHz. Measurement results agree well with those of simulations. A maximum insertion loss of 2dB is obtained for the single band filter. The effect of the connectors and feeding lines has not been extracted, which indicates the total loss of the filter might be less. Minute differences can be attributed to the tolerances in the fabrication process Frequency (GHz) S 21 S 11 S 21 S Frequency (GHz) C A Simulation Measurement Sim. Meas Frequency (GHz) S 21 S 21 Simulation Measurement S 11 B S 11 Simulation Measurement Frequency (GHz) D Figure Measurement and simulation results. A) Single band resonator. B) Single band two pole filter. C) Dual band resonator. D) Dual band two pole filter. 149

150 A B Figure Fabricated resonators and filters. A) Single band. B) Dual band Summary Single and dual band resonators are implemented using the half mode substrate integrated waveguides (HMSIWs) loaded with complementary split ring resonators (CSRR). Forward wave propagation is achieved below the characteristic cutoff frequency of the waveguide due to the evanescent wave transmission properties of the CSRR. Since no transition is needed, a very compact size is achieved. Single and dual band two pole filters are implemented by using the theory of coupled resonator filters. Full wave simulations are in good agreement with measurements. Minute differences are observed and mainly due to fabrication tolerance and additional loss due to the connectors. For comparison, the available external quality factors for the CSRR loaded HMSIW resonators in this work are lower than those offered by the CSRR loaded SIW resonators of reference [45], mainly due to the reduced size of the HMSIW structure. However, some advantages of using the HMSIW are that the size is reduced and the external quality factor can be easily controlled by three different geometrical parameters. In addition, since the HMSIW has an open side, tunable applications by loading varactor diodes are easy to implement. 150

151 6.2. Electrically Tunable Evanescent Mode Half Mode Substrate Integrated Waveguide Resonators This section explains the implementation of electrically tunable evanescent mode half mode substrate integrated waveguide (HMSIW) resonators for S band applications [97]. The previously studied CSRR loaded HMSIW resonator, which achieves forward electromagnetic wave transmission below the characteristic waveguide cutoff frequency due to evanescent wave amplification, is additionally loaded with a variable capacitor connected to one of the conductors of the CSRR. The capacitive loading changes the effective capacitance to ground, resulting in frequency tuning of the resonator. Three different configurations are investigated with a varactor diode connected between the ground and three different contact points of the CSRR. The external Q factor is slightly affected by the frequency tuning. Two electrical equivalent circuits, representing the three different cases, are used to model the behavior of the tunable resonators. More than 15% tunability is achieved around 3.4 GHz. Full wave structure simulation results are in good agreement with those of measurement The Tunable CSRR loaded HMSIW Resonator The single band evanescent mode HMSIW resonator introduced in section is recalled in this section and shown in Figure 6-16A. The width of the HMSIW, w, controls the waveguide cutoff frequency, while the resonance frequency is mainly controlled by the dimensions of the CSRR. Figure 6-17B shows the electric field distribution at resonance, which is mainly concentrated on the conductors of the CSRR with the higher concentration in the inner metal patch, suggesting three different tuning configurations by connecting an external variable capacitor to three different locations of the CSRR. 151

152 S21 S11 (db) Max Min A B S11 Sim S21 Sim -40 S11 Meas S21 Meas S11 HMSIW S21 HMSIW Frequency GHz 7 8 C Figure The electrically tunable resonator. A) Previously proposed CSRR-loaded HMSIW resonator. B) Electric field distribution at resonance. C) Simulated and measured frequency response of the CSRR loaded HMSIW resonator and a conventional HMSIW structure. D) Layout of the proposed electrically tunable resonator, where connection points A, B and C offer three different resonators (A, B and C). The geometrical parameters are h = 0.5mm, g = a = b = 0.25mm, w = 6.4mm, l x = l y = 4.8mm, l =1.2mm, w L = 1.55mm. All resonators are implemented on a substrate Arlon DiClad 880 with a thickness of 508 m and a dielectric constant of 2.2. The via diameter, d, is 0.75 mm with a pitch, s, of 1.4 mm. The width of the HMSIW, w, is 6.4 mm for a waveguide cutoff frequency of 7.1 GHz, as shown in Figure 6-16C. A 50 microstrip line is directly connected. The external quality factor is controlled by the offset distance l.figure 6-16C shows the measured and simulated results of the original CSRR loaded HMSIW resonator. The resonance frequency below the waveguide cutoff 152 D

153 frequency is observed at 3.6 GHz. The simulated and measured external Q factors are 19 and 23, respectively. As shown Figure 6-16B, the electric field at resonance is not uniformly distributed on the CSRR, therefore the electric coupling with the variable capacitance will depend on the location of the connection. In Figure 6-16D three different tunable configurations are possible. The variable capacitance, implemented by a varactor diode in series with a decoupling capacitance (C DC ), can be connected to three different points: The open sidewall of the waveguide in point A (the outer conductor of the CSRR, which has the weakest electric field at resonance frequency), the inner strip of the CSRR in point B (the conductor between the split rings, on which the electric field is slightly stronger), or the inner conductor of the CSRR in point C (with the strongest electric field, and therefore it is supposed to offer the largest shift in the resonance frequency with a given capacitance value). Table 6-3 shows the parameters of the varactor diode (SMV1231, Skyworks TM Inc.). Table 6-3. Parameters of the varactor diode Model SMV 1231 Voltage (V) Capacitance (pf) V R = 15 V R s = L s = 0.7 nh Figure 6-17C shows the simulated results of the shift of the resonance frequency for the three resonators and the electrical equivalent circuit in Figure 6-17A and Figure 6-17B when the capacitance of the diode is pf. The bias circuit and the diode model are considered in the simulations. The resonator C presents the largest frequency shift. Resonators A and B have slightly different frequency shifts and a dual band frequency behavior, on which the upper resonance is produced by the connection of the variable capacitance to the waveguide structure, implying a different electrical equivalent circuit compared to that of the resonator C, as depicted in Figure 6-17A. The electrical equivalent circuit of the resonator C, in Figure 6-17B, is similar 153

154 S 11, S 21 (db) to that used in [45], but for this case both the coupling capacitance C C and the CSRR capacitance C R are affected by the variable capacitance. In each case, the inductor L V represents the inductance of the metalized via row, the inductor L c and capacitor C c represent the inductive and capacitive coupling of the HMSIW with the CSRR, and the CSRR is represented by the inductor L r and the capacitor C r. A B S 11 No varactor -30 S 21 Resonator A Resonator B Resonator C S 21 Circuit Sim Frequency (GHz) C Figure Simulated results. A) Electrical equivalent circuit for resonators A and B. B) Electrical equivalent circuit for resonator C. Extracted parameters are C c = 0.6 pf, L c =1.07 nh, L v = 0.62 nh, C r = 3 pf, L r = 2 nh, L d = 1 nh. C T models the variable capacitance. C) Simulation of the resonators with different tuning configurations and the same capacitance value. C T = pf. 154

155 Implementation and Measurement Results The three tunable configurations are implemented and tested. Additional capacitors and RF choke inductors (Taiyo Yuden Inc.) are used to implement the bias circuit for the varactor diode. Three non-zero voltages of 2.5 V, 5 V and 15 V are applied to the diode and the frequency shifts measured. Measurements have been performed with a vector network analyzer HP8719D. Figure 6-18 shows the photograph of the fabricated tunable resonators A and C. Figure Photograph of the fabricated tunable resonators. Resonator A (left) and Resonator C (right). Figure 6-19 shows the measurement results for resonators A and B. For the selected capacitance values, tunable ranges of 17% from 3.36 GHz to 2.78 GHz and 22% from 3.21 GHz to 2.48 GHz are achieved for the resonators A, B, respectively. The external quality factor Q e is modified to lower values between 15 and 17 during the tuning process. Less than 2.18 db and 2.6 db insertion loss is observed within the tunable range for the resonators A and B, respectively, which can be attributed to the loss in the additional lumped elements and the connectors. As observed, the tunable resonators A and B present a degraded dual band behavior, 155

156 S 11 (db) S 21 (db) mainly due to the additional capacitive loading of the HMSIW provided by the varactor diode. Therefore, the electrical equivalent circuit shown in Figure 6-17A is a representation of the dual band behavior of these two resonators, in which the varactor diode is connected as an independent branch represented by the capacitance C T and the inductor L d No varactor 15V 5V 2.5V Frequency (GHz) A V Frequency (GHz) B 156 No Varactor Figure Measured results with applied DC voltage. A) and B) Return and insertion losses for resonators A (solid lines with markers) and B (dashed lines). 15V 5V

157 S 11, S 21 (db) Meanwhile, Figure 6-20 shows the measured results for the resonator C, featuring the largest resonance frequency shift from the primary resonance frequency of the original resonator, deep inside the evanescent region of the waveguide, which causes lower return losses at resonance and a less than 4 db insertion loss within the tunable range. As observed, the resonator C keeps the single band frequency response as the original CSRR loaded HMSIW resonator, which features the forward wave propagation band below the waveguide cutoff frequency and the degraded upper waveguide passband of the original HMSIW structure. Then, the electrical equivalent circuit previously introduced in Figure 6-17B is a representation of the single band behavior of the resonator C, in which the tuning effect of the diode affects not only the original resonance frequency, but also the out-off-band transmission zero and the degraded waveguide passband. A tunable range of approximately 14% has been obtained from around 2.7 GHz down to 2.3 GHz. Table 6-4 summarizes the measured results for the three implemented resonators. The size of the resonators, including the bias circuit, is also given V S 21 5V 2.5V Frequency (GHz) Figure Measured results with applied DC voltage for the resonator C. 157

158 Table 6-4. Summary of the measured results for the three implemented resonators Resonator Original f o (GHz) f o at 15 VDC f o at 5 VDC f o at 2.5 VDC Tunability A % B % C % Summary Three configurations of electrically tunable evanescent mode resonators implemented on the CSRR loaded HMSIW have been studied. The CSRR etched on the top surface of the HMSIW allows forward transmission below the waveguide cutoff frequency. Moreover, by connecting a varactor diode to the CSRR, the resonance frequency can be tuned when different DC voltages are applied. Measurements results are provided around 3.4GHz. A tunability of more than 14% for a variation of 12.5 V DC has been achieved. Additional insertion loss can be minimized by selecting low loss lumped components. The resonators can provide an easy way to implement planar tunable bandpass filters. Size at f o = 3.6 GHz 6.3. Dual-Band Filters Using CSRR and Capacitive Loaded Half-Mode Substrate- Integrated-Waveguide In the previous section, the CSRR loaded HMSIW resonator working below the g g waveguide cutoff frequency was used to implement a set of single and dual band two pole filters and an electrically tunable resonator. The approach demonstrated to be useful for the design of miniaturized, low cost bandpass filters. However, for dual band filters the independent control of the external quality factors and inter-resonator mutual coupling coefficients at both bands is challenging with the previous work. In this section, a different approach for implementing compact, planar dual band HMSIW filters is investigated. By loading the HMSIW with both a CSRR on the top surface and a capacitive metal patch, two independent frequency bands are generated. Moreover, the external Q factor and mutual coupling coefficient for both bands 158

159 present relatively independent control. The dual band resonator offers a compact size, low loss, and good selectivity, which makes it useful for filter applications. A dual band resonator and a two-pole miniaturized bandpass filter with a size of are demonstrated on the conventional printed circuit board (PCB) technology. The external and internal mutual coupling variations are fully investigated. Full wave structure simulation and measurement results are provided The Dual Band CSRR and Capacitive Loaded HMSIW Resonator Figure 6-21A illustrates the previously introduced single band CSRR loaded HMSIW resonator. The proposed dual band resonator is introduced in Figure 6-21B. Two loading structures, a CSRR and a capacitive metal patch, are integrated in the waveguide for providing two relatively independent frequency bands below the waveguide cutoff frequency. A row of metalized vias with a diameter d of 0.75 mm and a pitch s of 1.4 mm are used to realize the electrical sidewall of the HMSIW. The external Q factor of the lower frequency band is controlled by adjusting the input distance l, while the combination of the distance l and the width w s in the metal patch offers the controllability of the second band. The substrate Arlon Diclad 880 ( r =2.2 and a thickness of mm) is used for the implementation. The symmetric electrical equivalent circuit of the dual band resonator is introduced in Figure 6-21C. The resonant tank formed by L R and C R models the CSRR, while the capacitor C 2 models the capacitive coupling of the HMSIW with the CSRR [45]. The effect of the capacitive patch is modeled as a series reactance formed by L p and C p. The inductances L 1 and L 2 model the inductive contribution of the HMSIW, while the inductance L v models the inductive effect of the via wall. In order to validate the electrical equivalent circuit, Figure 6-22A shows the comparison of the electromagnetic and circuit simulation results. It is clearly observed that two independent 159

160 resonances are generated at 3.5GHz with an external Q factor of 20.5 and at 5.8GHz with an external Q factor of A B L 1 L 1 Port 1 L P Port 2 C p L V C 2 C 2 L V L 2 C R L R L 2 C Figure Dual band resonator. A) Previously proposed single band resonator. B) Proposed dual band resonator. C) Electrical equivalent circuit of the dual band resonator. To further understand the working principle, Figure 6-22B and Figure 6-22C show the electric field distribution at 3.5 GHz and 5.8 GHz, respectively, in which it is observed that the CSRR has its major contribution to the first resonance frequency with the field concentrated around the ring, while its contribution to the second resonance frequency is mainly due to the inductive and capacitive couplings with the HMSIW structure. In the same way, it is observed that the capacitive patch has a major effect in the second resonance frequency. 160

161 S 11, S 21 (db) S 11 HFSS sim. S 21 HFSS sim. -50 S 11 Circuit sim. S 21 Circuit sim Frequency (GHz) A B C Figure Simulated performance of the dual band resonator. A) Frequency response. B) Electric field distribution at 3.5 GHz. B) Electric field distribution at 5.8 GHz. Design parameters are: a = b = c = 0.25 mm, c 1 = c 2 = 4.5 mm, w = 6 mm, L C = 2.8 mm, W C = 5.5 mm, g = 0.25 mm, W s = 0.6 mm, l = 1.7 mm, t = 0 mm, p = 0.3 mm, w 1 = 1.55 mm. Extracted values are L v = 0.6 nh, L 1 = 1.07 nh, L 2 = 0.88 nh, C 2 = 0.53 pf, L P = 0.39 nh, C p = pf L R = 2.77 nh, C R = 3.05pF Dual Band Bandpass Filter Design and Measurement Results The procedures for coupled resonator Chevishev filter design highlighted in [90] are followed in this work. The dual band filter is implemented by cascading two resonators, as shown in Figure 6-23A. The mutual coupling coefficients of both frequency bands are controlled by the inter-resonator distance L IR, while the distance L IC fine tunes that of the second frequency band. The metal patch does not need to be centered with respect to the HMSIW, which allows 161

162 S 11, S 21 (db) S 11, S 21 (db) Mutual Coupling easy control of the mutual coupling coefficients. Figure 6-23b shows the variation of the coupling coefficients. L IC L IR A Coupling Lower band Coupling Upper band Inter-resonator distance L IR (mm) Figure Proposed dual band filter. A) Topology. B) Coupling coefficient variation. L IC is kept constant as 5.6mm. B S sim 11 S sim 21 S 11 meas S sim 11 S sim 21 S 11 meas S 21 meas Frequency (GHz) A S meas Frequency (GHz) B Figure Simulated and measured performance of the implemented devices. A) The dual band resonator. B) The two-pole filter. The measured 3dB bandwidths are: 10% at 3.5 GHz and 9 % at 5.8 GHz. 162

163 A two-pole coupled resonator narrow band Chevishev filter is designed for 3.5 GHz and 5.8 GHz operation and a 20 db bandwidth of 100 MHz at both bands. The filter is fabricated by using standard printed circuit board fabrication with a CNC milling machine. Figure 6-24 shows the full-wave simulation and measured results for the implemented filter. Less than 1.2 db measured insertion loss is obtained at both bands. It is worth to mention that since the filter has narrow bandwidth at both bands, the associate insertion loss is higher than that obtained for a wider bandwidth, because the resonator needs to have a higher external quality factor which implies a weaker external coupling to the feeding line. The slight frequency shifts and differences in the return loss band can be attributed to the tolerance in the fabrication process and the loss in the connectors. Nevertheless, simulated results are in good agreement with measurement ones. Figure 6-25 shows the implemented resonator and filter. B A Figure Photographs of the fabricated dual band devices. A) Resonator. B) Filter Summary In conclusion, the proposed alternative approach for implementing HMSIW dual band filters allows the relatively independent control of the external Q factor and coupling coefficient 163

164 at both frequency bands. It was demonstrated that the capacitive loading of the HMSIW also produces a resonant frequency below the original waveguide cutoff frequency, as the CSRR loading does. By using simultaneous loading, arbitrary frequencies of operations can be selected, demonstrating that the topology can also be a good candidate to implement miniaturized dual band filters. In addition, we believed the structure is useful for implementing narrow band and broadband filters with more than two poles and more than two bands Wireless Passive Sensing Application Using a Cavity Loaded Evanescent Mode HMSIW Resonator In this section, a compact cavity coupled CSRR loaded HMSIW resonator is used for implementing sensing applications at microwave frequencies by using the principle of electromagnetic transduction [99-100], in which the external perturbations in the electromagnetic field are converted into frequency shifts. A small PDMS cavity, whose upper side is confined with a metal coated thin membrane, is placed on top of the CSRR loaded HMSIW original resonator. The electromagnetic field in the close vicinity of the CSRR resonator is perturbed when the cavity upper membrane is pressed or deformed by an external event and a shift in the resonance frequency is produced. This resonance frequency shift can be sensed telemetrically, allowing wireless passive sensing of a variety of external events such as pressure and strain. Moreover, the sensing frequency is selected to be in the wireless carrier frequency and therefore no additional frequency conversion module or mixer is necessary for wireless data transmission. Furthermore, the use of an evanescent mode resonator allows significant size reduction in comparison with the use of conventional bulky waveguide, cavity, coplanar waveguide (CPW) or microstrip resonators. The planar circuit implementation of the half mode substrate integrated waveguide (HMSIW) architecture, combined with the easy microfabrication of the cavity and membrane in a variety of materials, such as PDMS in this case, makes the sensor compatible and 164

165 S11 and S21 (db) integrable with PCB and CMOS/MEMS processes while allowing the batch fabrication of multiple devices. Because it operates in a microwave frequency spectrum, it offers high pressure frequency sensitivity. Resonance frequency as a function of an applied pressure is presented. Also, a broadband antenna has been integrated to perform wireless interrogation of the sensor The Evanescent Mode Resonator The originally previously proposed evanescent mode resonator in section is used in this section as a sensor. The dimensions of the waveguide control the waveguide cutoff frequency and the quality (Q) factor. The dimensions of the complementary split ring resonator (CSRR) control the resonance frequency. As previously shown in section , the electric field at resonance is mainly concentrated on the conductors and the inner metal patch of the CSRR, which suggests that a perturbation of the electromagnetic field right on top or under the resonator can produce a shift in the resonance frequency S11 S21-40 Measurement Simulation Frequency (GHz) Figure Simulated and measured results for the resonator used in our study. 165

166 The measured and simulated results of the resonator are presented infigure The substrate Arlon Diclad 880 with a thickness of 0.508mm and a dielectric constant of 2.2 is used to implement a CSRR loaded HMSIW resonator. Metalized vias with a diameter of 0.75 mm and a separation s of 1.4 mm are used to provide the electric sidewall of the waveguide. The CSRR is etched on the metallic surface of the waveguide. A 50 feeding microstrip line is connected directly to the waveguide. The selected resonance frequency and dimensions of the resonator are summarized in Table 6-4. The obtained external quality factor Q e is Table 6-4. Dimensions in mm of the resonator. g=a=b=c l y =l x s p d w t f o =5.25 GHz w s =1.55mm l=1.8 mm Proposed Sensor Structures The cross sections of the proposed configurations of the sensors using the evanescent mode resonator are presented in Figure In Figure 6-27A, a dielectric cavity made of PDMS with a thin metal coated membrane on top is used to create perturbations in the electromagnetic field that shift the resonance frequency of the resonator. When the membrane is deformed, the height of the air gap changes, which produces a shift in the resonance frequency. For the architecture in Figure 6-27B, a cavity backed resonator is used, which uses a metal coated membrane as part of the ground plane that is deformed when the membrane is under pressure, changing the effective height of the dielectric bi-layer under the resonator and, at the same time, the effective dielectric constant under the resonator. These changes produce a shift in the resonance frequency of the resonator that can be telemetrically detected. In our study, the performance of the first structure is discussed, while the use of the second structure for sensing purposes is left as a future work. 166

167 Frequency (GHz) A B Figure Proposed sensor configurations. A) Resonator with cavity on top surface. B) Cavity backed resonator Air gap thickness h (um) Figure Variation of resonance frequency as a function of the air gap Effect of the Air Gap in the Resonance Frequency In order to analyze the performance of the resonator as a sensor, a numerical simulation of the resonance frequency on the variation of the air gap thickness, h, is performed by using the high frequency structure simulator (HFSS, Ansys Inc.), assuming a uniform variation of the deflection of the membrane as a first order approximation. The height of the cavity is selected to be 200 m. Figure 6-28 shows the change in the resonance frequency as a function of the air gap thickness. It is observed that the resonance frequency decreases as the height decreases. In this 167

168 simulation only the metallic membrane has been taken into consideration in order to analyze the effect of its interaction with the electromagnetic field on top of the resonator. It is observed that a linear change of 5 MHz/m in the resonance frequency is expected. A B Figure Mechanical simulation for the deflection as a function of an applied pressure. A) Simulated results. B) Deflection profile Mechanical Simulation of the Deflection In this application, the range of pressure that can be measured depends on the selected material to implement the metal coated membrane. For demonstration purposes, two widely used materials for MEMS applications have been selected: polydimethylsiloxane (PDMS) and Parylene with Young s modulus of 0.75 GPa and 3 GPa, respectively. Finite element COMSOL simulations of the maximum deflection of the membrane under uniform pressure have been performed. For the simulations, a metallic layer of 1 m Copper under the dielectric membrane has been used. A membrane thickness and area of 20 m and 5 mm 5 mm, respectively, have been selected. Figure 6-29A shows the simulation results of the maximum displacement as a function of pressure for coated and non-coated membranes. Since PDMS is a more elastic material than Parylene, the achievable deflection of PDMS is larger than that of Parylene for the 168

169 Return Loss (db) applied pressure. A pressure range of a few Pascal is used due to the low Young s modulus of the materials. On the other hand, when the membranes are coated with a thin metallic layer their flexibility is reduced, while allowing smaller deflections with the applied pressure. Figure 6-29B shows the simulated deflection profile for a uniformly applied pressure of 20 Pa to a non metal coated 20 m thick PDMS membrane. It is observed that the maximum deflection of the membrane at the center is 30 m. PW ML MW PL H1 Ground on back A B BroadBand Antenna Resonator with Antenna Frequency (GHz) C Figure Broadband antenna. A) Antenna. B) Integrated antenna and resonator. C) Measured return loss. 169

170 Wireless Interrogation Planar broad band antennas with a gain of 2 dbi are integrated to the sensor for interrogation purpose. Transmitting and receiving antennas are used to wirelessly determine the data of the sensor. Frequency sweep and far field measurements are used to interrogate the sensor. For demonstration purpose, the received signal, filtered by the sensor, is obtained by using a vector network analyzer HP8719D. Figure 6-30 shows the implemented integrated antenna and resonator with measurement results. It is observed that the return loss of the antenna is changed by integrating it with the resonator The dimensions of the antenna are: PW=11.82 mm, PL = mm, H1 = 0.67 mm, ML = 0.67 mm, MW = 5 mm Measurement Results A PDMS cavity with a height of 200m, and a PDMS membrane with an area of 5 mm 5 mm and a thickness of 90 m are fabricated. For sensing purpose, Ti/Cu layers with a thickness of 30 nm / 500 nm have been sputtered on the PDMS membrane. Also, a 20 m thick metal coated Parylene membrane is used for comparing the performance of the two materials. Measurement results of the integrated resonator and antenna under four different pressure conditions on the PDMS membrane are shown in Figure 6-31A. The sensitivity curve f/p and the calculated displacement sensitivity f/d curve are presented in Figure 6-31B. In Figure 6-31C the measured transmission response from the antenna to the sensor under pressure is presented, demonstrating that the sensor can be connected to another external antenna to completely perform wireless interrogation. Table 6-5 summarizes the obtained data for the two different membranes. Figure 6-32A shows the implemented sensor consisting of the integrated resonator and antenna with the cavity on top. Figure 6-32B shows the wireless transmission test at a distance of 10 mm. 170

171 S21(dB) Return Loss (db) Frequency (GHz) Displacement (um) 0 Pressure Increasing Cavity loaded Pressure (184Pa) Pressure (368 Pa) Pressure (552 Pa) Pressure (736Pa) No Cavity Frequency (GHz) A Pressure (Pa) B C Figure Measured results. A) Return loss as a function of frequency and pressure. B) Frequency and displacement sensitivity to pressure. C) Wireless transmission performance. Table 6-5. Summary of measurements results Membrane F/P F/d Thickness PDMS 16 MHz/Pa 6.5 MHz/m 90m Parylene 2.2 MHz/Pa 7.3 MHz/m 20m Summary Pressure Increasing No pressure Pressure (184Pa) Pressure (368Pa) Pressure (552Pa) Pressure (736Pa) Frequency (GHz) A passive wireless sensing scheme has been demonstrated using the CSRR loaded HMSIW resonator at microwave frequencies. The electromagnetic transduction principle is used 171

172 on a cavity loaded evanescent mode resonator for a pressure sensor. Size reduction of the resonator is achieved by using the evanescent wave amplification on the half mode substrate integrated waveguide. Perturbations on the electromagnetic field exciting the resonator are used for achieving resonance frequency shifts under pressure or deformation conditions. A broadband antenna is integrated to the sensor, demonstrating that the configuration can be a good candidate for implementing wireless passive sensing applications working at microwave frequencies. A B Figure Fabrication and test. A) Fabricated wireless sensor and antenna module. B) Wireless transmission test. 172

173 CHAPTER 7 MICROMACHINED EMBEDDED EVANESCENT MODE HMSIW BANDPASS FILTER This chapter presents the design and implementation of a 3D integrable, compact SU8 epoxy embedded evanescent wave half mode substrate integrated waveguide (HMSIW) microwave resonator and a two pole bandpass filter working at 12 GHz (Ku band) [83]. The devices were designed and fabricated by using a multilayer surface micromachining fabrication process using SU8 as the dielectric. Evanescent wave amplification with forward bandpass transmission below the characteristic waveguide cutoff frequency is achieved with the use of the previous complementary split ring resonator (CSRR) loaded half mode substrate integrated waveguide, resulting in compact resonators with controllable external quality factors and dual band operation capability. The in substrate implementation and embedded nature of the resonator and filter allows conventional handling and packaging of 3D filter microstructures without additional mechanical consideration, which otherwise would require a very delicate and expensive vacuum packaging process. In addition, the device is formed in a mold which is used for its fabrication and electrical performance degradation after further packaging is minimized [44]. Photopatternable SU8 epoxy has been selected as the dielectric for implementing embedded passive devices due to its electrical and mechanical properties that offer easy implementations of multiple layers for the fabrication 3D integrated passive devices, the capability for high aspect ratio vertical interconnection, low process temperature, good thermal stability, good adhesion, thick film forming capability with single coating, high resistance to solvents and electrochemical processes, the compatibility and integrability with CMOS/MEMS/MMIC processes, and the batch processability for multiple devices. Moreover, the use of integrated compact passive devices is suitable for System on Package (SoP) or System on Substrate (SoS) approaches to achieve compactness of the devices and systems. Additionally, since the printed circuit board 173

174 (PCB) FR4, silicon or glass can been selected as the supporting substrates for the micromachined filters, the compatibility with conventional microwave PCB implementations and integrated circuits is maintained. The design, fabrication and characterization of the proposed embedded resonator and filter are presented and discussed in detail. Full wave 3D electromagnetic structure simulations and circuital simulations based on the electrical equivalent circuit extraction at 12 GHz are presented. In the same way, measurement results at 12 GHz are provided. Variation of the external quality factor of the resonator from 12 to 19 is obtained by varying its geometrical parameters. The resonator and filter feature a great size reduction of more than 50% when compared with conventional PCB implementations. Further, due to the scalability and repeatability of the fabrication process, this 3D integrable implementation can be extended to the use of less lossy dielectric for microwave and millimeter wave frequencies, such as Benzocyclobutene (BCB). This approach for HMSIW bandpass filters can be used for implementing new multiband wireless and sensor applications and scaled from low GHz to millimeter wave frequencies, while allowing the use of diverse organic substrates for the fabrication of multilayer devices and systems D SU8 Embedded Resonator Design In Chapter 6, compact size, high selectivity and multiband capability resonators and bandpass filters were demonstrated based on the conventional printed circuit board (PCB) technology by integrating the half mode substrate integrated waveguide (HMSIW) technology with a complementary split ring resonator (CSRR), which provides forward wave propagation below the waveguide cutoff frequency and allows significant size reduction. The working principle of the resonator was fully explained. It was stated that the dimensions of the waveguide 174

175 control the waveguide cutoff frequency and the external quality factor of the resonator, while the dimensions of the complementary split ring resonator control the resonance frequency. On the other hand, the quasi-te 0.5,0 mode waveguide cutoff frequency for an HMSIW structure can be calculated with the previously introduced Equation Further optimizations provide the optimum dimensions. It is emphasized that the cutoff frequency of the waveguide is selected to be higher than the target resonance frequency of the resonator. Figure 7-1. Electrical equivalent circuit of the implemented CSRR loaded HMSIW resonator Figure 7-1 shows the electrical equivalent circuit for the CSRR loaded HMSIW resonator, which is an adapted version of the electrical circuit presented in [45]. The inductance L R and the capacitance C R model the CSRR, while the inductance Lv models the via wall of the waveguide, which gives its intrinsic highpass behavior. The capacitive and inductive coupling of the CSRR with the waveguide are modeled by the capacitance C C and L C, respectively. Insertion loss due to the transmission line and the quality factor of the resonator are modeled by the resistors R W and R R, respectively. 175

176 SU8 or BCB Metal Metal Figure 7-2. Cross section of the proposed SU8 embedded resonator. The design of the resonator is embedded in SU8 epoxy. SU (Micro-Chem, Inc.), is selected as the dielectric layer. Figure 7-2 shows the cross section of the proposed SU8 epoxy embedded resonator, previously shown as well in Chapter 3 and recalled in this section for clarity purposes. Multilayer fabrication is possible by adding multiple SU8 layers and interconnecting layers. For this design, the row of metalized vias is replaced by a long metalized via wall with its width of 100 m, more suitable for the microfabrication process. The heights h 1, h 2 and h 3 are selected to be 50 m each. Since the process uses surface micromachining, the supporting substrate can be either silicon, glass, or organic materials such as ones for printed circuit board. For this design, PCB substrate FR-4 with a dielectric constant of 4.1, comparable with the dielectric constant of the SU8 ( r = 3.4 at 10GHz), is selected as the carrier substrate. The resonator is implemented on top of the first SU8 layer. The ground plane is implemented on the second SU8 layer, and the electroplated vertical interconnects are used for the signal line and the vias to ground, as proposed in [44]. The upper layer of SU8 allows the implementation of GCPW or microstrip lines for feeding as well as different components such as wideband antennas. Here, two layers of SU8 and a top ground plane are used. 176

177 Figure D view of the embedded resonator. Figure 7-3 shows the 3D view of the SU8 embedded resonator structure drawn with the 3D high frequency structure simulator (HFSS, ANSYS, Inc.). The ground plane on top of the resonator has been removed for presentation purpose. The selected width of the HMSIW, w, is 1.4 mm which results from a calculated theoretical waveguide cutoff frequency of 29 GHz with r = 3.4. Table 7-1 shows the dimensions in millimeters of the resonator along with the geometrical dimensions of the complementary split ring resonator implemented on the top surface of the first SU8 layer of 50 m. Since no transition is necessary, a 50 SU8 embedded microstrip line is directly connected. It is observed that the overall size of the resonator, without taking into account the feeding lines, is smaller than the effective guided quarter wavelength in the medium at 12GHz ( mm), which offers a great size and area reduction in g comparison with conventional quarter-wavelength based planar implementations. Electroplated square vertical interconnects with a width of 100 m are used for connecting the embedded microstrip line with an upper 50 GCPW feeding line implemented on the top layer of SU8. 177

178 S11 and S21 (db) Table 7-1. Dimensions of the resonator W L (mm) H (mm) g (m) a (m) b (m) w (mm) l x =l y (mm) l (mm) Figure 7-4 shows the simulated results for the designed resonator. The resonance frequency is around 12 GHz, which is achieved by optimizing the dimensions of the CSRR resonator. It is observed that the transmission is obtained below the characteristic cutoff frequency of the waveguide, which is around 29GHz. The external Q factor of this doubly loaded resonator is calculated with Equation 7-1, where f r is the resonance frequency and BW 3dB is the 3 db bandwidth for S21. In this case Q = 14.3 for an offset distance of 0.2 mm. Q 2 f BW r (7-1) 3dB S11 EM S21 EM S11 Circuit S21 Circuit Frequency (GHz) Figure 7-4. Simulated performance of the resonator. The extracted circuital parameters are : L V = 0.5 nh, L C = 0.34 nh, L R = 1.17 nh, C R = pf, C C = 0.25 pf, R W = 4. Figure 7-5 shows the variation of the Q factor with the offset distance l. Electromagnetic simulations are performed for different distances and the external Q factor is calculated with Equation For a maximum distance of 0.25 mm, the Q factor is around 19. Although the 178

179 Q factor resonator shows a lower external Q factor than that of other SIW filter implementations, it shows advantages with regards to the size Figure 7-5. External Q factor variation with the input distance l Two Pole Embedded Filter Design Figure 7-6 shows the top view of the two pole embedded bandpass filter designed by using the coupled resonator filter design procedure [90]. The coupling coefficient between the resonators is controlled by the inter-resonator distance L IR, and is calculated with Equation 7-2 as follows: Distance l (mm) M f12 f 22 ( f f ), (7-2) with f 12 and f 22 as the lower and higher resonance frequencies that are obtained when the two resonators are close each other (even and odd mode). For example, a two pole single band Chebyshev filter for 12 GHz with a bandwidth of 500MHz for a return loss of 20 db is designed and implemented. Table 7-2 summarizes the calculation and dimensions. Table 7-2. Parameters of the embedded filter Q e M L i L IR BW mm 0.49mm 500MHz 179

180 S11 and S21 (db) S11 and S21 (db) Figure 7-7 shows the electromagnetic simulation results for the designed two pole filter. Since a relatively large loss tangent of the SU8 has been used (0.03), the simulated insertion loss is high. Nevertheless, the simulated results show that the embedded configuration is promising in terms of size reduction, scalability to higher frequencies and integrability with other devices. Finally, in Figure 7-7 is observed a 3 db bandwidth of around 750 MHz. L i Figure 7-6. Top view of the two pole embedded filter S11 S Frequency (GHz) A S11 S Frequency (GHz) B Figure 7-7. Electromagnetic simulation results of the two pole filter. A) Frequency response of the insertion and return loss. B) Close view of the 10 db bandwidth Fabrication Process In our study FR4 ( r = 4.1) printed circuit board (PCB) have been selected as the supporting substrate for the embedded devices, maintaining low cost and compatibility with 180

181 conventional microwave hybrids and integrated circuit implementations. Fig. 5.8 illustrates the fabrication procedure, which is very similar to that used in the fabrication of the multilayer CRLH devices in Chapter 4. First, the bottom and top copper layers of the FR4 are etched away with a solution of H 2 SO 4 :H 2 O 2 :Deionized-Water (1:1:3). Standard substrate cleaning is then performed with TCE (trichloroethylene), followed by rinse with Isopropanol, DI water and dehydration on top of a hot plate at 120 degrees for 10 minutes. SU8 SU8 SU8 Substrate Substrate Copper Ti/Cu seed layer Via interconnection SU8 SU8 SU8 SU8 Negative Photo Resist Electroplated copper SU8 SU8 Copper Figure 7-8. Fabrication process. The first layer of SU8-2025, which is used as an interface layer in order to minimize the electromagnetic field interactions with the carrier substrate, is spin-coated at 1400 RPM for 30 seconds to get an approximate thickness of 50 m. Edge bead is further removed. Soft bake is performed on a leveled hot plate with the temperature ramped up to 65C at a rate of 250C/hour and kept for 30 min. Temperature is ramped up to 95C at the same rate and kept for 20 minutes. 181

182 The samples are allowed to cool down to room temperature on the hot plate. Lithography with an MA6 Karl Suss Mask Aligner (UV wavelength of 365nm) is performed with an optical dose of 200 mj/cm 2. Post exposure bake is performed at 65C and 95C on leveled hot plate. The samples are allowed to cool down at room temperature, developed in SU8 developer for 30 minutes and finally rinsed with Isopropanol and blown dry with a nitrogen gun. A low temperature curing process of the SU8 is performed at 120C during 10 minutes. Further, a metallic seed layer of Ti/Cu/Ti (30 nm / 300 m / 30 nm) is sputtered on top of SU8 followed by negative photoresist patterning with NR4-8000p (Futurrex Inc.) for the embedded resonator and feeding lines. After the etching of the upper Titanium layer, Copper electroplating is performed for 5 m thickness, much thicker than the skin depth of copper at 12 GHz to minimize ohmic losses. At the end of the electroplating procedure, the Ti/Cu seed layer is sequentially removed. Ti is etched away using diluted hydrofluoridic acid ( HF:DI water, 1:10 ratio). The second layer of 50 um SU is then coated and the lithographical pattern of the interconnect vias and HMSIW via wall is performed. A second metallization of Ti/Cu/Ti seed layer is performed using sputtering, followed by negative photoresist patterning with NR for the upper CPW feeding lines and other devices. After copper electroplating up to 5 m, the seed layer is removed. Figure 7-9A and Figure 7-9B show scanning electron microscopy (SEM) images of the fabricated embedded resonator and filter without the top SU8 layer for presentation purpose. Figure 7-9C shows the patterned and metallized vias on SU8 prior Copper electroplating. Figure 7-9D shows the final fabricated filter with the top ground plane and the GCPW feeding lines already electroplated. This fabrication process can be easily extended to the implementation of more SU8 layers by repeating the coating, lithography and electroplating procedures. 182

183 A B C D Figure 7-9. Scanning electron microscopy (SEM) images of the embedded resonator and filter. A) The embedded resonator. B) the embedded filter without the second SU8 layer. C) patterned vias before electroplating. D) final device after the electroplating of the top ground plane and GCPW feeding lines Measurement Results Measurement results of the fabricated resonator and filter are performed by using the setup described in Chapter 4. Figure 7-10 shows the measurement results. In Figure 7-10A the passband of the reronator is obtained around 12 GHz as expected. Insertion losses of the resonator and filter show good agreement with the simulated results, although the differences might be due to different values of the dielectric constant and the dielectric loss tangent of the SU8, as well as the tolerances of the fabrication process. It is believed that the design of the filter is not completely optimized, since its return loss, shown in Figure 7-10B, is very low. This might 183

184 be caused by a low internal coupling coefficient between the coupled resonators. Additional work can be done in order to obtain a better response by optimizing the inter-resonator distance and thus, the internal coupling coefficient.. To highlight the advantages this micromachined embedded approach offer, Table 7-3 shows a comparison in size of waveguide resonators at 12 GHz fabricated in different technologies. It is observed that more than 95% size reduction in comparison with the conventional SIW resonator and more than 80% size reduction in comparison with the conventional HMSIW evanescent mode resonator implemented on a PCB approach are obtained, which confirm that the micromachined approach is useful for the miniaturization and 3D integration of passive devices at microwave frequencies and can be easily sacaled to higher frequencies of operation. Table 7-3. Comparison of resonator size in different technologies Type a (mm) b (mm) d (mm) Area (ad) mm 2 Filled Waveguide WR-90 with r = SIW on RO 4003, Thickness =0.508mm HMSIW evanescent mode resonator on RO4003, r = SU8 embedded HMSIW evanescent mode resonator Summary This chapter has presented the design, simulation and fabrication of compact multilayer micromachined, SU8 epoxy embedded evanescent wave half mode substrate integrated waveguide resonators and filters. A CSRR loaded on the top surface of the waveguide allows forward transmission below the waveguide cutoff frequency. Electromagnetic simulations show good agreement with measurement results as expected from the design specifications. The selection of SU8 as a dielectric layer, in despite of its relatively large loss, is based on good 184

185 thermal, mechanical and electrical properties and low cost. More than 80% size reduction, in comparison with the conventional PCB based devices is achieved for the resonator and filter. Although insertion loss seems to be large in comparison with conventional implementations, mainly due to the relatively large loss tangent of SU8, the surface micromachined process on SU8 still can be used for achieving compact circuits at microwave frequencies. The prototypes are fully fabricated and tested. The approach to implement embedded substrate integrated waveguide resonators and bandpass filters is suitable for System in Package or System in Substrate devices. Extentions of this work might include the use of BCB as a dielectric for implementing single and dual band micromachined embedded filters at millimeter wave frequencies. A Figure Measurement results. A) Insertion and return losses of the embedded resonator. B) Insertion and return losses of the embedded filter. B 185

186 CHAPTER 8 EVANESCENT MODE BROADBAND BANDPASS FILTERS As previously studied, bandpass filters are key components of communication systems. During decades, special attention has been paid to the design of high performance, compact size and low cost bandpass filters using standard printed circuit board (PCB) and low temperature cofired ceramic (LTCC) processes [90]. Recently, the substrate integrated waveguide (SIW) technique and its reduced mode versions that make use of fictitious magnetic walls, namely the half-mode (HMSIW) and the quarter-mode (QMSIW) substrate integrated waveguides, have been studied for the design of integrated planar filters with similar performance to that of the conventional bulky waveguide filters [41-43]. The advantages of the SIW technologies include modest loss, high power handling capability and integrability with other planar technologies. Moreover, the QMSIW has been used for the design of broadband filters with more than 5% bandwidth due to its reduced external quality (Q e ) factor, which is not easily achievable with the original SIW [43]. On the other hand, withto the negative properties offered by metamaterials, i.e. negative permittivity and permeability, the complementary split ring resonator (CSRR) and the composite right/left-handed (CRLH) technique have been combined with the SIW and HMSIW to realize bandpass filters working below the cutoff frequency of the dominant TE 10 mode [31, 45, 46]. In this chapter, a new reduced mode version of the SIW, the eighth-mode substrate integrated waveguide (EMSIW) cavity, which allows a size reduction of 87.5% with respect to the original SIW cavity, is proposed [101]. In addition, the EMSIW is loaded with a CSRR on its top surface in order to obtain a new resonance frequency below the resonance frequency of its original quasi-te dominant mode; therefore, its size is further reduced. Moreover, the CSRR loading technique is also applied to the QMSIW cavities for implementing broadband bandpass 186

187 filters. Section 8.1 studies the proposed EMSIW and CSRR-loaded EMSIWcavities, in which the working principle and parameters of the cavities are provided and then, a set of two-pole bandpass filters are implemented on PCB and tested. Section 8.2 proposes the CSRR loaded QMSIW cavitiy for broadband filters working at quasi-millimeter wave and millimeter wave frequencies. Two dielectric substrates are selected: Liquid crystal polymer (LCP) with a dielectric constant of 2.9, and Benzocyclobutene (BCB) with a dielectric constant of Surface micromachining techniques are used to implement these filters Broadband Bandpass Filters using CSRR Loaded Eighth-Mode Substrate Integrated Waveguide Cavities The working principle and parameters of the EMSIW cavity are studied theoretically and numerically. Also, a set of two pole bandpass filters is implemented by using the proposed cavities. The design concept is demonstrated with one direct fed filter and one filter with proximity feeding that allows the generation of two transmission zeros in the stop-band General Requirements for Broadband Bandpass Filter Design Broadband bandpass filters, which are considered to have more than 5% fractional bandwidth [43], are also widely used in radio-frequency engineering. Frequency bands such as Ultra Wide Band (UWB from 3.1 GHz to 10.6 GHz), ISM band from 57 GHz to 64 GHz, W band from 77 GHz to 110 GHz, WiFi applications from 5.25 GHz to 5.85 GHz, require not only narrow band filters that cover part of the frequency rage, but also broadband filters that cover the entire band. In addition, modern technologies require low cost, low loss and integrable filters. Conventional designs using waveguide or SIW cavities are not suitable for broadband filter design due to their intrinsic high unloaded and loaded quality factors. The general requirements for broadband filter design are summarized as follows: 187

188 Cavity resonators with a low external quality factors, which means more energy is coupled from the source to the cavities and wider bandwidths are achievable. High inter-resonators coupling coefficients, which means that the energy transfer between neighbor cavities is high, indicating that wide bandwidths can be achieved. The conventional waveguide or SIW cavities cannot offer the requirements for broadband filters, mainly because of their big sizes that feature high unloaded and loaded quality factors and low internal coupling coefficients. The next sections demonstrate that the proposed EMSIW cavities in our study have lower achievable external quality factors and higher internal coupling coefficients than the conventional SIW cavities, indicating that they are useful for the design of broadband bandpass filters The Eighth-Mode SIW Cavity Figure 8-1A illustrates the electric field distribution in an original SIW cavity designed for 11 GHz. Four fictitious magnetic walls are shown for analysis purposes. Metallic solid side walls are used for simulation purposes. All designs use the substrate Arlon DiClad 880 with a thickness of mm and a dielectric constant r = 2.2. The components of the electric and magnetic fields in the original SIW cavity resonator for the TE 101 mode (a = d) are given by [57] : E E H 0, (8-1) x y z E y x z E0 sin sin, (8-2) a a H x j10e0 x z sin cos, (8-3) k a a and H z je0 x z cos sin, (8-4) ka a a 188

189 where k,, and E 0 are the phase constant, the electrical permittivity and the magnetic permeability of the medium, and the amplitude of the electric field, respectively. The propagation constant is k / a, and the intrinsic impedance of the medium is [57]. A B C y x z d D D d/2 C B a A A a/2 B C D Figure 8-1. The eighth-mode substrate integrated waveguide (EMSIW). A) TE 101 mode (a = d) E-Field in the SIW cavity. B) E-Field in the proposed EMSIW cavity. C) E-Field in the proposed CSRR-loaded EMSIW at 8 GHz. D) E-Field in the CSRR-loaded EMSIW at 14 GHz. For quasi-te mode propagation and the existence of almost perfect magnetic walls in the reduced mode versions of the SIW, the thickness of the substrate, b, must be much smaller than the width of the waveguide, a, ( b << a). From Equations 8-3 and 8-4 it is observed in Figure 189

190 8-1A that perfect magnetic walls are available for the planes AA ( z = x ), BB ( x = a/2 ), CC ( z = a-x), and DD ( z = a/2 ). Then, by cutting the SIW cavity with the centered and diagonal magnetic walls, eight triangular cavities are created. As observed in Figure 8-1B, the triangular cavity, named here the eighth-mode SIW (EMSIW), supports the propagation of one eighth of the original TE 101 mode with its maximum intensity at the corner of the structure. In our study, we call the propagating mode in the EMSIW the quasi-te 0.125,0,0.125 mode because it resembles the eighth part of the original TE 101 mode. Therefore, the EMSIW features a reduced size with respect to the original SIW and can be useful for the design of compact bandpass filters The CSRR-loaded Eighth-Mode SIW Cavity To further reduce the size, a complementary split ring resonator (CSRR) is loaded on the top surface of the EMSIW, as shown in Figure 8-1C and Figure 8-1D. A single ring CSRR is chosen for simplicity. Figure 8-1C and Figure 8-1D also show the two first resonant modes in the CSRR-loaded EMSIW designed for 8 GHz. The working principle is similar to that studied in Chapter 6 for the CSRR loaded HMSIW resonator, and it is based on the epsilon negative behavior of a rectangular waveguide under the cutoff frequency. The lower mode at 8GHz, shown in Figure 8-1C, is due to the resonant negative permittivity property of the CSRR loaded on the surface of the waveguide, which causes the resultant effective dielectric constant eff of the EMSIW to be positive and provides a frequency band with forward wave propagation below the original cavity resonance. The original TE 101 -like mode, shown in Figure 8-1B for 11 GHz, is moved up to a frequency of 14 GHz due to the fact that the CSRR confines the electric field to a small area, as shown in Figure 8-1D. 190

191 External Q Factor L e L e (mm) A Figure 8-2. Parameters of the EMSIW cavities. A) External quality factors. B) The layout of the CSRR loaded EMSIW cavity. Physical dimensions are g = 0.35 mm, a =13 mm, and w f = 2.4 mm Resonator Analysis and Design With CSRR No CSRR A set of two-pole filters using magnetically coupled CSRR-loaded EMSIW cavities for operation at 8 GHz X-band. The classic methodology for the design of coupled resonators filters is followed [90], in which the filter is determined by the external quality factor Q of the resonator and the internal coupling coefficient k between the coupled resonators. First, the external quality factor Q is analyzed. Figure 8-2A shows the variation of the external Q factors with the offset of the feeding line for single loaded EMSIW and CSRR-loaded EMSIW cavities. An array of metalized vias is used for the real implementation with a via diameter of 0.75 mm and a center to center pitch of 1.4 mm. The resonator can be excited by using one of the open sides. A direct connection of a 50 feeding line to the open side derived from the magnetic wall BB is used to excite the cavities. A square single split CSRR is loaded on the top surface with a slot width of 0.25 mm, a split gap of 0.2 mm and a side of 2.6 mm. As observed in Figure 8-2A, the Q factors of both cases are varied with the offset distance L e of the feeding. Since the EMSIW cavities are smaller than their original SIW counterparts, the available internal and external quality factors of the EMSIW are small, which is useful for broadband filters w f g a/2 B 191

192 Coupling Coefficient k W i (mm) Figure 8-3. Characterization of the EMSIW cavities. A) Internal coupling coefficients. B) Layout of the coupled EMSIW cavities. The internal coupling coefficients k of the coupled cavities with and without the CSRR, which are controlled by a window of size W i in either side of the metalized via wall, are presented in Figure 8-3A. The method used in Chapter 6 to get the internal coupling coefficient is also used in this section, in which the cavities are excited with a weak external coupling that offers a high external quality factor. As observed, because of the small size of the cavities, higher coupling coefficients than those offered by the SIW and HMSIW cavities [41-43] are achievable, which is also useful for broadband filters. A With CSRR No CSRR B W i Two Pole Filters Designs The previous section has demonstrated that the proposed EMSIW cavities are suitable for broadband bandpass filter design, because of their low external quality factors and the high internal coupling coefficients. In this section, a set of two-pole Chevyshev filters with 20 db passband return loss is designed at 8 GHz for demonstration purposes. One direct-fed filter with a fractional bandwidth of 7.5 % and one proximity-fed filter with source-load coupling [90] and a 10 % fractional bandwidth are implemented. Figure 8-4A and Figure 8-4C show the layout of each filter. Standard PCB fabrication on a CNC milling machine is used. The side length of the 192

193 CSRR in the proximity fed filter is slightly decreased to 2.4 mm in order to compensate for the frequency shift due to the gap feeding [90]. Table 8-1summarizes the calculated parameters for the filter by following the design procedures in reference [90]. w i L e A B l c w t g t g c w c w i C D Figure 8-4. Two-pole filters. A) Direct fed filter. Geometrical parameters are : w i = 1.9 mm, L e = 2.5 mm. B) Photograph of the fabricated direct fed filter. C) Gap fed filter. Geometrical parameters are: w t = w c = 0.3 mm, g c = g t = 0.4 mm, l c = 6 mm, w i = 2.56 mm. D) Photograph of the fabricated gap fed filter. Table 8-1. Specification and calculated parameters of the filters Filter FBW/ Return Loss External Q Factor Internal Coupling Coefficient k Direct feeding 7.5%, 20 db Gap feeding 10%, 20 db

194 Results and Discussion The fabricated filters are shown in Figure 8-4B and Figure 8-4B. Measurements are performed with a vector network analyzer HP8719D after standard SOLT (short-open-load-thru) calibration. The effects of the connectors have not been de-embedded, which can cause additional loss. Figure 8-5A and Figure 8-5B show the frequency responses for the direct and proximity fed filters, respectively. The total area of both filters, excluding the feeding lines, is , with 0 as the free space wavelength at 8 GHz. Measured center frequencies of 7.92 GHz with a 3 db bandwidth of %, and 7.9 GHz with a 3 db bandwidth of % are obtained for the direct and proximity fed filters, respectively. The measured in band insertion and return loss are better than 1.5 db and 13 db for both filters. Measurement results agree well with simulated one. Minute differences are observed mainly due to the fabrication tolerances, especially the fabrication of the gaps, and connector loss. It is observed that the out of band rejection is better than 15 db and that the second order mode is suppressed in both cases Summary Eighth-mode substrate integrated waveguide (EMSIW) cavities are proposed and studied for broadband filter design. The EMSIW features a reduced size in comparison with the original SIW. Moreover, conventional implementations of two pole filters on microstrip show larger size than that achieved in this work. The size is further reduced by using CSRR-loaded EMSIW cavities. On the other hand, the size can also be reduced if the split gap and the ring width of the CSRR are reduced, which were limited by the fabrication technology used in this work. The Q- factors and internal coupling coefficients have been studied in detail. The concept has been demonstrated by implementing a set of CSRR-loaded EMSIW two-pole filters. Higher order filters and different coupling schemes could be also implemented. 194

195 S 11, S 21 (db) S 11, S 21 (db) 0 S S Measurement Simulation Frequency (GHz) A S 21 S 11 Measurement Simulation Frequency (GHz) B Figure 8-5. Measured and simulated results. A) Direct fed filter. B) Gap fed filter Surface Micromachined Broadband Millimeter Wave Bandpass Filters Using CSRR Loaded Quarter Mode Substrate Integrated Waveguide Cavities The previous section proposed the eighth mode substrate integrated (EMSIW) waveguide cavity for broadband bandpass filter design, which features a size reduction of 87.5% with respect to the original substrate integrated waveguide (SIW) cavity. In this section one more step 195

196 toward the miniaturization of broadband bandpass filters is taken. Due to the limits imposed by the standard printed circuit board fabrication technology for implementing devices working at millimeter wave frequencies, the use of surface micromachining techniques is proposed for the implementation of compact broadband filters working at millimeter wave frequencies. For this work, the quarter mode substrate integrated waveguide (QMSIW) loaded with a single ring complementary split ring resonator (CSRR) is selected as the waveguiding structure. It is also demonstrated that the CSRR loaded QMSIW is able to be excited with low external quality factor and that high inter-resonator coupling coefficients can be achieved, which is useful for broadband filter design. On the other hand, due to their good performance at microwave and millimeter wave frequencies, two dielectric materials have been selected for the implementation of the filters : The flexible liquid crystal polymer (LCP) and the photopaternable Benzocyclobutene resin (BCB). Flexible LCP Ultralam 3850 from Rogers Duroid with a thickness of 4 mil (101.6 m), a relative dielectric constant r = 2.9 and a loss tangent of is selected for the implementation of two pole and three pole QMSIW filters working at 25 GHz GHz. In order to achieve higher frequencies of operation, BCB with a thickness of 21m is used to implement two and four pole filters working at 60 GHz. At first, the QMSIW and the CSRR loaded QMSIW cavities are introduced. Nezt, the design and simulation of two and three pole CSRR loaded QMSIW filters using LCP as a dielectric material are presented. The filters are designed for operation at 25 GHz. The fabrication process is also studied. Finally, the design and simulation of two and four poles filters using BCB as a dielectric are studied. The fabrication of the filters is not done at this time, therefore only design concept and simulation results are presented at this time. However, the fabrication and measurement are left as a future work of this study. 196

197 The CSRR loaded Quarter Mode Substrate Integrated Waveguide Cavities The original SIW cavity shown in Figure 8-6A, previously shown in section 8.1, is recalled in this section for clarity purposes. In addition, the electric field distribution on two different versions of the QMSIW cavity are also presented in Figure 8-6B and Figure 8-6C. As observed, the QMSIW cavities feature one fourth of the size of the original SIW cavity while allowing the wave propagation in a mode that resembles one fourth of the original TE 101 mode of the SIW, called here the quasi-te 0.25,0,0.25 mode. The fictitious magnetic walls can be used to get square and triangular shape QMSIW cavities. On the other hand, when the CSRR is loaded on the top surface of the QMSIW, as shown in Figure 8-6D, the electric field is concentrated around the CSRR and the propagating mode is completely different to that of the original SIW and QMSIW cavities. Moreover, the CSRR loading creates a new resonance frequency below the original resonance of the QMSIW cavity, which also has the same working principle previously studied for the CSRR loaded EMSIW cavity. Similar results are obtained when a CSRR is also loaded on the triangular shape QMSIW cavity, although not shown here. In order to demonstrate the design concept, the next two sections present the design and simulation of the two, three and four pole filters using the proposed CSRR loaded rectangular QMSIW cavities The Design of CSRR loaded QMSIW Bandpass Filters on Flexible LCP Flexible liquid crystal polymer (LCP) Ultralam 3850 from Rogers Duroid, with a thickness of 4 mil (101.6 m) is selected as the dielectric material for implementing broadband filters working at 25 GHz frequency band. For broadband design, the fractional bandwidth (FBW) is selected to be higher than 5%. For demonstration purposes, two pole and three pole filters were designed and simulated. Table 8-2 summarizes the design specifications of the proposed filters. 197

198 The fabrication procedure is proposed based on the recommended processing guidelines given by Rogers Inc. for the liquid crystal polymer substrate [102]. Table 8-2. Design specification for the filters on LCP Filter f o (GHz) FBW Passband Return QMSIW Cavity Loss Two poles % 20 db Rectangular Three poles % 20 db Rectangular A B C y x z d D D d/2 C A B a A a/2 B a d/2 C D Figure 8-6. The quarter mode substrate integrated waveguide (QMSIW) cavities. A) Electric field (E-field) distribution in the original SIW cavity for TE 101 mode. B) The rectangular QMSIW cavity. C) The triangular QMSIW cavity. D) The proposed CSRR loaded QMSIW cavity. 198

199 Resonator study The filter design methodology explained in previous sections is used in this section [90]. The first step is the characterization of the resonator, from which the achievable external quality factors (Q e ) and internal coupling coefficients (k) are obtained. The inset in Figure 8-7 shows the designed resonator to work around 25 GHz. The original frequency of the QMSIW cavity with no CSRR loading is 48 GHz, which means that a size reduction of nearly 50% is achieved by the CSRR loading. Four different orientations of the CSRR are possible, while the one shown in Figure 8-7 has been chosen for this design. The external quality factor is controlled by the offset distance L q and also the offset position l o of the CSRR in the cavity. However, the offset position l o is selected to be 0.2 mm and is kept constant. Metalized vias with a diameter of 0.2 mm and a center-to-center pitch of mm create the metallic sidewalls. As observed, external quality factors as low as 6 can be obtained. Next, the internal coupling coefficient between magnetically coupled cavities is studied. The inset in Figure 8-8 shows the used configuration for such purpose. The cavities are excited with a high external Q factor feeding in order to have a weak coupling from the source. The internal coupling coefficient is controlled by the window in the via row W i. The extraction methodology explained in section is used in this section. Equation 6-17 is used to get the coupling coefficient. Figure 8-8 shows the obtained results. As observed, internal coupling coefficients higher than 0.15 are possible, indicating the CSRR loaded QMSIW cavity is useful for broadband filter applications. 199

200 40 35 External Quality Factor Q e Internal coupling coefficient k l o L q l o c g w Lq (mm) Figure 8-7. Extracted external quality factor (Q e ) of the resonator. The inset shows the configuration of the QMSIW cavity. Geometrical parameters are : l o = 0.2 mm, c = 0.1 mm, g = 0.15 mm, w = 1.35 mm W i W i (mm) Figure 8-8. Extracted internal coupling coefficient k. The inset shows the configuration of the magnetically coupled cavities. 200

201 S 11 (db) S 11 phase (deg) Finally, Figure 8-9A shows the simulated frequency response of the return loss of a single ended CSRR loaded HMSIW cavity. The phase-shift and additional loss provided by the feeding line are de-embedded. Figure 8-9B shows the phase angle of the return loss (S 11 ). These two plots are used to explain the procedure used in this research to extract the external quality factor (Q e ) of a single loaded cavity, which is explained in detail in reference [90]. Since the insertion loss is not available, the extraction is not based on obtaining the 3 db bandwidth as previously explained is section , but on the analysis of the phase angle of the return loss [90]. For this purpose, the resonance frequency of the cavity, f o, is obtained from the return loss plot and two frequency points, f L and f H, are extracted from the phase plot when the phase angles are ± 90 degrees, as observed in Figure 8-9B. Then, the external quality factor Q e is calculated as in Q e o. (8-4) f H f f L For this case in particular, f o = GHz, f H = GHz and f L = GHz, which results in Q e = f L deg deg f H Frequency (GHz) Frequency (GHz) A Figure 8-9. Simulated frequency response of a single ended CSRR loaded QMSIW cavity. A) Return loss. B) Phase angle. 201 B

202 Bandpass filter designs and simulations Based on the design specifications and the previous resonator study, two pole and three pole bandpass filters are designed. Table 8-3 summarizes the calculated design parameters [90]. The initial physical dimensions of the filter are selected based on the resonator study and optimized through full wave structure simulations in HFSS. Figure 8-10 shows the physical configuration of the two and three pole filters. Since the filters are to be tested with a Cascade Microtech probe station with GSG (ground-signal-ground) probes and a picth distance of 150 m, grounded coplanar waveguide (GCPW) launching feeding lines are used. The thickness of the metal trace (18 m of Copper) and the dielectric loss tangent of the LCP (tan = 0.002) have been taken into account for the simulations. In Figure 8-10B, the CSRR in the middle of the three pole filter is scaled up with a factor of in order to optimize the frequency response. Table 8-3. Design parameters of the filters on LCP Filter f 0 FBW Return Loss Q e (1,2) k 12 k 23 Two poles % 20 db Three Poles % 20 db W i W i W i A B Figure Physical layout of the bandpass filters on LCP. A) Two pole filter. Geometrical parameters are : w = 1.3 mm, g = 0.18 mm, W i = 0.51 mm, l x = l y = 0.9mm, l o = 0.18 mm, L q = 0. B) Three pole filter. Geometrical parameters are the same for the two pole filter except for : W i = 0.51 mm, l y2 = 1.01 mm. 202

203 S 11, S 21 (db) Figure 8-11 shows the simulated results for the designed two pole filter. A maximum insertion loss of 0.51 db is expected within the passband. Also, a fractional bandwidth of 11.7% for a 20 db return loss is obtained around a resonance frequency of 25.5 GHz. The simulation results agree well with the design specifications. The size of the filter is only , where 0 is the free space wavelength at 25.5 GHz. A 3dB bandwidth is obtained from 22 GHz up to 29 GHz, which covers the entire 24 GHz automotive band. The out of band rejection is better than 15 db at 32 GHz, which is 10% higher than the 3 db bandwidth upper frequency. The effects of the feeding lines have not been extracted, which indicates that the real insertion loss of the filter is lower S S Frequency (GHz) Figure Simulated results for the two pole bandpass filter on LCP. The simulated resonance frequency is 25.5 GHz. The obtained 20 db return loss fractional bandwidth is 11.7%. Less than 0.51dB insertion loss is expected within the 20 db passband. Figure 8-12 shows the simulated results for the three pole filter. As observed, a clear three pole response is obtained. A maximum insertion loss of 0.69 db is expected within the 203

204 S 11, S 21 (db) passband. A fractional bandwidth of 17% for a 20 db return loss is obtained around a resonance frequency of 25.7 GHz. The simulation results agree well with the design specifications. The size of the filter is only , where 0 is the free space wavelength at 25.7GHz. A 3dB bandwidth is obtained from GHz up to 28.57GHz. The out of band rejection is better than 15dB at 30 GHz, which is 5% higher than the 3 db bandwidth upper frequency. The effects of the feeding lines have not been extracted; therefore, the real insertion loss of the filter might be less S 11 S Frequency (GHz) Figure Simulated results for the three pole bandpass filter on LCP. The simulated resonance frequency is 25.7 GHz. The obtained 20 db return loss fractional bandwidth is 17%. Les than 0.68 db insertion loss is expected within the 20 db passband Proposed fabrication procedure on LCP The proposed fabrication of the bandpass filters is based on a surface micromachining process on the liquid crystal polymer (LCP) substrate Rogers Ultralam 3850 with a thickness of 204

205 4 mil and a dielectric constant of 2.9. Since only one layer of LCP is used in this project, the fabrication starts with the etching of the top Copper layer of a double clad LCP sheet. Then, mechanical drilling of the via holes, desmear of the vias in a heated ultrasonic bath, oxygen plasma cleaning, metal deposition, lithographical pattern and electroplating are used for fabricating the devices. Figure 8-13 summarizes the proposed fabrication process. Substrate: 4 mil LCP Rogers Ultralam Top Copper is removed Vias : CNC drilling of the vias and alignment marks. Via desmear and substrate cleaning. 1 minute Oxygen plasma cleaning and DC sputtering of Ti/Cu/Ti seed layer (30nm/300nm/30nm) Lithography patterning of the Copper layer with NR negative resist. 18 m Copper electroplating. Etch seed layer LCP Copper Ti/Cu/Ti seed layer NR Photomask Electroplated copper Figure Proposed fabrication process of the LCP filters The Design of CSRR loaded QMSIW Bandpass Filters on BCB Although LCP has good properties for the implementation of microwave and millimeter wave applications, as the frequency increases the integration with CMOS circuitry and processes becomes more important. Bulk and surface micromachining techniques offer an excellent way to fabricate small devices for millimeter wave applications, but some steps might require expensive equipment and processes in order to continue fabricating smaller features on non-photosensitive 205

206 dielectric materials, such as laser or deep RIE for through substrate via drilling. To overcome these limitations, Benzocyclobutene (BCB, Cyclotene from Dow Chemical) with a thickness of 21 m on CMOS grade low resistivity Silicon or glass is proposed as the dielectric material for implementing surface micromachined broadband filters working at the unlicensed 57GHz to 64 GHz frequency band. Different work has previously demonstrated micromachined narrow band waveguide cavity filters working at millimeter wave frequencies by using complicated elevated structures [103, 104] or multilayers of photoimageable thick pastes [105], which might not be completely compatible with conventional CMOS/MMIC/MEMS processes. However, no much work has been reported on wideband bandpass filters working with miniaturized cavities that make use of metamaterial concepts. We believed that the use of metamaterial concepts in combination with a post-cmos micromachined process using properly selected low loss polymers for interfacial layer, is useful for implementing low profile and low cost integrable filters for millimeter wave applications. Since the BCB is a photosensitive resin with multilayer capabilities, thicker thickness might be used in order to increase the quality factors of the resonators. In our study, a single coated layer of 21 m of BCB is used for demonstration purposes. The 3D cross section view of the proposed filters was previously introduced in Chapter 3. The thickness is achieved based on a modified coating and processing of the BCB. The Appendix gives the material specifications and the fabrication procedures. For demonstration purposes, two pole and four pole filters have been designed and simulated. Table 8-4 summarizes the design specifications and calculated parameters of the proposed filters. Table 8-4. Design specifications and calculated parameters of the filters on BCB Filter f 0 FBW Return Loss Q e (1,2) k 12 k 23 k 34 Two poles % 20 db Four Poles % 20 db

207 External Q factor Figure 8-14 shows the extracted external quality factors for a CSRR loaded QMSIW cavity resonator working at 60.5 GHz. External quality factors as low as 5.6 can be obtained, which is useful for broadband filters. The inset of Figure 8-14 shows the layout of the cavity resonator. Since a micromachining process is to be used, there is no necessity to follow the conventional SIW topology with a metalized via row. Therefore, a complete metalized wall with a width of 80 m is used for implementing the sidewalls of the cavity. The square CSRR has a side length of 0.4 mm, a ring gap of 0.1 mm and a ring width of 60 m. The side length of the square cavity is 0.6 mm. The resonance frequency of the original QMSIW cavity with no CSRR loading is 102 GHz, which indicates a size reduction of more than 40% when the CSRR is loaded on the cavity. The size can be further reduced if the width of the ring of the CSRR is selected smaller L q l y w Lq (mm) Figure Extracted external quality factor of a CSRR loaded QMSIW cavity on BCB. The internal coupling coefficient between magnetically coupled cavities is presented in Figure 8-15, where the inset shows the coupled cavities with a high external quality factor excitation. As observed, due to the small size of the cavities, internal coupling coefficients higher than 0.15 are also obtained, which makes the cavities on BCB are useful for broadband filters. 207

208 Coupling Coefficient (M) Window Window (mm) Figure Extracted internal coupling coefficient for magnetically coupled CSRR loaded QMSIW cavities on BCB. Window A w 23 l cx 2 3 l x w 12 w 34 L q 1 4 B Figure Layout of the proposed filters. A) The two pole filter. Geometrical parameters are: w = 0.6 mm, l y = mm, L q = 0.13 mm, window = 0.33mm. B) The four pole filter. Geometrical parameters are: w 12 = w 24 = mm, w 23 = mm, L q = 0.5 mm, l y = mm, l x = mm, l cx = 0.54 mm. 208