High Step-up Boost Converter Integrated with Voltage-Doubler

Transcription

1 High tep-up Boost Converter Integrated with oltage-doubler Ki-Bum Park, Gun-Woo Moon, and Myung-Joong Youn Department of Electrical Engineering, KAIT, Daejeon, Republic of Korea Abstract -- The voltage-doubler provides an additional stepup gain on top of that of the boost converter, while distributing voltage stresses on devices as well. The interface between the boost converter and the voltage-doubler is accomplished by a transformer and a balancing capacitor, which also constitute a resonant tank. ince this resonant operation shapes the current sinusoidal, a switch turn-off loss and a reverse recovery on diode can be reduced. Therefore, the proposed converter is promising for high step-up applications with high efficiency Index Terms Boost converter and voltage-doubler. I. INTRODUCTION For a battery powered system, electric vehicles, fuel cell system, and photovoltaic systems, where low voltage sources need to be converted to a high voltage of output, non-isolated high step-up conversion techniques find increasing necessities. A classical boost converter is generally used for its simple structure and continuous input current. However, it is hard to satisfy both high voltage conversion ratio and high efficiency at the same time with a plain boost converter. In high output voltage applications, moreover, high voltage stress on switch and diode degrades the performance of devices, causing a severe hard switching loss, a conduction loss, and a reverse recovery problem []-[8]. To relieve abovementioned limitations in high step-up applications, various types of step-up techniques can be applied [4]-[4]. A coupled-inductor boost converter is a favorable candidate for its simple structure, however an input current ripple is large and an auxiliary circuit is required to suppress the switch voltage spike [5]-[9]. A voltage multiplier cell or a switch-capacitor circuit can be useful to raise a stepup gain in collaboration with classical topologies [0]-[]. As the output voltage is increased, however, the number of stage is increased, requiring more capacitors and diodes. Besides, a current snubber is required to reduce the reverse recovery on diode. To raise step-up gain of a boost converter further in nonisolated applications, the alternative structure, which combines a boost converter with an auxiliary circuit in series, can be considered as shown in Fig. [5],[6]. Proper selection of an auxiliary module can give many advantages such as high step-up capability, design flexibility, and distributed voltage stress. In this paper, as an auxiliary stepup circuit, the voltage-doubler is integrated with a boost converter as shown in Fig. [7]. The voltage stresses on the diodes D o and D o3 in the volage-doubler are clamped to its Fig.. Boost converter with auxiliary step-up circuit. Fig.. Proposed boost converter integrated with voltage-doubler. Fig. 3. Key waveforms in BR region /0/$ IEEE 80

2 (a) (b) (c) (d) (e) (f) Fig. 4. Topological states in BR region. (a) Mode [ t 0 ~ t ]. (b) Mode [ t ~ t ]. (c) Mode 3 [ t ~ t 3 ]. (d) Mode 4 [ t 3 ~ t 4 ]. (e) Mode 5 [ t 4 ~ t 5 ]. (f) Mode 6 [ t 5 ~ t 6 ]. output, o + o3, therefore it is inherently suitable for high voltage application. The interface between the boost converter and the voltagedoubler is accomplished by the transformer, which also contribute to a step-up gain by the turn ratio n. ince a square voltage waveform, i.e., AC voltage, is applied across the switch Q, the transformer can be inserted in parallel with Q. Then, C R is inserted into the primary side of the transformer to make up for a flux-balance of the transformer. Thereby, the voltage-doubler is coupled with the boost converter by sharing the common switch. That is, with a switching action of Q, both the boost converter and the voltage-doubler are operated. Moreover, the leakage inductance of the transformer L lkg and C R constitute a resonant tank. ince the resonant operation between L lkg and C R shapes the current sinusoidal during Q on-state, a switch turn-off loss and a reverse recovery on diode can be reduced. As a result, the proposed converter is promising for high step-up isolated applications with high efficiency. II. OPERATION PRINCIPLE The proposed converter combines the boost converter and the voltage-doubler, with a common switching function of Q, employing a pulse-width modulation (PWM). ince the voltage-douber utilize the resonant operation between L lkg and C R, its operation can be divided into two regions according to the relationship between the resonant period T R in () and duty cycle D. That is, the above-resonant (AR) region [ T R / > DT ] and the below-resonant (BR) region [ T R / < DT ], in a similar way to conventional resonant converters. The detailed operation is presented as follows. T = p L C () R lkg R A. BR region ( T R / < DT ) The key waveform and the topological states in BR region are shown in Fig. 3 and 4, respectively. Mode [ t 0 ~ t ] : Q is on-state and is applied to the boost inductor L B. The boost inductor current I Lb flows through Q and is increased linearly. At the same time, the voltage-doubler is operated with a common switching action of Q. The powering path from C R to the lower output o of the voltage-doubler is formed through the transformer, Q, and D o, as presented by the dotted line. L lkg and C R constitute a resonant tank and derive a powering current with a sinusoidal shape. The resonant capacitor voltage Cr is decreased. The switch current I Q comprises I Lb and the resonant current of the voltage-doubler as well. ince D o is turned-off with very slow slope of I Do, a reverse recovery would be minimized. D o3 is blocked by o + o3. Mode [ t ~ t ] : ince T R / is shorter than switch oninterval DT, the resonant operation is finished at t before Q is turn-off. Therefore, only I Lb flows through Q and the resonant current does not increases a turn-off current. That is, it does not affect turn-off loss at all. ince no current flows through C R, Cr keeps its value. Mode 3 [ t ~ t 3 ] : Q is turned-off at t and I Lb flow through D o. Meanwhile, the voltage-doubler is starting to be conducted in the opposite direction. The resonant operation 8

3 Transformer turn ratio n Fig. 7. Transformer turn ratio n according to a variation of M. Fig. 5. Key waveforms in AR region. (a) (b) Fig. 6. Topological states in AR region. (a) t 0 ~ t. (b) t ~ t 3. path from the output of the boost converter, o, to the upper output of the voltage-doubler, o3, is formed through D o, the transformer and D o3, as presented by the dotted line. Therefore, by the resonant operation between L lkg and C R, I Do3 is increased. ince the resonant current flows through D o in the opposite direction to I Lb, I Do is decreased accordingly. D o is blocked by o + o3. Mode 4 [ t 3 ~ t 4 ] : I Do reaches zero at t 3, therefore the entire of I Lb flows through the transformer and D o3 of the voltage-doubler. The I Lb charges C R, increasing Cr linearly. D o is blocked. Mode 5 [ t 4 ~ t 5 ] : At t 4, Cr is increased enough to conduct D o again. In this mode, as contrary to mode 3, the powering path from o3 to o, is formed through D o3, the transformer and D o. Therefore, I Do is increased and I Do3 is decreased by the resonant operation between L lkg and C R. Here, a reverser recovery of D o3 can be reduced because of the slow slope of I Do3. Mode 6 [ t 5 ~ t 6 ] : I Do3 reaches zero at t 5 and the entire of I Lb flow through D o. ince no current flows through C R, Cr keeps its value. As D is increased, mode 5 and mode 6 fade gradually and disappear finally. B. AR region ( T R / > DT ) The operation in AR region is similar to that of BR region except that mode of BR region, where I Lb flows solely through Q, is ignored since T R / is longer than DT. The key waveform and the topological states in AR region are shown in Fig. 5 and 6, respectively. ince some topological states are the same to those of BR region, only different topological states, the intervals t 0 ~ t and t ~ t 3, are presented in Fig. 6. The topological states of the intervals t ~ t, t 3 ~ t 4, and t 4 ~ t 5 in AR region are corresponding to Figs. 4(a), 4(c), and 4(d), respectively. In AR region, a switch current at the turn-off instant of t comprises I Lb and I lkg as well, therefore a turn-off loss can be increased compared with that in BR region where only I Lb flows through at the switch turn-off instant. III. ANALYI AND CHARACTERITIC A. Input-Output oltage Gain For the sake of analysis, assuming the ripple of Cr is ignored and using a flux-balance on the boost inductor and the transformer, several voltage equations are obtained as follows. 8

4 witch rms current normalized by Io M = M = 0 M = 8.3 M = 7 M = 5 M = 3 M = (a) Duty Ratio (D) n = 3.5 (b) Fig. 8. (a) witch peak current stress and (b) switch rms current according to a function of M. o o o3 O = () - D = n (3) nd = (4) - D + n = (5) - D Cr _ avg = (6) The o is the same to the output voltage of a classical boost converter and the voltage-doubler provides n times higher voltage, n o ( = o + o3 ). Fig. 7 shows required turn ratio n according to the variation of D and the inputoutput voltage conversion ratio M. When D becomes zero, the voltage-doubler does not operated and I Lb flows through D o, D o, and D o3. That is, o and o3 become zero and O follows like that of a conventional boost converter. B. oltage and Current tress on Device In the boost converter, voltage stresses on Q and D o are o, i.e., /(-D). In the voltage-doubler, the voltage stresses on D o and D o3 are o + o3, i.e., n /(-D). That is, the Fig. 9. Transformer turn ratio n according to a variation of M. voltage stress on the voltage-doubler is n times higher compared with that on the boost converter. ince the voltagedoubler provides n/(+n) of the output voltage, the transformer handles n/(+n) of the total power accordingly. ince C o, C o, and C o3 are connected in series, an average current of each I Do, I Do, and I Do3 is the same to I O. The peak current of I Do is the same to the turn-off current of I Q. The peak current of I Do3 is similar to that of I Lb reflected to the transformer secondary. Assuming DT T R /, the peak current stress on D o and Q can be expressed as in (7) and (8), respectively. Fig. 8 shows the peak current stress and rms current of Q according to the variation of D and M. In the same M, as D is increased, the current stress in decreased. I Do _ peak pt IO p IO»» (7) T D R ( ) np T I ìm p + D -p D -p ü I» I +» í ý I î þ O Q _ peak in _ avg O TR D T R æ np T I ö O IQ _ rms» sin ( w ) 0 Rt Iin _ avg dt T ò ç + è TR ø (9) { ( 6 4 ) 9 }» IO M - M + + M - M D - M D 8D C. ZC on Diodes The diode currents in the voltage-doubler always flow through L lkg, which provides a current snubbing effect, therefore a reverse recovery on D o and D o3 can always be reduced. Especially in BR region, where the half-period resonant operation between L lkg and C R is ensured, a zerocurrent-switching (ZC) is achieved on D o minimizing a reverse recovery. In the mode analysis of BR region, during the switch offstate, I Do is decreased to zero and then is increased again. This operation is also caused by the resonant operation between L lkg and C R, therefore it depends on T R. As T R and DT get smaller, in both BR and AR regions, there is more (8) 83

5 To utilize a 00 switch with a sufficient margin, o is selected about to 50 and the output of the voltage-doubler, o + o3, is selected about to 50 accordingly, which allows a use of 00 rating diodes for the voltage-doubler. Therefore, n is designed to 3.5, which make the operating duty cycle change from 0.33 to 0.6 corresponding to 8 ~ 30 input variation as shown in Fig. 9. Fig. 0. Area-product A P of transformer according to variation of. Fig.. Resonant current waveforms according to a variation of T R. chance to re-increase of I Do. In this case, when Q is turned-on, an abrupt change in I Do is occurred to cause a reverse recovery. Unless I Do is increased again, a reverse recovery on D o would not be occurred. I. DEIGN CONIDERATION To illustrate the design procedure, 8 ~ 30 input, 00 output, 60 W prototype converter is presented. The required input-output voltage gain M is varied from 6.7 ( = 00/30 ), for the maximum input 30, to. ( = 00/8 ) for the minimum input 8. The nominal input voltage is 4, of which the required gain M is 8.3 ( = 00/4 ). A. Transformer Turn Ratio n and Duty Cycle In the proposed converter design, the selection of a switch, which is rather burdened by the sum of the boost inductor current and the resonant current, is primarily considered in terms of cost and an efficiency. As presented in (9) and Fig. 8(b), for the same M, an rms value of switch current is slowly decreased as a duty cycle is increased. On the other hand, a switch voltage stress, o = /(-D), is decreased with a decrease of the duty cycle, which lead to a use of lower voltage switch having a smaller on-resistance. However, a smaller duty cycle results in a larger turn ratio n as show in Fig. 7, which increases a voltage stress of the voltage-doubler. Therefore, a duty cycle should be selected to accommodate as low a voltage stress of switch as possible while not increasing a burden of the voltage-doubler too much. B. Inductor and Transformer The design of the boost inductor is the same as those of conventional ones. Considering a current ripple to be 5 % of the input current 6.7 A, L B is designed as 0 uh [8]. Normally, the area-product A P method can be used to predict the size of the magnetic core [8]. The A P represents the product between a cross-section area and the window area of the magnetic core. In the case of the proposed converter, the A P of one transformer can be obtained as in (0), where K u : the window utilization factor, J : the current density, and B max : the maximum flux density. Considering K u to be 0.3, J to be 300 A/cm, B max to be 0. T, I O to be 0.8 A, and F to be 00 khz, the A P of the transformer according to the function of is illustrated in Fig. 0, where the dot represents the case of n = 3.5. The A P is varied according to a change in the and the maximum A P is 0.73 cm 4 in case of = 4. ( - - ) p D O ( D) IO AP = + (0) B F K J 8D - D max C. Resonant Tank u Fig. shows the current waveform according to T R. In AR region, the switch turn-off loss is increased. On the other hand, in BR region, the switch turn-off loss is reduced and D o achieves a zero-current-switching (ZC) turn-off that minimizes the reverse recovery of the diode. However, the current stress and conduction loss of the devices are rather increased. Therefore, T R can be designed around the midpoint, T R / = DT, to achieve a ZC of the diode while minimizing the switch turn-off loss. Therefore, once L lkg is determined from the fabricated transformer, L lkg is set as it is and C R can be selected as in (). C R = D T () p Llkg. EXPERIMENTAL REULT To verify the proposed converter, the prototype is implemented. The specification and design parameters obtained from the design example are presented in Table I. Fig. shows the experimental waveforms at the nominal input 4 with full load condition. The duty cycle is about 0.5 and is similar to T R /. The resonant operation between L lkg and C R shapes I lkg sinusoidal, resulting in a reduced switch turn-off current. That is, although the peak switch 84

6 (a) (b) (c) Fig.. Experimental waveforms at = 4 with full load condition. (a) (b) (c) Fig. 3. Experimental waveforms at = 8 with full load condition. (a) (b) (c) Fig. 4. Experimental waveforms at = 30 with full load condition. current exceed 6 A, the turn-off current is only under 9 A. Moreover, both D o and D o3 achieve ZC turn-off, which alleviate the reverser recovery. The boost converter output o is about 50, therefore the voltage stresses on Q and D o is under 00 including voltage spike by parasitic inductances, which allows the use of a chottky diode for D o. The voltage stresses on D o and D o3 are clamped to o + o3, about 50. Fig. 3 and 4 show the experimental waveforms at = 8 and = 30, respectively. In case of = 8, the duty cycle is increased and the circuit is operated in BR region, i.e., DT > T R /. In case of = 30, the circuit is operated in AR region, i.e., DT < T R /. In both cases, the reverse recoveries on D o and D o3 are sufficiently suppressed by the current snubbing effect of L lkg. Fig. 4 shows the efficiency curves with respect to the variation of. ince the resonant tank is designed to satisfy 85

7 the condition DT = T R / at = 4, the proposed circuit shows the high efficiency over 93 % at this point along a wide load range. In the case of = 8, an increased conduction loss degrades the efficiency. On the other hand, in the case of = 30, the operation in AR region increases the switch turn-off loss, though a conduction loss is decreased. Consequently, it is noted that the efficiency is highly affected by the resonant tank design. Table I Experimental parameters Part alue Input voltage 8 ~ 30 ( nominal input : 4 ) Output voltage O 00 Output Power P O 60 W ( I O = 0.8 A ) witching frequency F 00 khz Boost inductor L B Inductance = 0 uh, high flux, outer diameter = 7 mm, μ = 5 Transformer turn ratio n 3.5, core : EER8/6/, A P = 0.9cm 4 Transformer leakage inductance L lkg uh Balance capacitance C R uf witch Q IRF540A ( R ds = 0.05 Ω, D = 00 ) D o: 6CTQ00 Diodes ( F = 0.58, RRM = 00 ), D o, D o3: 0ETF0 ( F =., RRM = 00 ) I. CONCLUION To raise step-up gain of a boost converter further in nonisolated applications, a voltage-doubler, which is inherently suitable for high voltage applications, is integrated with a boost converter in series as an auxiliary step-up circuit. The voltage-doubler provides an additional step-up ratio on top of the gain of the boost converter and distributes voltage stresses on devices as well. The interface between the boost converter and the voltage-doubler is accomplished by the transformer, which also contribute to a step-up gain by the turn ratio n. The transformer leakage inductance and the balancing capacitor constitute the resonant tank and its resonant operation shapes the current sinusoidal, resulting in a reduced switch turn-off loss and a reverse recovery on diode. Therefore, the proposed converter is promising for high stepup applications with high efficiency. It is also noted that other type of rectifier can also be integrated with a boost converter, being interfaced by a transformer and a balancing capacitor, in the same way to the voltage-doubler. REFERENCE [] R. W. Erickson and D. Maksimovic, Fundamentals of Power electronics, nd Ed., John Wiley, New York, UA, 950, pp [] K. M. mith and K. M. medly, Properties and systhesis of passive lossless soft-switching PWM converters, IEEE Trans. Power Electronics, vol. 4, no. 5, pp , ep [3] M. M. Jovanovic and Y. Jang, tate-of-the-art, single-phase, active power-factor-correction techniques for high-power applications an Fig. 5. Measured efficiency. overview, IEEE Trans. Industrial Electronics, vol. 5, no. 3, pp , Jun [4] F. L. Luo and H. Ye, Positive output cascade boost converters, IEE Proc. Electr. Power Appl., vol. 5, no. 5, sep [5] T.-F. Wu, Y.-. Lai, J.-C. Hung, and Y.-M. Chen, Boost converter with coupled inductors and buck-boost type of active clamp. IEEE Trans. Industrial Electronics, vol. 55, no., pp. 54-6, Jan [6] Q. Zhao and F. C. Lee, High-efficiency, high step-up dc-dc converters. IEEE Trans. Power Electronics, vol. 8, no., pp , Jan [7] K. C. Tseng and T. J. Liang, Novel high-efficiency step-up converter, IEE Proc. Electr. Power Appl., vol. 5, no., pp. 8-90, Mar [8] R.-J Wai and R.-Y. Duan, High step-up converter with coupledinductor, IEEE Trans. Power Electronics, vol. 0, no. 5, pp , ep. 005 [9] W. Li and X. He, A family of interleaved DC-DC converters deduced from a basic cell winding-cross-coupled inductors (WCCIs) for high step-up or step-down converters, IEEE Trans. Power Electronics, vol. 3, no. 4, pp , Jul [0] H. Ye and F. L. Luo, Positive output super-lift converters, IEEE Trans. Power Electronics, vol. 8, no., pp. 05-3, Jan [] E. H. Ismail, M. A. Al-affar, A. J. abzali, and A. A. Fardoun, A family of single-switch PWM converters with high step-up conversion ratio, IEEE Trans. Circuit and ystem I, vol. 55, no. 4, pp. 59-7, May 008. [] M. Prudente et al, oltage multiplier cells applied to non-isolated DC-DC converters, IEEE Trans. Power Electronics, vol. 3, no., pp , Mar [3] W. C. P. de Aragao Filho and I. Barbi, A comparison between two current-fed push-pull DC-DC converters analysis, design and experimentation, INTELEC, 996, pp [4] Y. Jang and M. M. Javanovic, New two-inductor boost converter with auxiliary transformer, IEEE Trans. Power Electronics, vol. 9, no., pp , Jan [5] K.-B. Park, H.-W. eong, H.-. Kim, G.-W. Moon, and M.-J. Youn, "Integrated boost-sepic converter for high step-up applications," in Proc. IEEE PEC, 008, pp [6] K.-B. Park, C.-E. Kim, G.-W. Moon, and M.-J. Youn, "Nonisolated high step-up converter based on boost integrated halfbridge converter," in Proc. INTELEC, 009, PC3-3. [7] K.-B. Park, C.-E. Kim, G.-W. Moon, and M.-J. Youn, PWM resonant single-switch isolated converter, IEEE Trans. Power Electronics, vol. 4, no. 8, pp , Aug [8] L. H. Dixon, Transformer and inductor design for optimum circuit performance, in Proc. Unitrode Power upply Design eminar,