FOR battery-powered systems, electric vehicles, fuel cell

Transcription

1 1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 High Step-up Boost Converter Integrated With a Transformer-Assisted Auxiliary Circuit Employing Quasi-Resonant Operation Ki-Bum Park, Member, IEEE, Gun-Woo Moon, Member, IEEE, and Myung-Joong Youn, Senior Member, IEEE Abstract Stacking an auxiliary step-up circuit on top of a boost converter is one of the most attractive structures for nonisolated high step-up applications. In this paper, in order to avoid the large input current ripple of coupled-inductor-based circuits, an auxiliary step-up circuit is integrated via an additional transformer and its balancing capacitor. A voltage-doubler is adopted as an auxiliary step-up circuit, which is inherently suitable for high-voltage applications due to its simple structure and low-voltage stress. Moreover, the transformer leakage inductor and the balancing capacitor constitute a resonant tank so that the quasi-resonant operation makes the current sinusoidal. As a result, a reduced switch turn- OFF loss and reverse recovery of the diode can be expected. The proposed converter is verified with a 24 V input, 160 W 200 V output prototype. Index Terms Coupled-inductor, high step-up converter, quasiresonant voltage-doubler. I. INTRODUCTION FOR battery-powered systems, electric vehicles, fuel cell systems, and photovoltaic systems, where low-voltage sources need to be converted into high voltages, the demand for nonisolated high step-up dc dc conversion techniques are gradually increasing [1] [6]. A classic boost converter is widely used due to its simple structure and its continuous input current. However, it is hard to achieve a high-voltage conversion ratio with just a plain boost converter, since the parasitic resistance of the circuit causes a severe loss as the duty cycle is increased which limits the step-up gain [7]. Especially in high output voltage applications, high-voltage stress on switches and diodes degrades the performance of devices, causing a severe hard switching loss, a conduction loss, and a reverse recovery problem [8] [10]. Moreover, an increased duty cycle to obtain a high step-up gain has a detrimental effect on the dynamic performance of a boost converter [7]. Therefore, to relieve the Manuscript received June 23, 2011; accepted September 14, Date of current version February 20, This paper was presented in part at the IEEE ECCE, 2010, under the title High step-up boost converter integrated with voltage-doubler. Recommended for publication by Associate Editor M. Vitelli. K.-B. Park is with the Power Electronic Systems Group, ABB Corporate Research, Segelhofstrasse 1K 5405 Baden-Daettwil, Switzerland ( ki-bum.park@ch.abb.com). G.-W. Moon and M.-J. Youn are with the KAIST, Electrical Engineering, 373-1, Kuseong-dong, Yuseong-gu, Daejeon , Republic of Korea ( gwmoon@ee.kaist.ac.kr, mmyoun@ee.kaist.ac.kr). Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TPEL abovementioned limitations on boost converters in high step-up applications, it is necessary to reduce the operating duty cycle and distribute the voltage stresses across the devices allowing for the use of low-voltage high-performance devices. Till now, various types of step-up techniques based on a boost converter have been developed [11] [29]. Cascading a boost converter is a simple way to achieve a high step-up gain by employing more components. However, an increased number of powering processes degrades the efficiency and the burden of a high-voltage stress still remains [11], [12]. Coupled-inductor boost converters are favorable candidates due to their simple structure. However, their input current ripple is large due to a coupled inductor effect and an additional voltage clamp circuit for the switch and diode is required [13] [20]. A voltagemultiplier cell or a switch-capacitor circuit can also be useful to raise the step-up gain in collaboration with basic topologies [21] [24]. However, when a higher output voltage is required, the number of step-up stages is increased, which requires more components. In addition, a current snubber is also required to suppress the excessive peak current for charging the capacitors. In terms of isolated type converters, current-fed converters easily offer a high step-up gain through the turn ratio of the transformer, which makes them inherently suitable for high step-up applications [25], [26]. As a price for utilizing a transformer, an additional voltage snubber is needed to limit the switch voltage spikes caused by the transformer leakage inductance. In addition, an auxiliary circuit for operation below 0.5 duty cycles is required. Among the abovementioned high step-up circuits, a coupledinductor-based circuit seems to be one of the most suitable candidates in low-to-medium power applications due to its simple structure. In order to remove the large input current ripple and improve the performance of this circuit further, an alternative structure, which is based on a boost converter integrated with a transformer assisted auxiliary step-up circuit, is investigated in this paper. Furthermore, a quasi-resonant operation is adopted to reduce the switch turn-off loss caused by the large switch current in a high step-up converter, without additional circuitry. II. PROPOSED HIGH STEP-UP CONVERTER In order to further raise the step-up gain of a boost converter, alternative structures which combines a boost converter with an auxiliary step-up circuit in series have been developed [15], [17], [27] [29]. Proper selection of an auxiliary module can offer advantages such as high step-up capability, design /$ IEEE

2 PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1975 Fig. 2. Proposed high step-up boost converter integrated with voltage-doubler as auxiliary step-up circuit, employing quasi-resonant operation. Fig. 1. High step-up converter employing auxiliary step-up circuit on top of boost converter. (a) Conventional coupled-inductor-assisted auxiliary circuit. (b) Proposed transformer-assisted auxiliary circuit. flexibility, and distributed voltage stress across devices which allows for low voltage high performance devices. Among them, the coupled-inductor-assisted auxiliary step-up circuit shown in Fig. 1(a) is promising due to its simple structure, having only one switch and an easy high step-up capability utilizing the turn ratio of the coupled-inductor [15], [17]. However, these structures also suffer from some side effects that come from the coupled-inductor as follows. As the output voltage of the auxiliary circuit is increased, more power is driven by the coupled-inductor to the secondary side. Since the input current consists of the boost inductor current, I Lb, and the reflected auxiliary circuit current, ni aux,the input current ripple can get considerably larger ( n is the turn ratio of the coupled-inductor). That is, as n increases, the input current ripple increases, and more input filtering might be required, which has a detrimental effect on the total efficiency. Moreover, since this input current flows through the switch, a large input current ripple could cause a large switch turn-off loss. In terms of the size of the coupled-inductor in high stepup applications, it can be rather large when compared with a plain inductor, since it requires many secondary turns and a large ac-current being superimposed to a dc-current flows through the windings. That is, the coupled-inductor needs to be designed like a flyback transformer [30]. So that the coupledinductor size is not increased too much, a smaller inductance design is one option. However, this increases the peak inductor current in return, which results in an even higher switch turn- OFF loss. As the input current increases, these drawbacks could impose a greater burden on the magnetic component design and efficiency. In order to relieve aforementioned drawbacks of coupledinductor-based circuits, a transformer-assisted auxiliary high step-up circuit, which has a continuous input current, is introduced in this paper as shown in Fig. 1(b). In addition, a quasi-resonant operation utilizing the transformer leakage inductance is adopted in part [31] [34], which provides partial soft-switching characteristics for both the switches and the diodes. Fig. 2 shows the proposed high step-up converter, and its main features are as follows. A. Voltage-Doubler as an Auxiliary Step-up Circuit Various types of rectifiers can be adopted as an auxiliary stepup circuit. Among them, a voltage-doubler is inherently suitable for high-voltage applications due to its simple structure, which consists of two diodes and two capacitors, and due to its lowvoltage stress on devices, which is clamped to the output voltage of the voltage-doubler. Therefore, they are widely adopted in many topologies as a part of the circuit [17], [19], [27], [29], [31], [35], [37]. For the proposed converter, a voltage-doubler is also employed as an auxiliary step-up circuit, but it is integrated with a boost converter in a different manner that allows for the following distinctive properties. B. Separated Boost-Inductor and Transformer In the proposed circuit, unlike a coupled-inductor-based circuit, the interface between the boost converter and the voltagedoubler is accomplished by an additional transformer, which also contributes to the step-up gain by means of the turn ratio n. Since a square voltage waveform, i.e., an ac voltage, is applied across the switch Q, the transformer can be inserted in parallel with Q. Then, the balancing capacitor, C R, is inserted into the primary side of the transformer to make up for the flux-balance of the transformer. Thereby, the voltage-doubler is coupled with the boost converter by sharing a common switch. Therefore, by the switching action of Q, both the boost converter and the voltage-doubler are operated at the same time. Compared with the large input current ripple of the coupledinductor assisted boost converters in [13] [17], the proposed converter maintains a continuous input current of I Lb, which

3 1976 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 could require less input filtering. In other words, the proposed circuit decouples the large ac-current from the input side to the transformer at the price of additional components, i.e., the transformer and the balancing capacitor. As a result, the input current ripple becomes continuous, and each inductor and transformer can be designed optimally compared with the coupledinductor [30]. C. Quasi-Resonant Operation Although the large input current ripple is removed, it is transferred to the transformer and this large current ripple still flows through the switch in addition to the boost inductor current I Lb, which could result in a high turn-off loss. To relieve this, a quasi-resonant operation, which considerably reduces the switch turn-off current, is adopted as follows. In a separated transformer structure followed by the voltagedoubler, the leakage inductance of the transformer, L lkg, and the balancing capacitor, C R, can constitute a resonant tank similar to that of the series resonant type converters in [31] [34]. The quasi-resonant operation between L lkg and C R makes the current on the transformer primary side and the voltage-doubler sinusoidal during the switch-on state. Owing to this sinusoidal current, the switch turn-off current can be considerably reduced resulting in less turn-off loss [36]. Moreover, since the sinusoidal current guarantees slow di/dt, the reverse recovery on the diode of the voltage-doubler can also be alleviated. III. OPERATION PRINCIPLES The proposed converter combines the operations of a boost converter and a voltage-doubler, with the common switching function of Q, employing pulse-width modulation (PWM). Since the voltage-doubler utilizes the quasi-resonant operation between L lkg and C R, its operation can be divided into two regions according to the relationship between the resonant period, T R, in (1) and the duty cycle, D. That is, the above-resonant (AR) region, where T R /2 > DT S, and the below-resonant (BR) region, where T R /2 < DT S, which is similar to conventional resonant converters [31] [34]. The detailed operation is presented as follows.. T R =2π L lkg C R (1) A. Below-Resonant Region (T R /2 < DT S ) The key waveform and the topological states in the BR region are shown in Figs. 3 and 4, respectively. Mode 1 [t 0 t 1 ]: QisintheON-state and V S is applied to the boost inductor L B. The boost inductor current, I Lb, flows through Q and is increased linearly. At the same time, the voltage-doubler is operated with the common switching action of Q. The powering path from C R to the lower output of the voltage-doubler, V o 2, is formed through the transformer, Q, and D o 2, as represented by the dotted line. The L lkg and C R constitute a resonant tank and derive a powering current with a sinusoidal shape. The resonant capacitor voltage, V Cr,is Fig. 3. Key waveforms in below-resonant region (T R /2 < DT S ). decreased. The switch current, I Q, comprises I Lb and the transformer primary current of the voltage-doubler, I lkg. Since D o 2 is turned-off with a very slow slope of I Do2, the reverse recovery can be minimized. D o3 is blocked by V o 2 + V o 3. Mode2[t 1 t 2 ]: Since T R /2 is shorter than the switch oninterval DT S, the resonant operation is finished at t 1 before Q is turned-off. OnlyI Lb flows through Q. Therefore, the switch turn-off loss is only affected by I Lb. Since no current flows through C R, V Cr keeps its value during this interval.emphasis Mode 3 [t 2 t 3 ]:Q is turned-off at t 2, then I Lb flows through D o1. Meanwhile, the voltage-doubler starts to conduct in the opposite direction. That is, the resonant powering path from the output of the boost converter, V o 1, to the upper output of the voltage-doubler, V o 3, is formed through D o 1, the transformer and D o 3, as represented by the dotted line. Therefore, by the resonant operation between L lkg and C R, I Do3 is increased. Since the resonant current flows through D o 1 in the opposite direction of I Lb, I Do1 is decreased accordingly. D o 2 is blocked by V o 2 + V o 3. Mode4[t 3 t 4 ]: I Do1 reaches zero at t 3. Then all of I Lb flows through the transformer and D o 3 of the voltage-doubler. The I Lb charges C R, increasing V Cr linearly. The D o 1 is blocked by V o 1 -V Cr -V o 3 /n, which is slowly decreased. The same amount of change can be observed in V Q. Mode 5 [t 4 t 5 ]: At t 4, V Cr is increased enough to conduct D o 1 again. In this mode, unlike mode 3, the powering path from V o 3 to V o 1, is formed through D o 3, the transformer and D o 1 by

4 PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1977 Fig. 4. Topological states of below-resonant region. (a) Mode 1 [t 0 t 1 ]. (b) Mode 2 [t 1 t 2 ]. (c) Mode 3 [t 2 t 3 ]. (d) Mode 4 [t 3 t 4 ]. (e) Mode 5 [t 4 t 5 ]. (f) Mode 6 [t 5 t 6 ]. the resonant operation between L lkg and C R. In other words, the flow of I Lb is shifted slowly in the resonant way from I Do3 to I Do1. Here, the reverser recovery on D o3 can be reduced by the slow slope of I Do3. Mode 6 [t 5 t 6 ]: I Do3 reaches zero at t 5. Then all of I Lb flow through D o 1. Since there is no current flowing through C R, V Cr keeps its value during this mode. As D is increased, modes 5 and 6 gradually fade and disappear. B. Above-Resonant Region (T R /2 > DT S ) Operation in the AR region is similar to that of the BR region except for mode 2 of the BR region, where I Lb flowing solely through Q, is ignored since T R /2 is longer than DT S.Thekey waveform and the topological states in the AR region are shown in Figs. 5 and 6, respectively. Since some of the topological states are the same as those of the BR region, only the different topological states, the intervals t 0 t 1 and t 2 t 3, are presented in Fig. 6. The topological states of the intervals t 1 t 2,t 3 t 4, and t 4 t 5 in the AR region correspond to Fig. 4(a), 4(c), and 4(d), respectively. It is noted that, in the AR region, since Q is turned-off while still powering through the transformer, the switch current at the turn-off instant of t 2 comprises I Lb and I lkg. Therefore, the turn-off loss can be increased compared with that in the BR region where only I Lb flows at the switch turn-off instant. Fig. 5. Key waveforms in above-resonanat region (T r /2 > DT s ).

5 1978 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 Fig. 6. Topological states of above-resonant region. (a) t 0 t 1.(b)t 2 t 3. on D o 2 and D o 3 are V o 2 +V o 3, i.e., nv S /(1-D). That is, the voltage stress on the voltage-doubler is n times higher than that of the boost converter. Since the voltage-doubler provides n/(1 + n) of the total output voltage, the transformer handles n/(1+n) of the total power accordingly. The voltage ripple and the peak voltage stress of V Cr are expressed as (7) and (8), respectively. V Cr rp = ni O T S C R (7) Fig. 7. Transformer turn ratio n according to a variation of M. IV. ANALYSIS AND CHARACTERISTICS A. Input-Output Voltage Gain For the sake of analysis, assuming the ripple of V Cr is ignored and using a flux-balance on the boost inductor and the transformer, the following voltage equations are obtained: V o1 = 1 1 D V S (2) V o2 = nv S (3) V o3 = nd 1 D V S (4) V O = 1+n 1 D V S (5) V Cr avg = V S. (6) The V o 1 is the same as the output voltage of a classical boost converter and the voltage-doubler provides a voltage that is n times higher, nv o 1 ( = V o 2 + V o 3 ). Fig. 7 shows the required turn ratio n according to the variation of D and the inputoutput voltage conversion ratio M. When D becomes zero, the voltage-doubler does not operated and I Lb flows through all of the series-connected diodes, D o 1, D o 2, and D o 3. That is, V o 2 and V o 3 become zero and V O follows V S in the same manner as a conventional boost converter. B. Voltage and Current Stress on the Device In the boost converter, the voltage stresses on Q and D o 1 are V o 1, i.e., V S /(1-D). In the voltage-doubler, the voltage stresses V Cr peak = V Cr avg + V Cr rp 2 = V S + ni O T S 2C R. (8) Since C o 1, C o 2, and C o 3 are connected in series, the average current of I Do1, I Do2, and I Do3 is the same as I O. The peak current of I Do1 is the same as the turn-off current of I Q, which is especially high in the AR region due to the remaining resonant current at the turn-off instant. The peak current of I Do3 is similar to that of I Lb reflected to the transformer secondary. Assuming that DT S T R /2, the peak current stresses on D o 2 and I Q can be expressed as in (9) and (10), respectively. I Q rms I Do2 peak πt S I O πi O T R 2D I Q peak I in avg + nπt S I O 1 T S T R (9) { } M (π +2D πd) π I O (10) 2D T R 2 0 ( ) 2 nπts I O sin (ω R t)+i in avg dt T R I O 1 8D {M 2 2M +1+(16M 2 14M) D 9M 2 D 2 }. C. ZCS on the Diodes (11) The diode currents in the voltage-doubler always flow through L lkg that provides a current snubbing effect. Therefore, the reverse recovery on D o 2 and D o 3 can always be reduced by a slow di/dt. The steepest slopes of I Do2 and I Do3 can occur at the switching transition in the AR region as shown in Fig. 5. Equations (12) and (13) represent the decreasing di/dt of I Do2 and I Do3, respectively, which are the same. In order to sufficiently

6 PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1979 Fig. 8. (a) Switch peak current stress and (b) switch rms current according to a function of M. Fig. 9. Operating duty cycle D according to variations of V S and n. Fig. 10. Area-product, A P, of transformer according to variation of V S. reduce the reverse recovery, di/dt should be at least less than 100 A/μs[9]. di Do2 dt di Do3 dt = V CR(t 2 )+V o3 /n = 1 ( 1 nl lkg nl lkg 1 D V S I ) O C R F S = V o1 + V o2 /n V CR (t 5 ) nl lkg = 1 ( 1 nl lkg 1 D V S I O C R F S (12) ). (13) On the other hand, in the BR region, where the half-period resonant operation between L lkg and C R provides an extremely slow slope on I Do2, zero-current-switching (ZCS) can be achieved on D o 2, minimizing the reverse recovery. In the case of D o 1, during the switch-off state, I Do1 is decreased to zero and then it is increased again as described in the mode analysis in the BR region of Section II. This operation is also caused by the resonant operation between L lkg and C R, therefore, it depends on T R and DT S. In both the BR and AR regions, as T R and DT S get smaller, there is a greater chance of reincreasing I Do1. In this case, when Q is turned-on, an abrupt change in I Do1 occurs causing a reverse recovery. Unless I Do1 is increased again, a reverse recovery on D o 1 will not occur. Fig. 11. Resonant current waveforms according to a variation of T R. V. DESIGN CONSIDERATION In general, the power level of a certain topology is mainly determined by the component count and rating. That is, the more switches are employed, the higher the power level. The proposed converter utilizes only one switch, therefore, it is expected to be suitable for low-to-medium power applications. To illustrate the design procedure for the proposed circuit, an V input, 200 V output, 160 W prototype converter is presented. The required input-output voltage gain M is varied from 6.7 ( = 200/30), for a maximum input of 30 V, to 11.1 ( = 200/18) for a minimum input of 18 V. The nominal input voltage is 24 V, for which the required gain M is 8.3 ( = 200/24).

7 1980 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 TABLE I EXPERIMENTAL PARAMETERS Fig. 12. Experimental waveforms at V S = 24 V with full load condition. A. Transformer Turn Ratio and Duty Cycle In the proposed converter design, the selection of a switch, which is burdened by the sum of the boost inductor current and the resonant current, is primarily considered in terms of cost and efficiency. As presented in (11) and Fig. 8(b), for the same M, the rms value of the switch current is slowly decreased as the duty cycle is increased. On the other hand, the switch voltage stress, V o1 = V S /(1- D), is decreased with a decrease in the duty cycle, which leads to the use of a lower-voltage switch having a smaller on-resistance. However, a smaller duty cycle results in a larger turn ratio n as show in Fig. 7, which increases the voltage stress of the voltage-doubler. Therefore, a duty cycle should be selected to accommodate as low a voltage stress on the switch as possible while not increasing the burden of the voltage-doubler too much. By selecting the transformer turn ratio n = 3.5, the duty cycle varies from in response to the V S change from 30 V

8 PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1981 Fig. 13. Experimental waveforms at V S = 18 V with full load condition. to 18 V, as can be seen in Fig. 9. The ratio of the boost converter output, V o 1, to the voltage-doubler output, V o 2 + V o 3, is always 1:n + 1 regardless of the input variation, as can be seen in (2) (5). When n = 3.5, V o 1 and V o 2 + V o 3 become about 45 V and 155 V, respectively. Therefore, 100 V devices for the boost converter and 200 V devices for the voltage-doubler are available. B. Inductor and Transformer The design of the boost inductor is the same as those of conventional ones. Assuming the current ripple to be 15% of the input current 6.7 A, L B is designed for 120 μh [30]. Since the balancing capacitor, C R, is inserted into the primary side and the voltage-doubler capacitors, C o 2 and C o 3,are located in the secondary side, no dc-current can flow through the transformer by the charge-balance of the capacitors, even though the voltage applied to the transformer is asymmetrical. Therefore, the magnetizing current has no dc-offset, which is beneficial for the transformer design [30]. Normally, the area-product A P method can be used to predict the size of the magnetic core [30]. The A P represents the product between a cross-section area and the window area of the magnetic core. In the case of the proposed converter, the A P of one transformer can be obtained as in (14), where K u is the window utilization factor, J is the current density, and B max is the maximum flux density. Assuming K u to be 0.3, J to be 300 A/cm 2, B max to be 0.1 T, I O to be 0.8 A, and F S to be 100 khz, the A P of the transformer according to the function of V S is illustrated in Fig. 10, where the dot represents the case of n = 3.5. The A P is varied according to a change in V S and the maximum A P is 0.73 cm 4 in the case of V S = 24 V. A P = D (V O (1 D) V S ) I O B max F S K u J C. Resonant Tank π 2 8D D. (14) Fig. 11 shows the current waveform according to T R.Inthe AR region, the switch turn-off loss is increased. On the other hand, in the BR region, the switch turn-off loss is reduced and D o 2 achieves a zero-current-switching (ZCS) turn-off that minimizes the reverse recovery of the diode. However, the current stress and the conduction loss of the devices are increased. Therefore, T R should be designed around the midpoint, T R /2 = DT S, to achieve ZCS of the diode while minimizing the switch turn-off loss and the conduction loss. To be designed at this point, once L lkg is obtained from the fabricated transformer,

9 1982 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 Fig. 14. Experimental waveforms at V S = 30 V with full load condition. L lkg can be set as it is and C R can then be selected as in (15). C R = D2 TS 2 π 2. (15) L lkg With respect to the reverse recovery of the diode, L lkg should also be considered as a current snubber. The minimum L lkg which guarantees di/dt below 100 A/μs is approximately obtained at 150 nh from (10). If the obtained L lkg from the fabricated transformer has a larger inductance than 150 nh, a severe reverse recovery will be relieved. As L lkg is increased, it is more beneficial to reverse recovery. However, a larger L lkg reduces C R to maintain the same T R as presented in (15), which lead to a larger voltage ripple in V Cr in return, as noted in (7). VI. EXPERIMENTAL RESULTS To verify the proposed converter, a 160 W prototype is implemented. The specifications and design parameters obtained from the design example are presented in Table I. Fig. 12 shows the experimental waveforms at a nominal input of 24 V under the full load condition. The duty cycle is about 0.5 and it is similar to T R /2. The resonant operation between L lkg and C R makes I lkg sinusoidal and only the boost inductor current I Lb flows through the switch Q at the turn-off instant, resulting in a reduced switch turn-off loss. That is, although the peak switch current exceeds 16 A, the turn-off current is under 9 A. Moreover, both D o 2 and D o 3 achieve ZCS turn- OFF, which alleviates the reverser recovery. The boost converter output V o 1 is about 50 V. Therefore, the voltage stresses on Q and D o 1 are under 100 V, including the voltage spikes caused by parasitic inductances, which allows for the use of a Schottky diode for D o 1. The voltage stresses on D o 2 and D o 3 are clamped to V o 2 +V o 3, at about 150 V. The input current, i.e., the boost inductor current I Lb, is continuous. Figs. 13 and 14 show the experimental waveforms at V S = 18 V and V S = 30 V, respectively. In the case of V S = 18 V, the duty cycle is increased to regulate V O and the circuit is operated in the BR region of DT S > T R /2. Here, the switch turn-off current is still the same as I Lb. In the case of V S = 30 V, the circuit is operated in the AR region, i.e., DT S < T R /2, and the switch turn-off current is about 12 A despite the fact that I Lb is only about 7 A. In both cases, the reverse recoveries on D o 2 and D o 3 are sufficiently suppressed by the current snubbing effect of L lkg. The voltage stress on the switch in the steadystate is still about 50 V regardless of the input variation. This implies that the switch turn-on loss is rarely affected by an input change. Fig. 15 shows the efficiency curves with respect to the variation of V S. Since the resonant tank is designed to satisfy the condition DT S = T R /2 at V S = 24 V, the proposed circuit shows high efficiency, over 93%, at this point along a wide load range. In the case of V S = 18 V, where the converter is operated in the BR region, the increased conduction loss caused by the increased input current would degrade the efficiency. On the other hand, in the case of V S = 30 V, the operation in the AR region increases the switch turn-off loss even though the conduction loss is decreased by a low input current. Consequently, it is noted that the efficiency of the proposed converter is affected by the

10 PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1983 Fig. 15. Measured efficiency. resonant tank design. That is, the proper selection of a balancing capacitor, C R, can improve the circuit performance at a certain operating point, which in this experiment is the nominal 24 V input, as expected. VII. CONCLUSION A coupled-inductor-assisted auxiliary step-up circuit is an attractive candidate in low-to-medium power applications due to its simple structure and it can easily achieve a high step-up gain by increasing the turn ratio of the coupled-inductor. However, it has large input current ripple, which may require more input filtering, and large switch current could cause a high turn-off loss. In order to remove the large input current ripple, an alternative structure, where the auxiliary step-up circuit is integrated via an additional transformer and its balancing capacitor, is introduced in this paper. 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11 1984 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 [32] B. Yang, F. C. Lee, A. J. Zhang, and G. Huang, LLC resonant converter for front end dc/dc converter, in Proc. IEEE APEC,2002,pp [33] K.-B. Park, C.-E. Kim, G.-W. Moon, and M.-J. Youn, PWM resonant single-switch isolated converter, IEEE Trans. Power Electron., vol. 24, no. 8, pp , Aug [34] D. Fu, F. C. Lee, Y. Qiu, and F. Wang, A novel high-power-density three-level LCC resonant converter with constant-power-factor-control for charging applications, IEEE Trans. Power Electron., vol. 23, no. 5, pp , Sep [35] Y. Jang and M. M. Javanovic, Interleaved boost converter with intrinsic voltage-doubler characteristic for universal-line PFC front end, IEEE Trans. Power Electron., vol. 22, no. 4, pp , Jan [36] L. Balogh, Design and application guide for high speed MOSFET gate drive circuit, in Proc. Unitrode Power Supply Design Seminar,2001,pp [37] H.-L. Do, A zero-voltage-switching dc dc converter with high voltage gain, IEEE Trans. Power Electron., vol. 26, no. 5, pp , May Gun-Woo Moon (S 92 M 00) received the M.S. and Ph.D. degrees in electrical engineering from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, in 1992 and 1996, respectively. He is currently a Professor in the Department of Electrical Engineering, Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea. His research interests include modeling, design and control of power converters, soft-switching power converters, resonant inverters, distributed power systems, power-factor correction, electric drive systems, driver circuits of plasma display panels, and flexible ac transmission systems. Dr. Moon is a member of the Korean Institute of Power Electronics (KIPE), Korean Institute of Electrical Engineers (KIEE), Korea Institute of Telematics and Electronics (KITE), Korea Institute of Illumination Electronics and Industrial Equipment (KIIEIE), and Society for Information Display (SID). Ki-Bum Park (S 07 M 10) was born in Korea, in He received the B.S., M.S., and Ph.D. degrees in electrical engineering from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 2003, 2005, and 2010, respectively. He is currently a Scientist with ABB Corporate Research Center, Baden-Dättwil, Switzerland. His research interests include power converters, multilevel inverter, server power supply, high power density adapter, battery management system, and display driver circuit. Dr. Park received the Second Prize Paper Award from the International Telecommunications Energy Conference (INTELEC) 2009 and the Third Prize Paper Award from the Energy Conversion Congress and Exposition (ECCE)- Asia Myung-Joong Youn (S 74 M 78 SM 98) was born in Seoul, Korea, in He received the B.S. degree in electrical engineering from Seoul National University, Seoul, in 1970, and the M.S. and Ph.D. degrees in electrical engineering from the University of Missouri, Columbia, in 1974 and 1978, respectively. In 1978, he joined the Air-Craft Equipment Division, General Electric Company, Erie, PA, where he was an Individual Contributor on Aerospace Electrical System Engineering. Since 1983, he has been a Professor at the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea. His research activities are in the areas of power electronics and control, which include the drive systems, rotating electrical machine design, and high-performance switching regulators. Dr. Youn is a member of the Institution of Electrical Engnieers, U.K., the Korean Institute of Power Electronics (KIPE), the Korean Institute of Electrical Engineers (KIEE), and the Korea Institute of Telematics and Electronics (KITE).