TLT-5806/RxArch2/1 Receiver Architectures - Part 2 Increasing the role of DSP in receiver front-ends Markku Renfors Department of Communications Engineering Tampere University of Technology, Finland markku.renfors@tut.fi www.cs.tut.fi/tlt Topics: - Classification of DSP-Based Receiver Architectures - Characteristics of Alternative Flexible Wideband Receiver Architectures - DSP for Flexible Receivers - Digital Channel Selection & Down-Conversion Techniques
TLT-5806/RxArch2/2 Classification of DSP-Based Receiver Architectures Location of ADC (A/D-converter) 1. Baseband or low IF 2. IF 3. RF Analog front-end bandwidth - ADC bandwidth 1. Single channel 2. Few channels 3. Frequency slice 4. Service band (e.g., GSM) 5. Frequency band (like 2 GHz range) The bandwidths of the analog front-end and ADC may or may not go hand in hand. Some examples below.
TLT-5806/RxArch2/3 Narrowband front-end Per-channel down-conversion (0, or IF) - Selectivity in analog part, good IF filters needed. - "Normal" frequency sythesizer needed. - No big demands for ADC dynamic range or jitter. - Sampling rate requirements are such that it is enough to attenuate aliasing to the desired band. - Narrowband (baseband or bandpass) ADC (like ΣΔ) can be utilized. RF-stages Mixer IF-stages BP/LP filter ADC LO f IF
TLT-5806/RxArch2/4 Wideband front-end, per-channel downconversion so that desired channel is around a fixed center frequency (0 or IF) - Analog front-end simplified in the sense that highly selective IF filters are not needed. - "Normal" frequency synthesizer needed. - Selectivity in digital part, with fixed center frequency. - High demands for ADC dynamic range and jitter. - Sampling rate requirements are such that it is enough to attenuate aliasing to the desired band. - Narrowband ADC (like ΣΔ) can be utilized. RF-stages Mixer IF-stages BP/LP filter ADC LO f IF B IF
TLT-5806/RxArch2/5 Wideband front-end, wideband (like slice or service band) down-conversion so that desired channel is located in a wider frequency range - Analog front-end simplified in the sense that highly selective IF filters are not needed. - Single or few LO frequencies needed for each service band -> simplified synthesizer; fast frequency hopping becomes feasible. - Selectivity in digital part, with tunable center frequency. - High demands for ADC dynamic range and jitter. - Sampling rate requirements are such that no aliasing into the whole band is allowed. - Wideband ADC (or ΣΔ with tunable center frequency!?) needed. RF-stages Mixer IF-stages BP filter ADC LO B IF
TLT-5806/RxArch2/6 About the Choice Between Lowpass and Bandpass Sampling Due to the I/Q gain and phase imbalance problems in practical analog circuitry, the wideband downconversion - wideband sampling approach is very difficult to implement at 0 (or low) IF. But utilizing a combination of different techniques for mitigating these effects, the mentioned approach is becoming feasible, but mostly on the basestation side. On the other hand, wideband IF sampling is very challenging due to the apperture jitter and other implementation problems concerning the sampling circuitry (usually track&hold). The ADC requirements (apart from the sampling process) concern mainly the spurious-free dynamic range, and are not so heavily depending on the choice between lowpass or IF sampling. However, the useful ADC bandwidth has a great impact, e.g., on power consumption. So the cost/complexity metrics for per-channel A/D-conversion (usually ΣΔ) and multichannel A/D-conversion (usually something else than ΣΔ) are quite different.
TLT-5806/RxArch2/7 Some Dependencies and Conclusions Considering sampling and A/D-conversion highest signal frequency determines the T/H bandwidth and jitter requirements signal bandwidth (after analog RF/IF/baseband filtering) determines the minimum sampling rate Increasing the degree of bandpass subsampling / ) leads to ( f f c s lower sampling rate more selectivity needed before sampling more noise aliasing => more gain needed before sampling lower processing gain -> more bits from ADC & tighter jitter requirements Implementing the receiver selectivity in DSP-part leads to simplified analog part hard requirements for the T/H and ADC dynamic range Wideband sampling has been mostly considered at IF due to I/Qimbalance problems; direct conversion/low-if becoming feasible, depending on system specs Using IF sampling sets hard requirements for the T/H circuitry and jitter of the sampling clock.
TLT-5806/RxArch2/8 Connection to Advanced Broadband Wireless System Developments The latest and future wireless communication systems use increasing bandwidths for data transmission. For example, 3GPP-LTE and WiMAX have the maximum bandwidth of 20 MHz, and the next generation ( IMTadvance ) is targeted to bandwidths of up to 100 MHz. Especially, LTE is using frequency-division multiplexing, and the spectrum entering the receiver resembles that of a multichannel receiver for more narrowband systems. Some characteristics and comments: - The wide bandwidth makes it possible to utilize fast frequency hopping and other forms of frequency diversity, to enhance the transmitted data rate. - In LTE (and other similar systems) the power levels of the frequency slots of different users are well-controlled (e.g., 20 db maximum variation in the power levels). This is in contrast to, e.g., multichannel GSM receiver, were the dynamic range is much bigger. This makes it feasible to implement the needed wideband receivers for such systems. - Direct conversion architecture is preferred. Actually, for most of the frequency channels, the low-if model is valid. - IQ-imbalance is significant, but not very critical because of the well-controlled power levels. DSP-based IQimbalance compensation is interesting in case of highorder modulations. - In these systems, and in OFDM systems in general, the frequencies at or close to DC in baseband processing are commonly not utilized in order to make direct conversion receiver design easier.
TLT-5806/RxArch2/9 About Direct Sampling Architecture In high-performance systems, it is necessary to have some selectivity and gain before sampling. The reasons are signal aliasing noise aliasing Sampling is inherently more noisy operation than mixing! Sampling directly from the antenna signal is usually not adequate. The Ultimate SW Radio Architecture Antenna a bank of RF filters and LNA s for different frequency bands T/H A/D DSP The needed technologies are not mature for challenging radio system specifications in the frequency bands used in mobile systems! However, direct sampling is already an interesting architecture in various applications o For example, satellite-based positioning (GPS/Galileo) where the dynamic range requirements are greatly reduced comparing with wireless communications.
TLT-5806/RxArch2/10 Direct Sampling & Analog Discrete-Time Processing Texas Instruments (TI) has introduced so-called digital radio processor (DRP) concept that is based on direct sampling, together with analog discrete-time processing to implement main part of the channel selectivity, downconversion, and sampling rate reduction. o For example, CIC/running-sum filters can be implemented with switched-capacitor techniques with analog processing. o Then the ADC is operating at relatively low rate and has reduced dynamic range requirements compared to digital direct-sampling approach. TI is marketing DRP-based transceiver chips for GPS, Bluetooth, and GSM/GPRS, i.e., for systems with relatively narrow bandwidth or reduced dynamic range, together with low-order modulation. In such architectures, also the sampling process may be designed to provide frequency selectivity. Then the idea of the sampling process is not anymore just taking instantaneous sample values, but to - Integrate the signal over a finite-length interval - Weighting the input signal by a proper window during the integration interval. Rectangular window results in sinc-response, other kind of windows can be designed for optimized performance.
TLT-5806/RxArch2/11 Multimode Receivers In flexible multi-mode receivers, the target is to use common blocks for different systems as much as possible. A long-term target is to make the transceiver configurable for any system. However, presently a combination of a few predetermined systems is more realistic, e.g., GSM/WCDMA/WLAN. A realistic approach has the following elements: - Separate RF stages for different systems. - Common IF/baseband analog parts; bandwidth according to the most wideband system. - Common ADC at IF or baseband; fixed sampling rate. - Especially in the terminal side: careful choice of IF frequency & sampling rate to make the downconversion simple. Typically, f IF =(2k+1) f s /4. - Digital channel selection filtering optimized for the different systems. LNA IF 1 filter AGC S&H I DSP IF 2 ADC Q 0,1,0,-1 DSP LO 1,0,-1,0
TLT-5806/RxArch2/12 Rephrasing Critical Issues in Modern Receiver Architectures: SWOT Analysis on Flexible Receiver Architectures with Wideband Analog Front-End Alternatives: 1. Multimode direct-conversion receiver 2. Multimode low IF receiver 3. Multimode IF-sampling receiver 4. Wideband IF sampling architecture 5. Wideband direct-conversion/low-if architecture 6. Direct-sampling architecture Common features for 1-3: Analog & A/D bandwidth according to the widest channel bandwidth Common features for 4 and 5: Tunable digital channel selection and down-conversion Interesting mostly for base-station applications
TLT-5806/RxArch2/13 1. Multimode direct-conversion receiver Strengths - Simple analog part - Sampling jitter not critical - Narrowband A/D-conversion can be used Weaknesses - DC-offset problems, especially difficult to handle in flexible receiver - 2nd-order intermodulation -> bigger demands for the linearity of the analog parts Oportunities - Fast and flexible DC-offset compensation techniques, facilitated by high resolution A/D-conversion techniques.
TLT-5806/RxArch2/14 2. Multimode low IF receiver Strengths - Rather simple analog part - Sampling jitter not critical - Narrowband A/D-conversion can be used - DC-problems avoided Weaknesses - 2nd-order intermodulation -> bigger demands for the linearity of the analog parts - Higher demands for I/Q balance - Multimode concept not very clear (different low-if's for different systems, or very hard demands for I/Q balance) Oportunities - Adaptive I/Q imbalance compensation can be used to loosen the requirements of the analog part.
TLT-5806/RxArch2/15 3. Multimode IF-sampling receiver Strengths - Well-known architecture, high-quality analog RF possible - 2nd-order intermodulation not a problem - Narrowband A/D-conversion can be used - DC-problems avoided Weaknesses - Challenging demands for sampling jitter and linearity - IF filter difficult to integrate Oportunities - New technologies for flexible IF/RF filter implementation (e.g., MEMS)
TLT-5806/RxArch2/16 4. Wideband IF sampling architecture Strengths - Reduced IF filtering requirements - Simplified frequency synthesizer - Possibility to use common blocks for multiple channels - Facilitates fast frequency hopping/channel switching - DC-problems avoided Weaknesses - Very challenging demands for sampling jitter and linearity - Wideband A/D-conversion needed - 2nd-order intermodulation may be a problem - Lot of DSP power needed - High power consumption Oportunities - Advances in ADC technologies and DSP HW
TLT-5806/RxArch2/17 5. Wideband direct-conversion/low-if architecture Strengths - Simplified frequency synthesizer - Possibility to use common blocks for multiple channels - Facilitates fast frequency hopping/channel switching - Rather simple analog part - Sampling jitter not critical Weaknesses - Hard demands for I/Q balance - Multimode concept not very clear (avoiding DC-offset problems in all different systems) - Wideband A/D-conversion needed - 2nd-order intermodulation -> bigger demands for the linearity of the analog parts - High power consumption Oportunities - Adaptive I/Q imbalance compensation can be used to loosen the requirements of the analog part.
TLT-5806/RxArch2/18 6. Direct-sampling receiver Strengths - Simplest possible analog part - Highly flexible for multi-standard receivers. Weaknesses - Very hard jitter requirements. - Currently not feasible for demanding system specs or high-order modulation. Oportunities - Novel ideas for sampling and ADCs - Analog discrete-time processing techniques.
TLT-5806/RxArch2/19 Case Study on Wideband IF Sampling in GSM Receivers* Introduction Specifications and Selectivity Requirements for GSM Sampling and Quantization Requirements as Functions of Analog Filter Bandwidth Requirements for Digital Filtering * This part is based on the diploma thesis work of Juho Pirskanen carried out at TUT/ICE during years 1999-2000.
TLT-5806/RxArch2/20 GSM Case: Introduction Today s receivers (like GSM) use narrowband ADconversion When only one system is to be implemented, this is not a problem However, when several systems with different bandwidths are desired to be used, we have the choices Several different analog front-ends in the receiver OR Wideband analog front-end and ADconversion
TLT-5806/RxArch2/21 GSM Case: Wideband Receiver Receiver with wideband front-end and wideband ADconversion Analog front-end can be simplified One AD-converter can be used for different systems Performance requirements of the ADC are increased Channelization filtering must be done in digital domain to obtain desired system characteristics
TLT-5806/RxArch2/22 GSM Case: Interference Mask Obtained from the GSM specifications Includes interference signals from Adjacent channel Out of band blocking Intermodulation test 20 GSM Interferer Mask 30 40 50 Signal Power [dbm] 60 70 80 90 100 Desired Signal Level for the Blocking Test Reference Sensitivity Level 110 120 5000 4000 3000 2000 1000 0 1000 2000 3000 4000 5000 Frequency Offset from the Carrier Frequency [khz]
TLT-5806/RxArch2/23 GSM Case: Attenuation Requirement Attenuation requirements for GSM can be found by ( ) A ( f) = P( f) P C/ I A s I sign c m P I is the interference signal P sign is the desired signal C/I c is the carrier to interference ratio A m is the extra noise margin Channelization Filtering for GSM 0 10 20 Magnitude Response [db] 30 40 50 60 70 80 90 5000 4000 3000 2000 1000 0 1000 2000 3000 4000 5000 Frequency Offset from the Carrier Frequency [khz]
TLT-5806/RxArch2/24 GSM Case: ADC Dynamic Requirements ADC dynamic range requirement can be calculated as dynamic max{ s ( ) ( )} SNR = A f + H f f A s is the attenuation requirement H(f) is the amplitude response of the analog filter Fourth-order Chebyshev type two filters : GSM Specifications and some 4th Order Bandpass Chebyshev Filters 0 10 20 Magnitude Response [db] 30 40 50 60 70 80 90 0 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 Frequency Offset from the Carrier frequency [khz] The red squares mark the critical points where the dynamic range requirement is maximized for each filter bandwidth.
TLT-5806/RxArch2/25 GSM Case: ADC Dynamic Requirements By combining equations, the number of bits can be found by b f SNR 1.76 10log s dynamic 10 2B = 6.02 Used sampling rates Multiples of GSM symbol rate 17.33 MHz, 34.66 MHz and 69.33 MHz Studied filter types Butterworth and Chebyshev type two filters Used filter orders Fourth and sixth order filters Filter bandwidth From 100 khz to 2.5 MHz
TLT-5806/RxArch2/26 GSM Case: Number of Bits Required in ADC Fourth-order Butterworth filters: Bits used in AD converter 14 12 10 8 6 4 2 14 12 10 8 6 4 2 14 12 10 8 6 4 2 AD conventers Bits as a Function of Filter Bandwidth. Fs = 92.16MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 AD conventers Bits as a Function of Filter Bandwidth. Fs = 69.333MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 AD conventers Bits as a Function of Filter Bandwidth. Fs = 34.6666MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 Filters bandwidth [khz] Sixth-order Chebyshev type two filters: Bits used in AD converter 14 12 10 8 6 4 2 14 12 10 8 6 4 2 14 12 10 8 6 4 2 AD conventers Bits as a Function of Filter Bandwidth. Fs = 92.16MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 AD conventers Bits as a Function of Filter Bandwidth. Fs = 69.333MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 AD conventers Bits as a Function of Filter Bandwidth. Fs = 34.6666MHz 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 Filters bandwidth [khz] Notice that in practice the minimum number of bits is higher than the lowest values indicated here, in order to be able to carry out the channel equalization properly.
TLT-5806/RxArch2/27 GSM Case: Jitter Noise The maximum signal power to be sampled is: ADC max{ I ( ) ( )} P = P f + H f f Using the standard white-noise model for the jitter effects, the maximum allowed standard deviation of the timing error is given by: T A = F s P C 2B sig Ic 4π f P 2 2 max ADC A m
TLT-5806/RxArch2/28 GSM Case: Jitter Requirements for f IF =156 MHz The timing jitter requirements when using fourth-order Butterworth filters: 10 3 Maximum Allowable Standart Deviation of The Timing Jitter Fs=69.333 MHz Fs=36.7 MHz Fs=17.8 MHz Standard Deviaton of Jitter [ps] 10 2 10 1 10 0 10 1 0 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 Filters Bandwidth [khz] The timing jitter requirements when using sixth-order Chebyshev type two filters: Maximum Allowable Standart Deviation of The Timing Jitter 10 3 Fs=69.333MHz Fs=36.7MHz Fs=17.8MHz Standard Deviaton of Jitter [ps] 10 2 10 1 10 0 10 1 0 500 1000 1500 2000 2500 3000 3500 4000 4500 5000 Filters Bandwidth [khz]
TLT-5806/RxArch2/29 GSM Case: Digital Filtering Requirements Channelization and noise filtering requirements for DSP in the GSM case. Sixth order Chebyshev type two analog filter Second order Sigma-delta modulator with 2 quantization bits Attenuation Requirements for Noise Filtering and Interfering Channels 0 10 20 Stopband Attenuation[dB] 30 40 50 60 70 RF Attenuation Requirements Attenuation Requirements for Noise Power at the Multiples of the Signal Band 2B 80 Attenuation Requirements for Total Quantization Noise Power 90 0 0.5 1 1.5 2 2.5 3 Frequency [khz] x 10 4 The total decimation factor can be divided to several stages, e.g., 256 = 64 * 2*2 First stage can be implemented by using CIC-filters Simple structure, no multipliers Good attenuation for aliasing signal bands 0 Filtering Requirements and CIC Decimation Filter 20 Magnitude Response [db] 40 60 80 100 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 Frequency Offset from Carrier Frequency [khz] x 10 4
TLT-5806/RxArch2/30 GSM Case: Conclusions Dynamic requirement of the ADC Highly effected by the analog filter bandwidth Analog filter order and type has only one bit effect on ADC requirement (together 2 bits in some cases) Standard deviation of timing jitter Highly effected by the analog filter bandwidth and used IF frequency (IF sampling) Analog filter order and type has only slight effect When considering GSM/WCDMA receivers The analog bandwidth should be about 2 MHz Fractional decimation has to be done
TLT-5806/RxArch2/31 DSP for Flexible Receivers In advanced SW radio concepts, the selectivity filtering and down-conversion are moved from analog continuoustime part to the discrete-time/dsp part. => Here efficient multirate filtering techniques become very important. It also helps to move as much as possible functionality from the analog or digital front-end to baseband processing. => All-digital synchronization concept becomes very interesting in this context. Some errors due to RF-front-end can be corrected by DSP. Example: Adaptive I/Q imbalance compensation.
TLT-5806/RxArch2/32 All-Digital Synchronisation Concept Free-running local oscillators for demodulation and frequency conversion. Free-running sampling clock. Errors are compensated in digital part. => All synchronisation functions can be implemented using digital techniques
TLT-5806/RxArch2/33 Digital Channel Selection & Down-Conversion
TLT-5806/RxArch2/34 Digital Channel Selection & Down-Conversion Digital Down-Conversion 1. Desired channel centered at fixed IF => Fixed down-conversion Special choices of f IF and f s make things easy. Especially when f IF =(2k+1) f s /4, the signal aliases to f s /4 and down-conversion is very easy. 2. Wideband sampling case => Tunable down-conversion and NCO (numerically controlled oscillator) needed. 3. Stepwise mixing and decimation => Tunable digital down-conversion is possible also without NCO, as demonstrated in the later example. However, it is not easy to find sufficiently efficient and flexible schemes. Channel Selection Filtering - After down-conversion, efficient lowpass decimator structure is needed. - CIC-filters are commonly used in the first decimation stages, FIR-filters and the last stages. Nth-band IIR filters also an efficient solution. Adjusting Symbol Rates - Different systems use different symbol/chip rates. - Common sampling clock frequency is preferred. => Decimaton by a fractional factor is needed. - This can be done at baseband or earlier in the decimation chain.
TLT-5806/RxArch2/35 CIC-Filters z -1 z -1 32 z -1 z -1 CIC = Cascaded Integrator - Comb Transfer function: H ( z) z R 1 = 1 z 1 N j2πf H e F Frequency response: s = e jπn ( R 1) f F s Rf sin π R f sin π N F s F s Here R is the decimation factor and N is the order of the CIC-filter. A first-order CIC-filter takes the avarage of R consequtive input samples and decimates by R. It is also called moving average or running sum filter. It is important to use modulo arithmetic (like 2's complement) in the implementation, because there will be inevitable internal overflows.
TLT-5806/RxArch2/36 CIC-Filters In CIC filter, those frequencies aliasing to 0-frequency are heavily attenuated. For a relatively narrowband signal, low-order CIC-filters are sufficient; more wideband signals neede higher CIC-filter orders Example (for a GSM application): N=2, R=32.
TLT-5806/RxArch2/37 NCO-Based Arbitrary Digital Down- Conversion Dedicated processors implementing the following kind of down-conversion and channel selection structure are available for several vendors (like Harris). Sampling rates in the 50... 100 MHz range are possible. However, the power consumption is still too high for terminal applications.
TLT-5806/RxArch2/38 Example Using Harris HSP50214 for GSM channel selection filtering. - Input sample rate: 39 MHz - CIC-filter: decimation by 18, order=5 - Two pre-designed FIR half-band filters are used for the next decimation stages. - The final filter stage is an FIR design. - Output sample rate: 541.667 khz Frequency responses of the filter stages:
TLT-5806/RxArch2/39 Example (continued) Overall frequency response and the effects of different stages:
TLT-5806/RxArch2/40 Stepwise Decimation and Mixing Approach With suitable choice of the key parameters (IF frequency, channel spacing, sampling rate) it is possible to do adjustable down-conversion also without NCO, just by using the frequency translations of multirate DSP. In the following, a special bandpass decimator structure is described, which gives some more flexibility in this kind of solutions. The idea is to do stepwise down-conversion, with decimation by 2 in each stage. Three different types of stages are used: 1. Lowpass decimation when the desired signal is at the lower frequencies. 2. Highpass decimation when the desired signal is at the higher frequencies. 3. The special bandpass decimator when the desired signal is in the mid-frequencies. Note: This example should be taken as an example of the possibilities of complex signal processing in specific designs, but not as a generic technique for SDR.
TLT-5806/RxArch2/41 The structure includes: - down-conversion by f s /8 - bandpass filtering - decimation by 2 Bandpass Decimator Complex filtering is needed in the basic model, but the structure can be optimized to an efficient form, where complex signal processing is not needed. Basic model: e -jπk/4 H 4 (ze -jπ/2 ) x Re[] 2 Magnitude Response H (ze -jπ/2 ) 4 π/4 π/2 3π/4 π Angular Frequency Magnitude Response π/4 π/2 3π/4 π
TLT-5806/RxArch2/42 Optimized form: Bandpass Decimator (continued) A 1 (z) 1,1,-1,-1,1,1,-1 A 0 (z) A 3 (z) - + - + x A 2 (z) - This is completely equivalent to the original form. - No complex signal processing actually needed. - All filtering operations running at quarter of the sampling rate. - Computational complexity roughly the same as for halfband decimators with the same selectivity.
TLT-5806/RxArch2/43 Basic Building Blocks for the SDM Approach 0 Magnitude responses of H 2 (z) and H 2 (-z) Magnitude [db] -20-40 -60-80 -100 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 Normalized Frequency Magnitude response of H 4 (ze -jπ/2 ) 0 Magnitude [db] -20-40 -60-80 -100 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 Normalized Frequency A (z) 0 + LP out A (z) 1 - + HP out A 1 (z) A 0 (z) A 3 (z) - + - + 1,1,-1,-1,1,1,-1 x BP out A 2 (z)