Low-Output-Voltage, 800mA, PWM Step-Down DC-DC Converters

Transcription

1 9-2527; Rev 0; 7/02 Low-Output-oltage, 800mA, Step-Down General Description The 800mA step-down converters power low-voltage microprocessors in compact equipment requiring the highest possible efficiency. The are optimized for generating low output voltages (down to 7m) at high efficiency using small external components. The supply voltage range is from 2.6 to 5.5 and the guaranteed minimum output current is 800mA. MHz pulse-width modulation () switching allows for small external components. A unique control scheme minimizes ripple at light loads, while maintaining a low 40µA quiescent current. The include a low on-resistance internal MOSFET switch and synchronous rectifier to maximize efficiency and minimize external component count. No external diode is needed. % duty-cycle operation allows for a dropout voltage of only 340m at 800mA. Other features include internal soft-start, power-ok (POK) output, and selectable forced operation for lower noise at all load currents. The MAX928 is available with several preset output voltages:.5 (MAX928-5),.8 (MAX928-8), and 2.5 (MAX928-25). The MAX927R has adjustable output range down to The are available in a tiny 0-pin µmax package. WCDMA Handsets PDAs and Palmtops DSP Core Power Battery-Powered Equipment Applications 800mA Output Current Output oltages from 0.75 to to 5.5 Input oltage Range Power-OK Output No Schottky Diode Required Selectable Forced Operation MHz Fixed-Frequency Operation 40µA Quiescent Current Soft-Start 0-Pin µmax Package PART Features Ordering Information PRESET OUTPUT OLTAGE TEMP RANGE PIN- PACKAGE MAX927REUB Adj. to C to +85 C 0 µmax MAX928EUB C to +85 C 0 µmax MAX928EUB C to +85 C 0 µmax MAX928EUB C to +85 C 0 µmax Pin Configuration Typical Operating Circuit TOP IEW 0 POK IN 2.6 TO 5.5 C BATT L 0.75 AT 800mA C2 GND REF FB MAX927R MAX BATT PGND C C R C C f COMP MAX927R FB POK COMP 5 6 µmax REF GND PGND Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at , or visit Maxim s website at

2 ABSOLUTE MAXIMUM RATINGS BATT,, POK, COMP, to GND to +6 PGND to GND to +0.3, REF, FB to GND to ( BATT + 0.3) Continuous Power Dissipation (T A = +70 C) 0-Pin µmax (derate 5.6mW/ C above +70 C)...444mW Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS Operating Temperature Range C to +85 C Junction Temperature...+ C Storage Temperature Range C to + C Lead Temperature (soldering, 0s) C ( BATT = 3.6, = BATT, C REF = 0.µF, T A = 0 C to +85 C, unless otherwise noted. Typical values are at T A = +25 C.) PARAMETER CONDITIONS MIN TYP MAX UNITS BATT Input oltage Undervoltage Lockout Threshold BATT rising or falling (35m hysteresis) Quiescent Current No load, pulse skipping, = GND µa MHz switching 2 ma Quiescent Current in Dropout µa Shutdown Supply Current = GND 0. 0 µa REFERENCE AND ERROR AMP FB oltage Accuracy FB Input Current Transconductance (g m ) MAX927R MAX MAX MAX MAX µa MAX927R 0 na MAX927R 2 MAX MAX MAX Reference oltage Accuracy Reference Supply Rejection 2.6 < BATT < m CONTROLLER P-Channel On-Resistance N-Channel On-Resistance BATT = BATT = BATT = BATT = Current-Sense Transresistance (R CS ) 0.48 /A P-Channel Current-Limit Threshold..3.6 A P-Channel Pulse-Skipping Current Threshold A N-Channel Negative Current-Limit Threshold A µs Ω Ω 2

3 ELECTRICAL CHARACTERISTICS (continued) ( BATT = 3.6, = BATT, C REF = 0.µF, T A = 0 C to +85 C, unless otherwise noted. Typical values are at T A = +25 C.) PARAMETER CONDITIONS MIN TYP MAX UNITS N-Channel Synchronous Rectifier Turn-Off Threshold 20 ma Leakage Current BATT = 5.5, = GND or BATT µa Maximum Duty Cycle % Minimum Duty Cycle = GND 0 = BATT 5 Internal Oscillator Frequency MHz Thermal Shutdown Threshold 5 C hysteresis 60 D eg r ees POK COMPARATOR BATT Operating oltage Range I POK = 0. ma 5.5 Output Low oltage FB = 0.5, I POK = ma Output High Leakage Current POK = 5.5 µa POK Threshold Output alid to POK Release Delay LOGIC INPUTS (, ) MAX927R MAX MAX MAX POK transitions to high impedance 20ms after FB > POK ms Logic Input High 2.6 < BATT < Logic Input Low 2.6 < BATT < Logic Input Current BATT = µa % ELECTRICAL CHARACTERISTICS ( BATT = 3.6, = BATT, C REF = 0.µF, T A = -40 C to +85 C, unless otherwise noted.) PARAMETER CONDITIONS MIN MAX UNITS BATT Input oltage Undervoltage Lockout Threshold BATT rising or falling (35m hysteresis) Quiescent Current No load, pulse skipping, = GND 240 µa Quiescent Current in Dropout 340 µa Shutdown Supply Current = GND 0 µa REFERENCE AND ERROR AMP MAX927R FB oltage Accuracy MAX MAX MAX FB Input Current MAX µa 3

4 ELECTRICAL CHARACTERISTICS (continued) ( BATT = 3.6, = BATT, C REF = 0.µF, T A = -40 C to +85 C, unless otherwise noted. Typical values are at T A = +25 C.) PARAMETER CONDITIONS MIN MAX UNITS FB Input Current MAX927R na Reference oltage Accuracy Reference-Supply Rejection 2.6 < BATT < m CONTROLLER P-Channel On-Resistance N-Channel On-Resistance BATT = BATT = BATT = BATT = P-Channel Current-Limit Threshold A P-Channel Pulse-Skipping Current Threshold A Leakage Current BATT = 5.5, = GND or BATT µa Maximum Duty Cycle % Minimum Duty Cycle = GND 0 % Internal Oscillator Frequency MHz POK COMPARATOR BATT Operating oltage Range I POK = 0. ma 5.5 Output Low oltage FB = 0.5, I POK = ma 0. Output High Leakage Current POK = 5.5 µa POK Threshold Output alid to POK Release Delay LOGIC INPUTS (, ) MAX927R MAX MAX MAX POK transitions to high impedance 20ms after FB > POK 5 25 ms Logic Input High 2.6 < BATT < Logic Input Low 2.6 < BATT < Logic Input Current BATT = 5.5 µa Ω Ω 4

5 (Circuits of Figure 3 and 4, T A = +25 C, unless otherwise noted.) EFFICIENCY (%) MAX927R EFFICIENCY vs. LOAD CURRENT IN = 3.6 IN = LOAD CURRENT (ma) = 3.3 MAX927 toc0 EFFICIENCY (%) MAX EFFICIENCY vs. LOAD CURRENT IN = 3.6 IN = LOAD CURRENT (ma) Typical Operating Characteristics MAX927 toc02 EFFICIENCY (%) MAX928-8 EFFICIENCY vs. LOAD CURRENT IN = 2.7 IN = 5 IN = 3.6 = LOAD CURRENT (ma) MAX927 toc03 EFFICIENCY (%) MAX928-5 EFFICIENCY vs. LOAD CURRENT IN = 2.7 IN = 3.6 IN = 5 = LOAD CURRENT (ma) MAX927 toc04 EFFICIENCY (%) MAX927R EFFICIENCY vs. LOAD CURRENT IN = 2.7 IN = 3.6 IN = 5 = 0 0 LOAD CURRENT (ma) MAX927 toc05 DROPOUT OLTAGE (m) MAX DROPOUT OLTAGE vs. LOAD CURRENT IN = LOAD CURRENT (A) MAX927 toc06 OUTPUT OLTAGE () MAX928-8 OUTPUT OLTAGE vs. LOAD CURRENT IN = LOAD CURRENT (ma) MAX927 toc07 INPUT CURRENT (µa) NO-LOAD INPUT CURRENT vs. INPUT OLTAGE INPUT OLTAGE () MAX927 toc08 OSCILLATOR FREQUENCY (MHz) OSCILLATOR FREQUENCY vs. INPUT OLTAGE T A = +85 C T A = +25 C T A = -40 C INPUT OLTAGE () MAX927 toc09 5

6 Typical Operating Characteristics (continued) (Circuits of Figure 3 and 4, T A = +25 C, unless otherwise noted.) MAXIMUM LOAD CURRENT (A) MAXIMUM LOAD CURRENT vs. INPUT OLTAGE = =.8 = INPUT OLTAGE () MAX927 toc0 I IN STARTUP WAEFORM ms/div MAX927 toc 5/div /div 200mA/div POK POK WAEFORM 20ms/div MAX927 toc2 5/div 2/div 2/div HEAY-LOAD SWITCHING WAEFORMS MAX927 toc3 LIGHT-LOAD SWITCHING WAEFORMS MAX927 toc4 (AC-COUPLED) 0m/div (AC-COUPLED) 0m/div I L 200mA/div 5/div 5/div I L 200mA/div 400ns/div 2ms/div LOAD TRANSIENT MAX927 toc5 LINE TRANSIENT MAX927 toc6 (AC-COUPLED) m/div (AC-COUPLED) 0m/div I LOAD 900mA 0mA/div 2mA IN /div µs/div ms/div 6

7 PIN NAME FUNCTION Pin Description Forced- Input. Drive to GND to use at medium to heavy loads and pulse-skipping at light loads. Drive to BATT to force operation at all loads. 2 GND Ground 3 REF Internal.25 Reference. Bypass to GND with a 0.µF capacitor. 4 FB Output Feedback Sense Input. To set the output voltage to the preset voltage (MAX928), connect FB directly to the output. To adjust the output voltage (MAX927R), connect FB to the center of an external resistordivider between the output and GND. FB regulation voltage is COMP Compensation Input. See the Compensation, Stability, and Output Capacitor section for compensation component selection. 6 Shutdown Control Input. Drive low to shut down the converter. Drive high for normal operation. 7 PGND Power Ground 8 Inductor Connection to the drains of the internal power MOSFETs. 9 BATT Supply oltage Input. Connect to a 2.6 to 5.5 source. Bypass to GND with a low-esr 0µF capacitor. 0 POK Power-OK Open-Drain Output. Once the soft-start routine has completed, POK goes high impedance 20ms after FB exceeds 90% of its expected final value. BATT COMP SLOPE COMPENSATION COMPARATOR BIAS P P MAX927 MAX928 MHz OSC PFM CURRENT COMPARATOR CONTROL ILIM COMPARATOR N N N-CHANNEL CURRENT COMPARATOR PGND TO COMP REF.25 REFERENCE POWER-OK CONTROL POK FB MAX927R ONLY MAX928 ONLY GND Figure. Simplified Functional Diagram 7

8 Detailed Description The step-down DC-DC converters accept inputs as low as 2.6, while delivering 800mA to output voltages as low as These devices operate in one of two modes to optimize noise and quiescent current. Under heavy loads, MAX927/ MAX928 operate in pulse-width modulation () mode and switch at a fixed MHz frequency. Under light loads, they operate in PFM mode to reduce power consumption. In addition, both devices provide selectable forced operation for minimum noise at all load currents. PFM Operation and Control Scheme The PFM mode improves efficiency and reduces quiescent current to 40µA at light loads. The MAX927/ MAX928 initiate pulse-skipping PFM operation when the peak inductor current drops below 30mA. During PFM operation, the switch only as necessary to service the load, reducing the switching frequency and associated losses in the internal switch, synchronous rectifier, and inductor. During PFM mode, a switching cycle initiates when the error amplifier senses that the output voltage has dropped below the regulation point. If the output voltage is low, the P-channel MOSFET switch turns on and conducts current to the output filter capacitor and load. The PMOS switch turns off when the comparator is satisfied. The then wait until the error amplifier senses a low output voltage to start again. Some jitter is normal during the transition from PFM to with loads around ma. This has no adverse impact on regulation. At loads greater than 30mA, the use a fixed-frequency, current-mode, controller capable of achieving % duty cycle. Current-mode feedback provides cycle-by-cycle current limiting, superior load and line response, as well as overcurrent protection for the internal MOSFET and synchronous rectifier. A comparator at the P-channel MOSFET switch detects overcurrent conditions exceeding.a. During operation, the regulate output voltage by switching at a constant frequency and then modulating the power transferred to the load using the comparator (Figure ). The error-amp output, the main switch current-sense signal, and the slope compensation ramp are all summed at the comparator. The comparator modulates the output power by adjusting the peak inductor current during the first half of each cycle based on the output-error voltage. The have relatively low ACloop gain coupled with a high-gain integrator to enable the use of a small, low-valued, output filter capacitor. The resulting load regulation is 0.3% (typ) from 0 to 800mA. Forced Operation To force -only operation, connect to BATT. Forced operation is desirable in sensitive RF and data-acquisition applications to ensure that switching noise does not interfere with sensitive IF and data sampling frequencies. A minimum load is not required during forced operation because the synchronous rectifier passes reverse inductor current as needed to allow constant frequency operation with no load. Forced operation has higher quiescent current than PFM (2mA typ compared to 40µA) due to continuous switching. % Duty-Cycle Operation The maximum on-time can exceed one internal oscillator cycle, which permits operation at % duty cycle. As the input voltage drops, the duty cycle increases until the internal P-channel MOSFET stays on continuously. Dropout voltage at % duty cycle is the output current multiplied by the sum of the internal PMOS onresistance (typically 0.25Ω) and the inductor resistance. Near dropout, switching cycles can be skipped, reducing switching frequency. However, voltage ripple remains small because the current ripple is still low. Synchronous Rectification An N-channel synchronous rectifier eliminates the need for an external Schottky diode and improves efficiency. The synchronous rectifier turns on during the second half of each cycle (off-time). During this time, the voltage across the inductor is reversed, and the inductor current falls. In normal mode, the synchronous rectifier is turned off when either the output falls out of regulation (and another on-time begins) or when the inductor current approaches zero. In forced mode, the synchronous rectifier remains active until the beginning of a new cycle. Shutdown Mode Driving to GND places the in shutdown mode. In shutdown, the reference, control circuitry, internal switching MOSFET, and synchronous rectifier turn off and the output becomes high impedance. Drive high for normal operation. Input current falls to 0.µA (typ) during shutdown mode. POK Output POK is an open-drain output that goes high impedance 20ms after the soft-start ramp has concluded and FB is within 90% of the threshold. POK is low impedance when in shutdown. 8

9 MAX927R Applications Information Output oltage Selection The have preset output voltages. In addition, the MAX927R has an adjustable output. To set the output voltage at the preset voltage, connect FB to the output. See Table for a list of the preset voltages and their corresponding part numbers. The output voltage for the MAX927R is adjustable from 0.75 to the input voltage by connecting FB to a resistor-divider between the output and GND (Figure 2). To determine the values of the resistor-divider, first select a value for feedback resistor R2 between 5kΩ to kω. R is then given by: where FB is FB R R OUT = 2 FB Input Capacitor Selection Capacitor equivalent series resistance (ESR) is a major contributor to input ripple in high-frequency DC-DC converters. Ordinary aluminum-electrolytic capacitors have high ESR and should be avoided. Low-ESR aluminum electrolytic capacitors are acceptable and relatively inexpensive. Low-ESR tantalum capacitors or polymer capacitors are better and provide a compact solution for space-constrained surface-mount designs. Ceramic capacitors have the lowest ESR overall. The input filter capacitor reduces peak currents and noise at the input voltage source. Connect a low-esr bulk capacitor ( 0µF typ) to the input. Select this bulk capacitor to meet the input ripple requirements and voltage rating rather than capacitance value. Use the R R2 kω Figure 2. Setting the Adjustable Output oltage Table. FB Regulation oltages MAX927R MAX928-5 MAX928-8 MAX PART PRESET OUTPUT OLTAGE 0.75, Adjustable following equation to calculate the maximum RMS input current: I IRMS = OUT OUT IN OUT IN ( ) Compensation, Stability, and Output Capacitor The are externally compensated with a resistor and a capacitor (see Figure 3, R C and C C ) in series from COMP to GND. An additional capacitor (C f ) may be required from COMP to GND if high- ESR output capacitors are used. The capacitor integrates the current from the transimpedance amplifier, averaging output capacitor ripple. This sets the device speed for transient response and allows the use of small ceramic output capacitors because the phaseshifted capacitor ripple does not disturb the current regulation loop. The resistor sets the proportional gain of the output error voltage by a factor g m R C. Increasing this resistor also increases the sensitivity of the control loop to output ripple. The resistor and capacitor set a compensation zero that defines the system s transient response. The load creates a dynamic pole, shifting in frequency with changes in load. As the load decreases, the pole frequency decreases. System stability requires that the compensation zero must be placed to ensure adequate phase margin (at least 30 at unity gain). The following is a design procedure for the compensation network: ) Select an appropriate converter bandwidth (f C ) to stabilize the system while maximizing transient response. This bandwidth should not exceed /0 of the switching frequency. 2) Calculate the compensation capacitor, C C, based on this bandwidth: For the MAX927: R CC = OUT gm I OUT MAX RCS 2 R + R f ( ) 2 2π C 9

10 For the MAX928: CC = OUT gm I OUT MAX R CS ( ) f ( ) 2π C Resistors R and R2 are external to the MAX927 (see the Setting the Output oltage section). I OUT(MAX) is the maximum output current, R CS = 0.48/A, and g m = 2µS for the MAX927. See the Electrical Characteristics table for MAX928 g m values. Select the closest standard C C value that gives an acceptable bandwidth. 3) Calculate the equivalent load impedance, R L, by: RL = OUT IOUT( MAX) 4) Calculate the compensation resistance (R C ) to cancel out the dominant pole created by the output load and the output capacitance: Solving for R C gives: = 2π R C 2π R C L OUT C C R C R L OUT C = CC 5) Calculate the high-frequency compensation pole to cancel the zero created by the output capacitor s ESR: = 2π RESR COUT 2π RC Cf IN 2.6 TO 5.5 C 0µF BATT MAX928-8 FB L CDRH4D8 4.7µH.8 AT 800mA C2 0µF C C 200pF R C 8kΩ C3 0.µF C f 22pF COMP REF GND POK PGND Figure 3. Applications Circuit for the MAX928 C C 680pF IN 2.6 TO 5.5 C 0µF R C 5kΩ C3 0.µF C f 22pF BATT COMP REF MAX927R GND FB POK PGND L CDRH4D8 4.7µH R 6.5kΩ % R2 49.9kΩ % AT 800mA C2 0µF Figure 4. Applications Circuit for the MAX927 0

11 Solving for C f gives: C R ESR C OUT f = RC or 22pF, whichever is greater. Standard Application Circuits Figures 3 and 4 are standard applications circuits for the. Figure 3 illustrates the preset output voltages (MAX928), while Figure 4 shows the adjustable configuration (MAX927). Table 2 lists part numbers and suppliers for the components used in these circuits. PC Board Layout and Routing High switching frequencies and large peak currents make PC board layout a very important part of design. Good design minimizes EMI, noise on the feedback paths, and voltage gradients in the ground plane, all of which can result in instability or regulation errors. Connect the inductor, input filter capacitor, and output filter capacitor as close together as possible and keep their traces short, direct, and wide. Connect their ground pins at a single common node in a star ground configuration. The external voltage feedback network should be very close to the FB pin, within 0.2in (5mm). Keep noisy traces, such as those from the pin, away from the voltage feedback network. Position the bypass capacitors as close as possible to their respective pins to minimize noise coupling. For optimum performance, place input and output capacitors as close to the device as possible. Connect GND and PGND to the highest quality system ground. The MAX928 evaluation kit illustrates an example PC board layout and routing scheme. TRANSISTORS: 3282 PROCESS: BiCMOS Chip Information Table 2. Suggested Parts/Suppliers PART PART NUMBER MANUFACTURER PHONE WEBSITE Inductor CDRH3D6-4R7 Sumida USA Japan Input/Output Capacitors JMK22BJ06MG Taiyo Yuden COMP Capacitor GRM88XH56J Murata REF Capacitor EMK07BJ04KA Taiyo Yuden

12 Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to 0.6±0. 0 e ÿ 0.±0. 0.6±0. TOP IEW 4X S H BOTTOM IEW 0 DIM A A MIN MAX MIN MAX A D D2 E E2 H L L b e c S α INCHES MILLIMETERS REF REF BSC 0.0 BSC REF REF LUMAX.EPS D2 E2 GAGE PLANE A2 A c D b A α E L L FRONT IEW SIDE IEW PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE, 0L umax/usop APPROAL DOCUMENT CONTROL NO. RE I Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 2 Maxim Integrated Products, 20 San Gabriel Drive, Sunnyvale, CA Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.