TR (draft) V0.0.0 (2003-xx)

Transcription

1 Technical Report Transmission and Multiplexing (TM); Spectral management on metallic access networks; Part 2: Technical methods or perormance evaluations Work Item Reerence Permanent Document DTS/TM TM6(01)20 Filename m01p20a4.pd (rev 4) Date may 28, 2003 Rapporteur/Editor Rob F.M. van den Brink tel: (on behal o KPN) TNO Telecom ax: PO Box R.F.M.vandenBrink@telecom.tno.nl 2260 AK Leidschendam The Netherlands

2 2 Reerence DTS/TM Keywords spectral management, unbundling, access network, local loop, transmission, modem, POTS, IDSN, ADSL, HDSL, SDSL, VDSL, xdsl 650 Route des Lucioles F Sophia Antipolis Cedex - FRANCE Tel.: Fax: Siret N NAF 742 C Association à but non lucrati enregistrée à la Sous-Préecture de Grasse (06) N 7803/88 Important notice Individual copies o the present document can be downloaded rom: The present document may be made available in more than one electronic version or in print. In any case o existing or perceived dierence in contents between such versions, the reerence version is the Portable Document Format (PDF). In case o dispute, the reerence shall be the printing on printers o the PDF version kept on a speciic network drive within Secretariat. Users o the present document should be aware that the document may be subject to revision or change o status. Inormation on the current status o this and other documents is available at I you ind errors in the present document, send your comment to: editor@etsi.r Copyright Notiication No part may be reproduced except as authorized by written permission. The copyright and the oregoing restriction extend to reproduction in all media. European Telecommunications Standards Institute All rights reserved.

3 3 Contents Intellectual Property Rights...5 Foreword Scope Reerences Deinitions and abbreviations Deinitions Abbreviations Transmitter signal models or xdsl Generic transmitter signal model Cluster 2 transmitter signal models transmitter signal model or "ISDN.2B1Q" transmitter signal model or "ISDN.MMS.43" transmitter signal model or "Proprietary.SymDSL.CAP.QAM" Cluster 3 transmitter signal models Transmitter signal models or "HDSL.2B1Q" Transmitter signal models or "HDSL.CAP" Transmitter signal model or "SDSL" Transmitter signal model or "Proprietary.SymDSL.CAP.A::Fn" Transmitter signal model or "Proprietary.SymDSL.CAP.B::Fn" Transmitter signal model or "Proprietary.SymDSL.CAP.C::Fn" Transmitter signal model or "Proprietary.SymDSL.PAM::Fn" Transmitter signal model or "Proprietary.SymDSL.2B1Q::Fn" Transmitter signal model or "Proprietary.PCM.HDB3.2M.SR" Transmitter signal model or "Proprietary.PCM.HDB3.2M.SQ" Cluster 4 transmitter signal models Transmitter signal model or "ADSL over POTS" Transmitter signal model or "ADSL over ISDN" Transmitter signal model or "ADSL.FDD over POTS" Transmitter signal model or "ADSL.FDD over ISDN" Cluster 5 transmitter signal models Transmitter signal model or "VDSL" Generic receiver perormance models or xdsl Generic input models or eective SNR Linear input model or eective SNR Advanced input models or eective SNR Generic detection models Generic Shited Shannon detection model Generic PAM detection model Generic CAP/QAM detection model Generic DMT detection model Speciic receiver perormance models or xdsl Receiver perormance model or "HDSL.2B1Q" Receiver perormance model or "HDSL.CAP" Receiver perormance model or "SDSL" Receiver perormance model or "ADSL over POTS" Receiver perormance model or "ADSL.FDD over POTS" Receiver perormance model or "ADSL over ISDN" Receiver perormance model or "ADSL.FDD over ISDN Receiver perormance model or "VDSL" Transmission and relection models Summary o test loop models Basic model or echo loss...18

4 4 8 Cross talk models Overview o dierent network topologies Validity limitations o cross talk modeling Generic cross talk models or two-node co-location Basic diagram or two-node topologies Models or cross talk cumulation FSAN sum or cross talk cumulation Models or cross talk coupling Basic models or equivalent NEXT and FEXT Models or cross talk injection Forced noise injection Current noise injection Generic cross talk models or multi-node co-location Measurement methods Examples o evaluating various scenarios Example scenario A Assumed coniguration Assumed conditions Evaluated perormance or scenario A Example scenario B Example scenario C Example scenario D...26 Annex A: Bibliography...26 History...27

5 5 Intellectual Property Rights IPRs essential or potentially essential to the present document may have been declared to. The inormation pertaining to these essential IPRs, i any, is publicly available or members and non-members, and can be ound in SR : "Intellectual Property Rights (IPRs); Essential, or potentially Essential, IPRs notiied to in respect o standards", which is available rom the Secretariat. Latest updates are available on the Web server ( Pursuant to the IPR Policy, no investigation, including IPR searches, has been carried out by. No guarantee can be given as to the existence o other IPRs not reerenced in SR (or the updates on the Web server) which are, or may be, or may become, essential to the present document. Foreword This Technical Report (TR) has been produced by Technical Committee Transmission and Multiplexing (TM). The present document is part 2 o a multi-part deliverable covering Transmission and Multiplexing (TM); Acces networks; Spectral management on metallic access networks, as identiied below: Part 1: Part 2: Part 3: "Deinitions and signal library". "Technical methods or perormance evaluations. "Construction methods or spectral management rules. NOTE: Part 3 is under preparation.

6 6 1 Scope The present document gives guidance on a common methodology or studying the impact on xdsl perormance (maximum reach, noise margin, maximum bitrate) in noisy cables when changing parameters within various Spectral Management scenarios. These methods enable reproducible results and a consistent presentation o the assumed conditions (characteristics o cables and xdsl equipment) and coniguration (choosen technology mixture and cable ill) o each scenario. The technical methods include computer models or calculating: xdsl receiver capability o detecting signals under noisy conditions; xdsl transmitter characteristics; cable characteristics cross talk cumulation in cables, originating rom a mix o xdsl disturbers; The objective is to provide the technical means or evaluating the perormance o xdsl equipment within a chosen scenario, such as calculations and measurements. This includes the description o perormance properties o equipment. Another objective is to assist the reader with applying this methodology by providing examples on how to speciy the coniguration and the conditions o a scenario in an unambiguous way. The distinction is that a coniguration o a scenario can be controlled by access rules while the conditions o a scenario cannot. Possible applications o this document include: Studying access rules, or the purpose o bounding the cross talk in unbundled networks. Studying deployment rules, or the various systems present in the access network. Studying the impact o cross talk on various technologies within dierent scenarios The scope o this Spectral Management document is explicitly restricted to the methodology or deining scenarios and quantiying the perormance o equipment within such a scenario. All judgement on what access rules are required, what perormance is acceptable, or what combinations are spectral compatible, is explicitly beyond the scope o this document. The same applies or how realistic the example scenarios are. 2 Reerences For the purposes o this Technical Report (TR) the ollowing reerences apply: SpM ISDN [1] TR " Transmission and Multiplexing (TM); Spectral Management on metallic access networks; Part 1: Deinitions and signal library V1.2.1 ( ), august [2] ANSI T1E1.4/ R6 "Spectrum Management or loop transmission systems" drat; revision 6, November 2000 (or a more recent version) [3] TS (V1.3.2): "Transmission and Multiplexing (TM); Integrated Services Digital Network (ISDN) basic rate access; Digital transmission system on metallic local lines".

7 7 HDSL SDSL ADSL [4] TS (V1.5.3): "Transmission and Multiplexing (TM); High bit-rate Digital Subscriber Line (HDSL) transmission systems on metallic local lines; HDSL core speciication and applications or combined ISDN-BA and kbit/s transmission". [5] TS , v1.1.3: "Transmission and Multiplexing (TM); Access transmission system on metallic access cables; Symmetrical single pair high bitrate Digital Subscriber Line (SDSL)". Nov [6] ITU-T Recommendation G.991.2: "Single-Pair High-Speed Digital Subscriber Line (SHDSL) transceivers". [7] TS , v1.3.1, ( ): "Transmission and Multiplexing (TM); Access transmission systems on metallic access cables; Asymmetric Digital Subscriber Line (ADSL) - European speciic requirements", may [8] ITU-T Recommendation G (1999): "Asymmetric digital subscriber line (ADSL) transceivers". 3 Deinitions and abbreviations 3.1 Deinitions For the purposes o the present document, the ollowing terms and deinitions apply: upstream transmission: transmission direction rom an NT-port to an LT-port, usually rom the customer premises, via the access network, to the telecommunication exchange downstream transmission: transmission direction rom an LT-port to an NT-port, usually rom the telecommunication exchange via the access network, to the customer premises Noise margin: the ratio by which the received noise may increase until the recovered signal does not meet the predeined quality criteria. This ratio is commonly expressed in db. Signal margin: the ratio by which the received signal may decrease until the recovered signal does not meet the predeined quality criteria. This ratio is commonly expressed in db. Max datarate: the maximum data rate that can be recovered according to predeined quality criteria, when the received noise is increased with a choosen noise margin (or the received signal is decreased with a choosen signal margin). Loop provider: company acilitating access to the local loop wiring. (In several cases the loop provider is historically connected to the incumbent network operator, but other companies may serve as loop provider as well) Network operator: company that makes use o a local loop wiring or transporting telecommunication services. This deinition covers incumbent as well as competitive network operators. Access Rule (or metallic access rule): Mandatory rule or achieving access to the local loop wiring, equal or all network operators that make use o the same network cable, that bounds the cross talk in that network cable. Deployment Rule: Voluntary rule, irrelevant or achieving access to the local loop wiring and proprietary or each individual network operator. A deployment rule relects a network operators own view about what the maximum length or maximum bitrate may be or oering a speciic transmission service to ensure a chosen minimum quality o service.

8 8 3.2 Abbreviations For the purposes o the present document, the ollowing abbreviations apply: ADSL BER CAP DMT FDD HDSL ISDN LT-port LTU NT-port NTU PAM PSD QAM REC SDSL SNR TRA VDSL xdsl 2B1Q Asymmetric Digital Subscriber Line Bit Error Ratio Carrierless Amplitude/Phase modulation Discrete Multitone modulation Frequency Division Duplexing/Duplexed High bit rate Digital Subscriber Line Integrated Services Digital Network Line Termination port (commonly at central oice side) Line Termination Unit Network Termination port (commonly at customer side) Network Termination Unit Pulse Amplitude modulation Power Spectral Density (single sided) Quadrature Amplitude modulation Receiver Symmetrical (single pair high bitrate) Digital Subscriber Line Signal to Noise Ration (ratio o powers) Transmitter Very-high-speed Digital Subscriber Line (all systems) Digital Subscriber Line 2-Binary, 1-Quarternairy (Special variant o a 4-level PAM linecode) 4 Transmitter signal models or xdsl A transmitter model in this clause is mainly a PSD description o the transmitted signal under matched conditions, plus an output impedance description to cover mis-matched conditions as well. PSD masks o transmitted xdsl signals are speciied in several documents or various purposes, or instance in Part 1 o Spectral Management [1]. These PSD masks, however, cannot be applied directly to the description o a transmitter model. One reason is that masks are speciicing an upper limit, and not the expected (averaged) values. Another reason is that the deinition o the true PSD o a time limited signal requires no resolution bandwidth at all (it is deined by means o an autocorrelation, ollowed by a Fourier transorm) while PSD masks do rely on some resolution bandwidth. They describe values that are (a slightly) dierent rom the true PSD, especially at steep edges (e.g. guard bands), and or modeling purposes this dierence is sometimes very relevant. To dierentiate between several PSD descriptions, masks and templates o a PSD are given a dierent meaning. Masks are intended or proving compliance to standard requirements, while templates are intended or modeling purposes. This clause summarizes various xdsl transmitter models, by deining template spectra o output signals. 4.1 Generic transmitter signal model A generic model o an xdsl transmitter is essentially a linear signal source. The Thevenin equivalent o such a source equals an ideal voltage source U s having a real resistor R s in series. The output voltage o this source is random in nature (as a unction o the time), is uncorrelated with any other transmitter signal, and occupies a relatively broad spectrum. This generic model can be made speciic by deining: The output impedance R s o the transmitter. The template o the PSD, measured at the output port, when terminated with an external impedance equal to R s. This is identiied as the matched condition, and under these conditions the output power equals the

9 9 maximum power that is available rom this source. Under all other (mis-matched) termination conditions the output power will be lower. 4.2 Cluster 2 transmitter signal models transmitter signal model or "ISDN.2B1Q" transmitter signal model or "ISDN.MMS.43" transmitter signal model or "Proprietary.SymDSL.CAP.QAM" 4.3 Cluster 3 transmitter signal models Transmitter signal models or "HDSL.2B1Q" Transmitter signal models or "HDSL.CAP" The PSD templates or modeling signals generated by HDSL.CAP transmitters are dierent or single-pair and two-pair HDSL systems. The PSD templates or modeling the "HDSL.CAP/2" and "HDSL.CAP/1" transmit spectra or two-pair and single-pair systems are deined in terms o break requencies, as summarized in table 1. These template are taken rom the nominal shape o the transmit signal spectra, as speciied in the HDSL standard [4] The associated values are constructed with straight lines between these break requencies, when plotted against a logarithmic requency scale and a linear dbm scale. The source impedance equals R s =135Ω. HDSL.CAP/2 2 pair HDSL.CAP/1 1- pair 135 Ω 135 Ω [Hz] [dbm/hz] [Hz] [dbm/hz] ,98 k -57 <TBD> <TBD> 21,5 k ,02 k ,58 k ,10 k ,62 k ,00 k -90 1,188 M M -120 Table 1. PSD template values at break requencies or modeling "HDSL.CAP/2" and "HDSL.CAP/1" NOTE: A PSD template or HDSL.CAP/1 is currently or urther study.

10 Transmitter signal model or "SDSL" The PSD templates or modeling the spectra o "SDSL" transmitters is deined by the theoretical sinc-shape o PAM encoded signals, plus additional iltering and a noise loor. The transmit spectrum is deined in three distinct requency bands, as summarized in expression 1 and the associated table 2. The break requency int is the requency where the curves or P 1 () and P 2 () intersect. This PSD template is taken rom the nominal shape o the transmit signal spectrum, as speciied in the SDSL standard [5]. The source impedance equals R s =135Ω. < int : K sdsl P1 ( ) = R s X 2 sinc X N H ( ) 1 ( L + ) H 2 [ W / Hz] int 1,5 MHz : P ( ) = K 2 x 0 1,5 [ W / Hz] > 1,5MHz : P ( ) = [ dbm / Hz] R s = 135 Ω sinc(x) = sin(π x) / (π x) int = is the lowest requency above H where the expressions or P 1() and P 2() intersect Parameter values are deined in table 2 Expression 1. PSD Tempate values or modeling both the symmetric and asymmetric modes o SDSL Mode Data Rate R TRA Symbol Rate sym X H L 0 N H K SDSL K X [kb/s] [kbaud] [khz] [Hz] [V 2 ] [W/Hz] Sym < 2048 both (R+ 8 kbit/s)/3 sym X/ Sym 2048 both (R+ 8 kbit/s)/3 sym X/ Asym 2048 LTU (R+ 8 kbit/s)/3 2 sym x 2/ Asym 2048 NTU (R+ 8 kbit/s)/3 sym x 1/ Asym 2304 LTU (R+ 8 kbit/s)/3 2 sym x 3/ Asym 2304 NTU (R+ 8 kbit/s)/3 sym x 1/ Table 2. Parameter values or the SDSL templates, as deined in expression Transmitter signal model or "Proprietary.SymDSL.CAP.A::Fn" Transmitter signal model or "Proprietary.SymDSL.CAP.B::Fn" Transmitter signal model or "Proprietary.SymDSL.CAP.C::Fn" Transmitter signal model or "Proprietary.SymDSL.PAM::Fn"

11 Transmitter signal model or "Proprietary.SymDSL.2B1Q::Fn" Transmitter signal model or "Proprietary.PCM.HDB3.2M.SR" Transmitter signal model or "Proprietary.PCM.HDB3.2M.SQ" 4.4 Cluster 4 transmitter signal models Transmitter signal model or "ADSL over POTS" Transmitter signal model or "ADSL over ISDN" Transmitter signal model or "ADSL.FDD over POTS" Transmitter signal model or "ADSL.FDD over ISDN" 4.5 Cluster 5 transmitter signal models Transmitter signal model or "VDSL" 5 Generic receiver perormance models or xdsl A receiver perormance model is capable o predicting up to what perormance a data stream can be recovered rom a noisy signal. In all cases it assumes that this recovery meets predeined quality criteria such as a maximum BER (Bit Error Ratio). Values like BER<10 7, during a time interval o several minutes, are not uncommon. The word perormance reers within this context to a variety o quantities, including noise margin, signal margin and max datarate. When the internal receiver noise is zero and the echo cancellation is ininite, quantities like noise margin and signal margin become equal. Perormance models are implementation and linecode speciic. Perormance modeling becomes more convenient when broken down into a cascade o smaller submodels: a linecode independent input (sub)model that evaluates the eective SNR rom received signal, received noise, and various receiver imperections.

12 12 a linecode dependent detection (sub)model that evaluates the perormance (e.g. the noise margin at speciied bit rate) rom the eective SNR. This clause describes various sub models, being used or evaluating the perormance o receivers under noise conditions. NOTE Generic models are deined with various parameters, to express various receiver properties. They include parameters to express the amount o echo suppression, receiver noise level, and SNR gap. This clause 5 is dedicated to generic perormance models only. Clause 6 is dedicated to speciic models by assigning values to all parameters o a generic model. 5.1 Generic input models or eective SNR This clause describes (sub)models or xdsl perormance that enable the description o the line code independent behavior o xdsl receivers. They describe how to evaluate the eective SNR, as intermediate result, rom various input quantities and linear imperections. When combined with a (sub)model o a line code dependent detection block a complete perormance model can be ormed (see succeeding subclauses) Linear input model or eective SNR This model is restricted to linear evaluations o the eective SNR. When non-linear behavior o the input block is relevant, such as or gain controlled analog rontends, more advanced input models may be required. received signal received noise P RS P RN Linear input model or eective SNR Receiver Eective SNR detection echo P RE 1/η e echo suppression P RN0 internal receiver noise block transmitted signal echo loss P TS Transmitter block (or opposite direction) xdsl transceiver Figure 1: Flow diagram o a transceiver model, that incorporates a linear model or eective SNR. On input, the linear input model or eective SNR requires values or signal, noise and echo. The low diagram in igure 1 illustrates this or an xdsl transceiver that is connected via a common wire pair to another transceiver (not shown). The received signal power P RS carries the data that is to be recovered. This signal originates rom the transmitter at the other side o the wire pair, and its level is attenuated by cable loss. The received noise power P RN is all that is received when the transmitters at both sides o the link under study are silent. The origin o this noise is mainly cross talk rom internal disturbers connected to the same cable (cross talk noise), and partly rom external disturbers (ingress noise). The received echo power P RE is all that is received when the transmitter at the other end o the wire pair is silent, as well as all internal and external disturbers. It is a residue that will be received when a transmitter and a receiver are combined into a transceiver en co-connected via a hybrid to the same wire pairs. When the hybrid o that transceiver is unbalanced due to mismatched termination impedances (o the cable), then a

13 13 portion (P RE ) o the transmitted signal (P TS ) will leak into the receiver which is identiied as echo. The echo loss building block models this eect. The transer unction o echo loss can be modeled by one o the models described in 7.2 (see expression 6 ), and is related to the cable characteristics and the transceiver termination impedances on both ends o the cable. On output, the linear input model or eective SNR evaluates a quantity called SNR (Signal to noise Ratio) that indicates to what degree the received signal is deteriorated by noise and residual echo. Due to signal processing by the receiver the input SNR (the ratio between signal power, and the powersum o noise and echo) will change into the eective SNR at some virtual internal point at the receiver. The eective SNR can be better or worse then the input SNR. Receivers with build-in echo cancellation can take advantage o a-priori knowledge on the echo, and can suppress most o this echo and thus improving the eective SNR. On the other hand, all analog receiver electronics produce shot noise and thermal noise, while the A/D-converter produces quantization noise. The combination o all these individual noise sources deteriorates the eective SNR. The low diagram o igure 1 illustrates how this eective SNR is evaluated by this model o the input block. It incorporates two parameters: (a) an echo suppression actor η e that indicates how eective echo cancellation is implemented, and (b) an equivalent receiver noise power P RN0 that indicates how much noise is added by the receiver electronics. This input model evaluates the eective SNR as ollows: SNR RS ( PRS, PRN, PRE, PRN 0, ηe ) = 2 In principle all parameters o the eective SNR can be assumed as requency dependent, but this dependency has been omitted here. In addition, external change o signal and noise levels will modiy the value o this eective SNR. P RN + P P RN 0 + P RE η e To simpliy urther analysis o perormance quantities like noise margin and signal margin, a short-cut is used or the eective SNR by applying dedicated oset ormats. The simpliied SNR ormula is now parameterized by a single oset parameter m and an optional requency parameter. The oset eective SNR is the eective SNR, evaluated when the received signal or the received noise power has been modiied by a actor m. The convention is that when m=1 (equals zero db) the eective oset SNR equals the eective SNR itsel. When the value o parameter m increases, the eective oset SNR decreases. Two oset ormats or this SNR are identiied in expression 2. Noise oset ormat: SNRos, N ( m, ) = SNR( PRS ( ), PRN ( ) m, PRE ( ), PRN 0 ( ), ηe ( )) Signal oset ormat: SNR ( m ) = SNR( P ( ) / m, P ( ), P ( ), P ( ), η ( )) os, S, RS RN RE RN 0 e Expression 2: Shortcuts or SNR, resulting rom the linear input model, using oset ormats. These shortcuts are used or modeling the detection block o a receiver. Mark that when the receiver noise becomes zero and the echo suppression ininite, the noise oset and signal oset ormats become the same Advanced input models or eective SNR <let or urther study> ED NOTE These input models may address imperections that cannot be represented by simple linear modelling. For example the non-linear aspects o gain controlled analog rontends

14 Generic detection models This clause identiies several generic (sub) models or the detection block: one line code independent model derived rom the Shannon capacity limit, and various line code dependent models dedicated to PAM, CAP/QAM or DMT line coding. Table 3 summarizes the naming convention or input and output quantities. Input quantities linear In db remarks Signal to Noise Ratio SNR 10 log 10(SNR) Ratio o powers (requency dependent) Output quantities Noise margin m n 10 log 10(m n) Ratio o noise powers Signal margin m s 10 log 10(m s) Ratio o signal powers Table 3. Symbols used or input and output quantities o detection models On input, the detection block requires an eective SNR, as provided by the input block. This SNR is a unction o the requency. When the oset ormat is used or describing the SNR (see expression 2), it will also be a unction o the oset parameter m. On output, the detection block evaluates a signal margin m n (or a noise margin m s when more appropriated). This margin parameter is a dominant measure or the transport quality that is achieved under noisy conditions. The Noise Margin m n indicates how much the received noise power can increase beore the transmission becomes unreliable. The Signal Margin m s indicates how much the received signal power can decrease beore the transmission becomes unreliable. Unless explicitly speciied otherwise, the word margin reers in this document to noise margin. NOTE From an xdsl deployment point o view, the analysis o noise margin is preerred over signal margin, since the (cross talk) noise is the quantity that may increase when more systems are connected to the same cable. Many xdsl implementations, however, do report margin numbers that are not exactly equal to this noise margin, since the detection circuitry cannot make a distinction between external noise (due to cross talk) and internal noise (due to imperect electronics). These margins are oten an estimate closer in value to the signal margin then the noise margin Generic Shited Shannon detection model The calculation o the margin m using the generic Shited Shannon detection model, is equivalent with solving the equation in expression 3. It has been derived rom Shannon's capacity theorem, by reducing the eective SNR ("shiting" on a db scale) by the SNR-gap Γ, to account or the imperections o practical detectors. The associated parameters are summarized in table 4. The eective SNR is to be evaluated by using one o the input models described in clause 5.1. Depending on what oset ormat is used or the SNR expression (see expression 2), the calculated margin m will represent the noise margin m n or the signal margin m s. b + B / 2 c = B / 2 c SNRos log 2 1+ Γ ( m, ) d Expression 3: Equation o the Shited Shannon detection model, or solving the margin m.

15 15 Model Parameters linear In db remarks SNR gap Γ 10 log 10(Γ) Data rate d all payload bits that are transported in 1 sec Line rate b = DateRate + overhead bitrate Bandwidth B Width o most relevant spectrum Table 4. Parameters used or Shited Shannon detection models. The various parameters used within this generic detection model are summarized in table 4. The model can be made speciic by assigning values to all these model parameters. The SNR-gap (Γ) is a perormance parameter that indicates how close the detection approaches the Shannon capacity limit. The linerate is usually higher then the data rate (0 30%) to transport overhead bits or error correction, signaling and raming. The Bandwidth is a parameter that indicates what portion o the received spectrum is relevant or data transport. The model assumes that this portion passes the receive ilters Generic PAM detection model The calculation o the margin m using the generic PAM detection model is equivalent with solving the equation in expression 4. This model assumes ideal decision eedback equalizer (DFE) margin calculations. The associated parameters are summarized in table 5. The eective SNR is to be evaluated by using one o the input models described in clause 5.1. Depending on what oset ormat is used or the SNR expression (see expression 2), the calculated margin m will represent the noise margin m n or the signal margin m s. SNR req s N H 2 b = Γ ( ) = exp ln1+ SNRos ( m, + n s ) s 0 n= N L d Expression 4: Equation o the PAM-detection model, or solving the margin m. The SNR gap Γ, being used in the above expression 4, is a combination o various eects. This Γ parameter is oten split-up into the ollowing three parts: Its theoretical value Γ PAM (in the order o 9.75 db, at BER=10 7 ) A theoretical coding gain Γ coding (usually in the order o 3-5 db), to indicate how much additional improvement is achieved by the chosen coding mechanism. An empirical implementation loss Γ impl (usually 1.6 db or more), indicating how much overall deterioration is caused by implementation dependent imperections in echo cancellation, equalization, etc, without identiying its true cause. When Γ is split-up into the above three parts, its value shall be evaluated as ollows: SNR gap (linear): Γ = Γ PAM / Γ coding Γ impl SNR gap (in db): Γ _db = Γ PAM_dB Γ coding_db + Γ impl_db

16 16 Model Parameters linear In db remarks SNR gap (eective) Γ 10 log 10(Γ) = SNR req / (2 2 b 1) SNR gap in parts: Γ PAM 10 log 10(Γ PAM) Theoretical linecode value Γ coding 10 log 10( Γ coding) Coding gain Γ impl 10 log 10( Γ impl) Implementation loss Required SNR SNR req 10 log 10(SNR req) = Γ (2 2 b 1) Data rate d all payload bits that are transported in 1 sec Line rate b = DateRate + overhead bitrate Symbol rate s = b / b Bits per symbol b = b / s (can be non-integer) Summation range N L, N H On deault: N L= 2 and N H=+1 Table 5. Parameters used or PAM detection models. The various parameters in table 5 used within this generic detection model have the ollowing meaning: The SNR-gap (Γ) and required SNR (SNR req ) are equivalent parameters and can be converted rom one to the other. The advantage o using Γ over SNR req is that Γ can be deined with similar meaning or all theoretical models in the requency domain (Shited Shannon, CAP, PAM, DMT). The advantage o using SNR req over Γ is that this quantity is closer related to the SNR observed at the decision point o the detection circuitry. The line rate is usually higher then the data rate (0 30%) to transport overhead bits or error correction, signaling and raming. The symbol rate is the line rate divided by the number o bits packed together in a single symbol. The summation range or n is rom N L to N H, and this range has to be deined to make this generic model speciic. Commonly used values or PAM, using over sampling, are N L = 2 and N H =+1. This correspond to T/3-spaced equalization. Wider ranges are not excluded Generic CAP/QAM detection model The calculation o the margin m using the generic CAP/QAM detection model is equivalent with solving the equation in expression 5. This model assumes ideal decision eedback equalizer (DFE) margin calculations. The associated parameters are summarized in table 6. The eective SNR is to be evaluated by using one o the input models described in clause 5.1. Depending on what oset ormat is used or the SNR expression (see expression 2), the calculated margin m will represent the noise margin m n or the signal margin m s. SNR req s N H b ( ) 1 Γ 2 1 = exp ln1+ SNRos ( m, + n s ) s 0 n= N L d Expression 5: Equation o the CAP/QAM-detection model, or solving the margin m. The (eective) SNR gap Γ, being used in the above expression 5, is a combination o various eects. This Γ parameter is oten split-up into the ollowing three parts: Its theoretical value Γ CAP (in the order o 9.8 db or BER=10-7 ) A theoretical coding gain Γ coding (usually in the order o 3-5 db), to indicate how much additional improvement is achieved by the chosen coding mechanism. An empirical implementation loss Γ impl (usually 1.6 db or more), indicating how much overall deterioration is caused by implementation dependent imperections in echo cancellation, equalization, etc, without identiying its true cause.

17 17 When Γ is split-up into the above three parts, its value shall be evaluated as ollows: SNR gap (linear): Γ = Γ CAP / Γ coding Γ impl SNR gap (in db): Γ _db = Γ CAP_dB Γ coding_db + Γ impl_db Model Parameters linear In db remarks SNR gap (eective) Γ 10 log 10(Γ) = SNR req / (2 b 1) SNR gap in parts: Γ CAP 10 log 10(Γ PAM) Theoretical linecode value Γ coding 10 log 10( Γ coding) Coding gain Γ impl 10 log 10( Γ impl) Implementation loss Required SNR SNR req 10 log 10(SNR req) = Γ (2 b 1) Data rate d all payload bits that are transported in 1 sec Line rate b = DateRate + overhead bitrate Symbol rate s = b / b Bits per symbol b = b / s (can be non-integer) Summation range N L, N H On deault: N L=0 and N H=+3 Table 6. Parameters used or CAP/QAM detection models. The various parameters in table 6 used within this generic detection model have the ollowing meaning: The SNR-gap (Γ) and required SNR (SNR req ) are equivalent parameters and can be converted rom one to the other. The advantage o using Γ over SNR req is that Γ can be deined with similar meaning or all theoretical models in the requency domain (Shannon, CAP, PAM, DMT). The advantage o using SNR req over Γ is that this quantity is closer related to the SNR observed at the decision point o the detection circuitry. The line rate is usually higher then the data rate (0..30%), to transport overhead bits or error correction, signaling and raming. The symbol rate is the line rate divided by the number o bits packed together in a single symbol. The summation range or n is rom N L to N H, Commonly used values or CAP/QAM systems using over sampling are N L =0 and N H =+3. This holds when the carrier requency positions the spectrum low in the requency band (e.g. CAP-based HDSL). Other values may be more appropriated when the carrier requency moves the spectrum to higher requencies (e.g CAP based VDSL) Generic DMT detection model <let or urther study> 6 Speciic receiver perormance models or xdsl This clause 6 deines parameter values or the generic perormance models o the previous clause 5, to provide speciic models or various xdsl modems. ED NOTE This will be the main portion o the document. The validity o each model that get the predicate compliant must be demonstrated by showing how close it can predict the perormance requirements speciied in the associated xdsl standard. For instance SDSL: Gap=6.6 db, Echo=-50dB, Noise=-110 dbm, BitDensity=3 bits/symbol, Overhead=, etc.

18 Receiver perormance model or "HDSL.2B1Q" 6.2 Receiver perormance model or "HDSL.CAP" 6.3 Receiver perormance model or "SDSL" 6.4 Receiver perormance model or "ADSL over POTS" 6.5 Receiver perormance model or "ADSL.FDD over POTS" 6.6 Receiver perormance model or "ADSL over ISDN" 6.7 Receiver perormance model or "ADSL.FDD over ISDN 6.8 Receiver perormance model or "VDSL" 7 Transmission and relection models 7.1 Summary o test loop models ED NOTE This clause reers to various testloops or ADSL, SDSL, VDSL, as deined in published documents like standards. I required reerences to additional cable models can be added, but when possible we should try to keep this clause as short as possible. In practice, each country will avor its own cable models, and they are too numerous (and too proprietary) to mention them all here. 7.2 Basic model or echo loss A model or echo loss describes a property o the hybrid in a transceiver, and models what portion o the transmitted signal relects directly into the receiver. When the hybrid is perectly balanced, no echo will low into the receiver. When the cable impedance diers rom the value where the hybrid is designed or, the hybrid will be out o balance and some transmitted signal relects into the receiver. The basic model or echo loss assumes that (a) the output impedance o the transceiver equals some value R v, that (b) the hybrid is balanced when terminated with a load impedance Z L equal to R v, and that the hybrid can be represented by a Wheatstone bridge. This is illustrated in igure 2. The associated transer unction H E is speciied in expression 6.

19 19 R R V U S R U E + + U T Z L U U E T U = T U U T s / 2 = RV + Z Z L L Z L R = 2 Z L V Figure 2: Flow diagram o the basic model or echo loss H E Z L ( jω) R ( jω) = 2 Z ( jω) L V P P RE = TS H E ( jω) 2 Expression 6: Transer unction o the basic model or echo loss. The identiiers P RE and P TS reer to power low values used in igure 1. When using this basic model or echo loss in a ull simulation, value R V can be made equal to the design impedance o the modem under test, and value Z L can be made equal to the complex and requency dependent input impedance o the cable, terminated at the other cable end with a load impedance equal to R V. 8 Cross talk models Cross talk models account or the act that the transmission is impaired by cross talk originated rom discrete disturbers distributed over the local loop wiring. In practice this is not restricted to a lineair cable topology, since wires may an out into dierent directions to connect or instance dierent customers to a central oice The most simple topology models assume that all disturbers are co-located at only two locations; one at each end o a cable. These approximations may be adequate or situations above or instance 1 km in which the an out o the wires can be ignored. More advanced topology models require a multi-node co-location approach. An example is the insertion o repeaters, that introduces co-located disturbers in-between. Another example is deploying VDSL rom the cabinet or the situation that all customers are distributed along the cable. This clause summarizes dierent cross talk models or dierent topologies, sorted by complexity, and provide several cross talk models to predict how much noise is coupled into a victim wire pair. 8.1 Overview o dierent network topologies 8.2 Validity limitations o cross talk modeling

20 Generic cross talk models or two-node co-location The cross talk models in this sub clause apply to scenarios in which it can be assumed that all customers are virtually co-located, and that they are all served rom the central oice. The result is that such a cross talk model requires only two nodes (one on the LT side, and another one on the common NT side). These nodes are interconnected by means o a multi wire pair cable. Cross talk models are built up rom several building blocks, and the way these blocks are interconnected is deined by means o a topology diagram Basic diagram or two-node topologies The basic low diagram or describing a topology in which xdsl equipment is assumed to be co-located at two nodes (the two ends o a cable) is shown in igure 3 and 4. Up and downstream perormance are evaluated separately. The approach o this diagram can be described in three distinct steps. The diagram combines or each node the output disturbance o individual disturbers (P d1, P d2, ) by modeling cross talk cumulation as an isolated building block. This is because the cumulation rom dierent disturbers cannot be modeled by a simple linear power sum o all individual disturbers. Since each wire pair couples at dierent ratio to the victim wire pair, the cumulation requires some weighed power sum that accounts or the statistical distribution o all involved cross talk coupling ratios. By modeling cross talk cumulation as an isolated building block, the cumulated disturbance can be thought as i it was virtually generated by a single equivalent disturber (P d.eq ). This has been indicated in igure 3 and 4 by a box drawn around the involved building blocks. Using the equivalent disturber concept as intermediate yields an elegant concept to break down the complexity o a ull noise scenario into smaller pieces. Next, the diagram evaluates what noise level (P XN ) is coupled into the victim wire pair. Figure 3 and 4 illustrate what portion o the equivalent disturbance is coupled into the victim wire pair by using models or NEXT and FEXT. On top o this, background noise (P bn ) can be added to represent all remaining unidentiied noise sources. Since it is a generic diagram, the power level o this background noise level is let undeined here, but commonly used values are zero, or levels as low as P bn = 140 dbm/hz. When all building blocks are modeled or the same impedance as implemented in the modem under study, the noise level (P RN ) received by the modem under test equals the level derived so ar (P XN ). In practice, these models are normalized at some chosen reerence impedance R n, and this R n may be dierent rom the impedance implemented in the modem under study (targeted at its design impedance R V ). This mismatch will cause a change in the level o the disturbance, and this eect is modeled by the noise injection building block. The succeeding clauses summarizes some generic models or the individual building blocks o igure 3 and 4. The transer unctions H next and H ext o the building blocks or NEXT and FEXT are linear and requency dependent. The model or the topology assumes that all disturbers are uncorrelated, which causes that the cross talk power P XN behind the summation block is the sum o all individual powers. This transer unctions are speciied in expression 7. P P XN, NT XN, LT = = P d. eq, NT P d. eq, LT H H next 2 2 next + + P P d. eq, LT d. eq, NT H H 2 ext 2 ext + + P bn, NT P bn, LT Expression 7: Evaluation o the cross talk power levels, that low into the noise injection blocks o the two-node topology models in igure 3 and 4.

21 21 modem under study P RN,NT noise injection model TRANSFER (INSERTION LOSS) victim wire pair modem under study Background Disturber P XN,NT FEXT model P bn,nt NEXT model P d.eq,nt P d.eq,lt NT-side crosstalk cumulation model upstream crosstalk cumulation model LT-side P d1,nt P d2,nt P d3,nt mixture o xdsl disturbers equivalent disturber at NT side downstream equivalent disturber at LT side mixture o xdsl disturbers P d1,lt P d2,lt P d3,lt Figure 3: Flow diagram o the basic model or two-node topologies, or evaluating downstream perormance modem under study TRANSFER (INSERTION LOSS) victim wire pair noise injection model P RN,LT modem under study FEXT model P XN,LT Background Disturber NEXT model P bn,lt P d.eq,nt P d.eq,lt NT-side crosstalk cumulation model upstream crosstalk cumulation model LT-side P d1,nt P d2,nt P d3,nt mixture o xdsl disturbers equivalent disturber at NT side downstream equivalent disturber at LT side mixture o xdsl disturbers P d1,lt P d2,lt P d3,lt Figure 4: Flow diagram o the basic model or two-node topologies, or evaluating upstream perormance

22 Models or cross talk cumulation The noise that couples into a victim wire pair, and originates rom several co-located disturbers connected to dierent wire pairs, cumulate in level. This cumulation cannot be modeled by a simple linear power sum o all individual disturbers, because each wire pair couples at dierent ratio to the victim wire pair. Thereore the cumulation requires some weighed power sum that accounts or the statistical distribution o all involved cross talk coupling ratios. On input, the cumulation building block requires the levels (P d1 P dn ) o all involved individual disturbers that are colocated. On output, the cumulation building block evaluates the level o the equivalent disturbance (P d.eq ). This sub clause provides expressions to model building blocks or cross talk cumulation FSAN sum or cross talk cumulation The FSAN sum is one o the possible expressions to model cross talk cumulation, and is speciied in expression 8. The (requency dependent) power level o the equivalent disturbance, that cumulates rom M individual disturbers, is expressed below. The actor K n weighs this sum when K n 1. For K n >1 the FSAN sum results in a power level that s is always equal or less then the linear sum (K n ) o these powers. This actor is cable dependent, and assumed to be requency independent. Values ranging between K n =1/0,6 and K n = 1/0,8 have been observed in practice. On deault, K n =1/0,6 is commonly used, but this parameter must be explicitly speciied when using this model or cross talk cumulation in a perormance evaluation. P Kn Kn Kn K 1 K n n ( P + P + P + P ) d. eq = d1 d 2 d 3 L + Expression 8: FSAN sum or cumulating the power levels o M individual disturbers into the power level o an equivalent disturber In the special case that all M disturbers generates equal power levels (P d ) at all requencies o interest, the FSAN sum simpliies into P d.eq = P d M 1/Kn. The FSAN sum ignores dierences in source impedances o dierent disturber types. For cumulating disturbance rom sources with dierent impedances, their available power levels are to be combined according to the FSAN sum. This available power o a source is the power dissipated in a load resistance, equal to the source impedance. dm Models or cross talk coupling The spread in cross talk coupling between wire pairs in a real twisted pair cable is signiicant, and the coupling luctuates rapidly when the requency increases. The cross talk rom a single disturber is thereore random in nature. When the number o co-located disturbers increases, the luctuations reduce signiicantly. Models or cross talk coupling take advantage o this eect and their simplicity increases when the number o co-located disturbers increases. Equivalent cross talk coupling o a cable is the ratio between the level o the cross talk in the victim wire pair and the level o an equivalent disturber evaluated by some cross talk cumulation model, while connecting as much individual disturbers as possible to the cable under study. This cross talk sum will be dierent or each wire pair, due to the random nature o the coupling. Commonly accepted models or equivalent cross talk coupling represent 99% o the victim wire pairs. This is to approximate 100% o the cases, without being pessimistic or the very last extreme 1% case. This sub clause provides expressions to model the building blocks or equivalent cross talk coupling.

23 Basic models or equivalent NEXT and FEXT Expression set 9 speciies how to model the transer unctions o the equivalent NEXT and FEXT building blocks. The speciication is based on the ollowing constants, parameters and unctions: Variable identiies the requency. Constant 0 identiies a chosen reerence requency, commonly set to 0 = 1 MHz. Variable L identiies the physical length o the cable between the two nodes in meters. Constant L 0 identiies a chosen reerence length, commonly set to L 0 = 1 km. Function s T (, L) represents the requency and length dependent amplitude o the transmission unction o the actual test loop, normalized to a reerence impedance R n. This value equals s T = s 21, where s 21 is the transmission s- parameter o the loop normalized to R n This R n is commonly set to 135Ω. Constant K xn identiies an empirically-obtained number that scales the NEXT transer unction H next (, L). Constant K xn identiies an empirically-obtained number that scales the FEXT transer unction H ext (, L). H H next ext (, L) (, L) = = K K xn x s L / L T 0 (, L) s T 4 (, L) Expression 9: Transer unctions o the basic models or NEXT and FEXT Models or cross talk injection Several sub models or various building blocks within the cross talk model ignore the act that when the modem and cable impedance will change, the noise (and signal) observed by the receiver will change as well. For instance, when the input impedance (Z xdsl ) o the receiver under test decreases, the received noise level will decreases as well. To account or this eect, a cross talk injection block is included in the topology models in igure 3 and 4. The transer unction o the cross talk injection block identiied as H xi, and is requency and impedance dependent. Expression 10 illustrates how to use this transer unction or evaluating the power level P RN rom power level P XN. P = P RN XN H xi 2 Expression 10: Evaluation o the receive noise level rom the cross talk noise level under matched conditions, by a transer unction o the noise injector. A transer unction that models the impact o impedance mismatch can be very complex, and thereore several simpliied transer unctions are commonly used to approximate this eect. This sub clause summarize a ew o these approximations Forced noise injection When cross talk is modelled by means o orced noise injection, then all impedance and requency dependency o noise injection is ignored. The associated transer unction is shown in expression 11. H xi ( ) = 1 Expression 11: Transer unction or orced noise injection.