IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 1

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1 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION Self-Tuning Variable Frequency Controller for Inductive Electric Vehicle with Multiple Power Levels Masood Moghaddami, Student Member, IEEE, and Arif Sarwat, Member, IEEE Abstract A self-tuning controller for contactless electric vehicle (EV) charging systems based on inductive power transfer (IPT) with multiple power levels is proposed. The multiple charging levels (consisting of charging levels) are achieved by controlling the energy injection frequency of the transmitter coil of the inductive power transfer (IPT) system. The proposed controller is capable of self-tuning the operations to the natural resonance frequency of the IPT system and benefits from soft- operations (zero-current ), which ensures the maximum performance of the IPT system. The proposed controller has such a simple design which can be implemented based on a simplified control circuit. The simulation of the proposed controller for an inductive charging system at different charging levels is carried out in MATLAB/Simulink. Also, the proposed controller with an AC/DC/AC converter is implemented experimentally on an IPT charging system to verify the effectiveness of the controller at different charging levels. The experimental test results conform with the simulation results and verify that the proposed controller effectively enables self-tuning capability and soft- operations at different charging levels for IPT based contactless EV charging systems. Index Terms inductive power transfer, multi-charging-level, self-tuning control, soft-, variable frequency control. I. INTRODUCTION CONTACTLESS electric vehicle (EV) charging based on inductive power transfer (IPT) systems is a new technology that brings more convenience and safety to the use of EVs. Since it eliminates the electrical contacts, it would not get affected by rain, snow, dust and dirt, it is a safe, reliable, robust and clean way of charging electric vehicles, reduces the risk of electric shock. Therefore, it has recently found a significant interest in residential and commercial sectors [] [5]. Contactless EV charging is divided into two main categories: static charging and dynamic (in-motion) charging [6] []. In the static charging the goal is to charge the EV using a contactless charger while the EV is parked in a charging station, which can be used for light-duty or heavy-duty EVs [2]. This is a solution that enables safe, efficient, and convenient automated charging process without the interaction of the driver. On the other hand, a dynamic EV charging Manuscript received July 3, 26; revised September 24, 26; accepted November 27, 26. The authors are with the Department of Electrical and Computer Engineering, Florida International University, Miami, FL, 3374 USA. The authors are with the Department of Electrical and Computer Engineering, Florida International University, Miami, FL, 3374 USA ( mmogh3@fiu.edu; asarwat@fiu.edu) This work was funded by the National Science Foundation under grant number CAREER Transmitter Pad Secondary Compensation AC/DC Converter Primary -85kHz Compensation Receiver Pad Battery Storage System AC/AC Converter 5/6Hz Three-phase mains Fig.. Typical inductive electric vehicle charging system. system is designed to charge the EVs while they are moving. A typical static inductive EV charging system is shown in Fig.. This system is composed of a power supply, primary and secondary converters, transmitter and receiver pad structures, and compensation circuits. Different control methods for IPT systems and resonant converters have been proposed. These methods include powerfrequency control [3], [4], phase-shift and frequency control [5] [8], load detection [9], power flow control [2], and sliding mode control (SMC) [2] [24]. In [7], a selfoscillating phase-shift control for regulating the IPT systems is proposed, which matches the operation to the resonant current. The phase-shift is achieved by the use of a tunable leading phase-shifter in the current sensing circuit. However, the phase-shift control method leads to deviation of the operations from soft- points (zerocurrent and zero-voltage points), which may adversely affect the efficiency of the converter and significantly increase its electromagnetic interference (EMI). The self-tuning capability is specifically essential for dynamic IPT systems where the resonance frequency of the system may have small variations due to load variations on the receiver side. Furthermore, it can find many other applications such as energy encryption in IPT systems [25], where the IPT system has a variable resonance frequency with the use of variable compensation capacitors. One of the methods that can be employed to effectively adjust the resonant current in an IPT system is the energyinjection and free-oscillation method which has been successfully employed in many studies [26] [32]. This method can be employed in different converter topologies such as twostage AC/DC/AC converters [3] and matrix converters [29], (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

2 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 2 [3]. Using this method, the resonant current is adjusted by controlling the energy injection rate which is transferred to the primary LC tank. This is achieved by constantly between two free-oscillation and energy-injection operation modes of the converter. In this paper, a new variable frequency controller based on energy-injection free-oscillation technique for IPT systems is proposed. This controller provides multiple charging levels for inductive EV charging applications. In this study, the controller is designed for three-phase to single-phase fullbridge two-stage AC/DC/HFAC (high-frequency AC) converters. However, the proposed control method can be applied to other types of converter topologies. The controller enables contactless charging of an EV at multiple charging levels based on a user-defined preset. The charging levels include the standard wireless charging levels for light-duty EVs (Levels, 2, 3 and 4) as defined by SAE TIR J2954 [33]. Also, the proposed controller is capable of self-tuning the operations to the resonance frequency of the IPT system and benefits from zero-current (ZCS), which ensures the maximum power transfer efficiency. The proposed controller can be implemented either based on a digital or analog control circuit. MATLAB/Simulink is used to simulate the proposed controller at different charging levels and the results are presented. Also, the proposed controller is built and tested experimentally to evaluate the effectiveness of the proposed controller at different charging levels. II. THE PROPOSED CONTROLLER Using the energy injection method the amplitude of the resonant current in an IPT system can be controlled by varying the rate of the energy injection to the primary coil. On this basis, a self-tuning variable frequency energy injection control 3~ input S sgn S sgn!q Q + _ Rectifier Multiport Switch Clk< D Multiport Switch 2 N 2 N2 3!Q Q 2 N3 4 N4 select 4 P 3 P2 P3 P4 select Clk< D!Q Q Clk< D NOT S sgn + _ zero-cross detector i r GND Levels Fig. 2. The proposed simplified control circuit designed for a two-stage fullbridge AC/DC/HFAC converter. V dc V dc V dc Mode Mode 2 Mode 3 V dc Mode 4 Fig. 3. The resonant current path in four modes of operation. Modes and 2 are energy injection modes and modes 2 and 3 are free oscillation modes. TABLE I FOUR OPERATION MODES OF A FULL-BRIDGE INVERTER CORRESPONDING SWITCHING STATES. Mode Type Current direction (S sgn) injection (S inj) mode injection mode 2 injection mode 3 Free oscillation mode 4 Free oscillation method for IPT systems is proposed. The proposed method employs a variable frequency energy injection to the IPT system to achieve multiple charging levels. Fig. 2 shows the proposed controller which is designed for a two-stage AC/DC/HFAC converter. The two-stage converter is composed of a three-phase rectifier and a single-phase full-bridge highfrequency inverter. The operation of the full-bridge inverter can be described in four modes based on the direction of the resonant current (positive or negative) and the type of the (energy injection or free-oscillation) operation mode, as it is presented in Fig. 3 and Table I. Based on Table I, four signals of the full-bridge inverter,,, and can be expressed in terms of current direction (S sgn ) and energy injection (S inj ) states as follows: = S sgn, = S sgn, = S sgn S inj = S sgn S inj () Based on (), regardless of the type of operation mode (whether it is an energy injection or free-oscillation mode) is switched ON when the direction of current is positive, and is switched ON when the direction of current is negative. Furthermore, () shows that is switched ON in positive energy injection modes and is switched ON in negative energy injection modes. The presented controller is designed based on a simplified circuit and it is composed of D-type flip-flops, logic gates, a differential comparator, and multi-port switches. The sign of the resonant current (S sgn ) which has the resonance frequency (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

3 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 3 Current P P2 P3 P4 N N2 N3 N4 Fig. 4. The energy injection signals with frequency of f r (resonance frequency), f r/2, f r/4 and f r/8 for positive (P, P 2, P 3 and P 4 ) and negative (N, N 2, N 3 and N 4 ) half-cycles which are generated generated by the proposed controller. Free Oscillation Free Oscillation Free Oscillation Current Inverter output voltage Free Oscillation Fig. 5. Typical waveforms of the resonant current and the converter output voltage applying the proposed control method for charging Level 2-4. f r, is determined by the zero-cross detector which is the differential comparator. Based on (), is connected to S sgn and is connected to S sgn as shown in Fig 2. The series flip-flops form three frequency dividers and by using the sign signal S sgn as the clock source, signals with f r /2, f r /4 and f r /8 frequencies are generated. By combining the sign signal S sgn with the signals generated by the frequency divider with the use of and NOT gates, four signals for positive energy injection half-cycles (P,P 2,P 3,P 4 ) and four signals for negative energy injection halfcycles (N,N 2,N 3,N 4 ) are generated. Based on (), since P, P 2, P 3 and P 4 are energy injection signals in positive halfcycles, can be connected to any of them, enabling energy injection to the IPT system at different rates. Similarly, based on (), since N, N 2, N 3 and N 4 energy injection signals in positive half-cycles, can be connected to any of them, enabling energy injection to the IPT system at different rates. Therefore, the multi-port switches are employed to connect and to the proper energy injection signals based on a user-defined charging level preset. In Fig. 4, typical energy injection signals for positive and negative half-cycles are presented. Using two multi-port switches in time the proposed control circuit the frequency of energy injection in positive and negative half-cycles of the resonant current can be controlled separately. The energy injection frequency in positive and negative half-cycles can be f r (resonance frequency), f r /2, f r /4 and f r /8. Thereby, enabling energy injection to the IPT system at distinct levels. These charging levels are presented in Table II. Although, these 6 (4 4) charging levels can be reduced to distinct charging levels as there are 6 charging levels are equivalent to the other charging levels which are given in Table II, e.g. Level -2 is equivalent to Level 2-, and Level -4 is equivalent to Level 4-, etc. Typical waveforms of the resonant current and output voltage of a full-bridge inverter with the use of energy injection control method is shown in Fig. 5. As it can be seen, each mode starts and ends at zero-crossing points, which leads to zero-current current (ZCS) of the inverter. Therefore, the converter will have higher efficiency compared to conventional converters. Also, the simple design of the controller not only simplifies its implementation but also enables higher operating frequencies, at which conventional digital controllers (DSP/FPGA) may not be able to achieve. In charging levels with different energy injection frequencies for positive and negative half-cycles, the output voltage of the full-bridge inverter will have a DC component. Since the impedance of the series RLC circuit (shown in Fig. 2), for a DC input is infinite (the capacitor acts as an open circuit). In other words, the DC component of the voltage is eliminated. As a result, the output resonant current will only include harmonic (non-dc) components, without any DC component, which will result in a symmetrical resonant current. In an IPT system if the converter operates at the resonance frequency a perfect matching will be obtained, as the impedance of the compensation capacitors cancels out the equivalent impedance of the transmitter and receiver coils, resulting in a purely resistive network; Thereby, there would be no phase difference between the resonant current and input voltage. As a result, the zero-current (ZCS) will ensure the zero-voltage (ZVS). Since using the proposed controller the operations are synchronized with the resonant current and therefore, the ZCS and ZVS are both achieved. However, if the resonance frequency of the primary and secondary are a little different due to imperfect resonance matching, the equivalent circuit would not be purely resistive. Since the proposed controller uses zero-current crossing points for generating the signals, in such a case only ZCS will be achieved. III. THEORETICAL ANALYSIS In this section theoretical calculation of different measures for the contactless charging system based on the proposed controller is presented. Analytical solutions for the resonant voltage, resonant current and the output power can be found as follows: A. Current and Voltage Calculation The differential equation of a series compensated IPT system can be expressed based on the resonant current i as (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

4 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 4 follows: d 2 i dt 2 + L di dt + L C i = (2) where L is the self-inductance of the transmitter, and C is the compensation capacitor of the transmitter, and is the equivalent resistance of the load on the receiver side, reflected to the transmitter side. At each zero-crossing point of resonant current, the initial conditions of the circuit are as follows: i =, L di dt () = V t v c (3) where i is the initial resonant current and V t is the output voltage of the inverter. The resonant current and the resonant voltage (v c ) of the compensation capacitor can be found by solving (2), based on the initial conditions given in (3) [3]: i(t) = Ke t/τ sin(ωt) (4) v c (t) = v c + Kτ ( ) C( + τ 2 ω 2 τω e t/τ [sin(ωt) + τωcos(ωt)] ) where τ = 2L/R is the damping time constant, ω = ω 2 α 2 is the natural damped frequency, ω = / LC is the resonant frequency, α = /2L is the damping coefficient, and the coefficient K is expressed as: K = V t v c (6) ωl In order to calculate the the initial condition for the resonant voltage v c at each current zero-crossing in a steady-state condition, a full control cycle consisting of 2n half-cycles of the resonant current which includes one energy injection halfcycle (Fig. 5) is considered. The resonant voltage at the end of the energy injection half-cycle (t = π/ω) can be calculated using (5) as follows: where β is defined as: (5) v c = V t + β (V t v c ) (7) β = e π τω (8) At the end of free-oscillation half-cycles (half-cycles from 2 to 2n), the resonant voltage can be calculated based on (5) as follows: v ck = v c β k ( ) k (9) TABLE II DIFFERENT CHARGING LEVELS WHICH CAN BE ACHIEVED USING THE PROPOSED CONTROLLER. Frequency of energy injection Level Positive half-cycles Negative half-cycles Level - f r f r Level -2 f r f r/2 Level -4 f r f r/4 Level -8 f r f r/8 Level 2-2 f r/2 f r/2 Level 2-4 f r/2 f r/4 Level 2-8 f r/2 f r/8 Level 4-4 f r/4 f r/4 Level 4-8 f r/4 f r/8 Level 8-8 f r/8 f r/8 TABLE III THE SIMULATION RESULTS AT CHARGING LEVEL-, -2, Level RMS THD of Current (%) Output Power (kw) Level Level Level Level Using (7), equation (9) can be rewritten as follows: v ck = V t ( + β)β k ( ) k + v c β k () By assuming that the system has reached a steady-state condition, it can be concluded that the resonant voltage at the beginning of each control cycle (v c at k = ) should be equal to the its value at the end of the control cycle (v ck at k = 2n). Therefore, using () the following equations can be derived: v c = V t ( + β)β 2n + v c β 2n () ( + β) v c = β 2n β2n V t (2) Equation (2) is the initial condition for the resonant voltage in the steady-state condition and can be used in (6), (5) to calculate the resonant current and the resonant voltage at any time. B. Output Power Calculation The maximum output power of the converter is achieved when the controller is set to level -. In this case, all of the half-cycles of the resonant current would be in energy injection mode. Using the same method for calculation of the initial condition for the resonant voltage in steady-state conditions which is presented in section III.A, the initial condition for the resonant voltage can be calculated as follows: v c = + β β V t (3) Using (6) and (3), the resonant current i for any half-cycle can be written as follows: i = 2V t ωl( β) e t/τ sin(ωt) (4) The output power can be calculated using (4) as follows: P = π/ω V t 2V t ωl( β) e t/τ sin(ωt) dt π/ω P = 2V 2 t τ 2 ω ( + e π/τω) πl( β)( + τ 2 ω 2 ) (5) (6) Using (6), the output power can be calculated based on the input voltage and the circuit parameters (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

5 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION ,.5, ,.5, Voltage (V) 2-2 Voltage (V) Time (µs) (a) Time (µs) (b) ,.5, ,.5, Voltage (V) Time (µs) (c) Voltage (V) Time (µs) (d) Fig. 7. The resonant current, inverter output voltage, signals and battery charging current in different charging levels: (a) Level - charging with 4 kw output power, (b) Level -2 charging with 22. kw output power, (c) Level 2-2 charging with 2.3 kw output power, (d) Level 4-8 charging with 3.6 kw output power. IV. SIMULATION RESULTS An inductive EV charging system based on the proposed self-tuning controller which is shown in Fig. 2, is simulated 3~ mains + _ Rectifier Switching Signals Magnetic structure and Compensations Self-tuning Controller M i r C s Level Settings EV Battery Charger Battery Fig. 6. Inductive electric vehicle battery charging model with the proposed controller simulated in MATLAB/Simulink. in MATLAB/Simulink. The simulation model is presented in Fig. 6 and it is composed of a three-phase power supply, a fullbridge three-phase rectifier, a full-bridge single-phase inverter which is switched by the proposed controller, transmitter and receiver coils with their compensation capacitors, and a battery charger for an EV at the receiver. The self-inductances of the transmitter and receiver each are assumed to be 72 µh, with compensation capacitors of.2 µf. Therefore, the resonance frequency of the IPT system is 35 khz. The three-phase power supply is assumed to have a line-to-line voltage of 28 V and 6 Hz power frequency. The EV battery is modeled as a 36 V battery with 22 kwh capacity. The secondary converter is modeled as a conventional two-stage (AC/DC/DC) battery charger that can operate at constant current (CC) mode at a desired charging power level. The simulations are carried out in four charging levels (levels -, -2, 2-2 and 4-8 according to Table II) and the results are presented in Table III. Also, the resonant current, inverter output voltage, battery charging current and (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

6 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 6 Receiver pad Therefore, the proposed controller effectively self-tunes the operations with the resonant current and achieves soft- operations at each charging level. Transmitter pad Three-phase input Three-phase AC/DC/AC Controller board converter Fig. 8. The case study IPT setup consisting of two circular transmitter and receiver power pads, an AC/DC/AC converter as the primary converter and the proposed controller. TABLE IV EXPERIMENTAL TEST RESULTS ON THE CASE STUDY IPT SYSTEM USING THE PROPOSED CONTROLLER AT DIFFERENT CHARGING LEVELS. Frequency of energy injection Level Positive half-cycles Negative half-cycles current (A) Output power (W) Level - f r f r Level -2 f r f r/ Level -4 f r f r/ Level -8 f r f r/ Level 2-2 f r/2 f r/ Level 2-4 f r/2 f r/ Level 2-8 f r/2 f r/ Level 4-4 f r/4 f r/ Level 4-8 f r/4 f r/ Level 8-8 f r/8 f r/ corresponding signals are shown in Fig. 7. As it is shown, using the proposed controller, different charging levels can be achieved with low harmonic distortions (THD) in the resonant current. It is important to note that the charging levels -2 and 4-8 respectively correspond to the charging levels 4 and, as defined by SAE TIR J2954 standard [33]. V. EXPERIMENTAL ANALYSIS The proposed controller for IPT systems is implemented experimentally and tested on an IPT system. In Fig. 8, the IPT test-bed which is comprised of two circular transmitter and receiver power pads, compensation capacitors, a three-phase AC/DC/AC converter as the primary converter connected to the proposed controller, and a variable load at the secondary. The self-inductance of the power pads is 72 µh, and each pad has a.2 µf compensation capacitor. As a result, the resonance frequency of the IPT system would be 35 khz. The three-phase input of the rectifier is connected to a three-phase power supply with a reduced line-to-line voltage of 25 V, with 6 Hz frequency. The proposed controller is tested at multiple charging levels according to Table II and the results are presented in Table IV. The resonant current and the energy injection signals for each charging level are shown in Fig. 9. This figure shows that due to the self-tuning capability of the converter, the operations are all synced with the resonant current (at 35 khz resonance frequency). Also, Fig. 9 verifies the ZCS operations which occur at the current zero-crossing points. VI. CONCLUSION This paper has introduced a controller that can be used in inductive EV charging systems to achieve multiple charging levels. The proposed controller has user-defined charging levels and is capable of self-tuning the operations of the converter to the resonance frequency of the IPT system, and therefore eliminates the need for frequency tuning. Also, it enables soft- operations (ZCS) in the converter, which will result in a significant increase in the efficiency of the power electronic converter. Furthermore, the implementation of the proposed controller based on a simplified circuit eliminates the need for DSP/FPGA based solutions and enables high operating frequencies which are usually required in IPT systems. The experimental test results on the IPT test-bed conform with the simulation results and verify the effectiveness of the proposed controller at different charging levels. 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7 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 7 (a) (b) (c) (d) (e) (f) (g) (h) (i) (j) Fig. 9. Experimental test results on the IPT system setup: the resonant current and energy-injection signals at different charging levels: (a) Level -, (b) Level -2, (c) Level -4, (d) Level -8, (e) Level 2-2, (f) Level 2-4, (g) Level 2-8, (h) Level 4-4, (i) Level 4-8, (j) Level 8-8. [3] U. K. Madawala, M. Neath, and D. J. Thrimawithana, A powerfrequency controller for bidirectional inductive power transfer systems, IEEE Trans. Ind. Electron., vol. 6, no., pp. 3 37, Jan 23. [4] Z. U. Zahid, Z. M. Dalala, C. Zheng, R. Chen, W. E. Faraci, J. S. J. Lai, G. Lisi, and D. Anderson, Modeling and control of seriesseries compensated inductive power transfer system, IEEE Journal of Emerging and Selected Topics in Power Electronics, vol. 3, no., pp. 23, March 25. [5] A. Berger, M. Agostinelli, S. Vesti, J. A. Oliver, J. A. Cobos, and M. Huemer, A wireless charging system applying phase-shift and amplitude control to maximize efficiency and extractable power, IEEE Trans. Power Electron., vol. 3, no., pp , Nov 25. [6] J. T. Matysik, The current and voltage phase shift regulation in resonant converters with integration control, IEEE Trans. Ind. Electron., vol. 54, no. 2, pp , April 27. [7] A. Namadmalan, Self-oscillating tuning loops for series resonant inductive power transfer systems, IEEE Trans. Power Electron., vol. 3, no., pp , Oct 26. [8] A. Namadmalan and J. S. Moghani, Tunable self-oscillating technique for current source induction heating systems, IEEE Trans. Ind. Electron., vol. 6, no. 5, pp , May 24. [9] Z. H. Wang, Y. P. Li, Y. Sun, C. S. Tang, and X. Lv, Load detection model of voltage-fed inductive power transfer system, IEEE Trans. Power Electron., vol. 28, no., pp , Nov 23. [2] J. M. Miller, O. C. Onar, and M. Chinthavali, Primary-side power flow control of wireless power transfer for electric vehicle charging, IEEE (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

8 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TTE , IEEE IEEE TRANSACTIONS ON TRANSPORTATION ELECTRIFICATION 8 Journal of Emerging and Selected Topics in Power Electronics, vol. 3, no., pp , March 25. [2] F. F. A. van der Pijl, M. Castilla, and P. Bauer, Adaptive sliding-mode control for a multiple-user inductive power transfer system without need for communication, IEEE Trans. Ind. Electron., vol. 6, no., pp , Jan 23. [22] L. G. G. de Vicuna, M. Castilla, J. Miret, J. Matas, and J. M. Guerrero, Sliding-mode control for a single-phase ac/ac quantum resonant converter, IEEE Trans. Ind. Electron., vol. 56, no. 9, pp , Sept 29. [23] M. Castilla, L. G. de Vicuna, J. M. Guerrero, J. Matas, and J. Miret, Sliding-mode control of quantum series-parallel resonant converters via input-output linearization, IEEE Trans. Ind. Electron., vol. 52, no. 2, pp , April 25. [24] X. Chen, T. Fukuda, and K. D. Young, Adaptive quasi-sliding-mode tracking control for discrete uncertain input-output systems, IEEE Trans. Ind. Electron., vol. 48, no., pp , Feb 2. [25] Z. Zhang, K. T. Chau, C. Qiu, and C. Liu, encryption for wireless power transfer, IEEE Trans. Power Electron., vol. 3, no. 9, pp , Sept 25. [26] D. Budgett, A. Hu, H. LEUNG, and J. MCCORMICK, Inductive power transfer control using energy injection, Apr. 24, wo Patent App. PCT/NZ23/,84. [Online]. Available: [27] A. Körner and F. Turki, Compact, safe and efficient wireless and inductive charging for plug-in hybrids and electric vehicles, in Advanced Microsystems for Automotive Applications 24. Springer, 24, pp [28] F. Turki, A. Körner, J. Tlatlik, and A. Brown, Compact, safe and efficient wireless and inductive charging for plug-in hybrids and electric vehicles, SAE International Journal of Alternative Powertrains, vol. 3, no , pp. 39 5, 24. [29] H. L. Li, A. P. Hu, and G. A. Covic, A direct ac-ac converter for inductive power-transfer systems, IEEE Trans. Power Electron., vol. 27, no. 2, pp , Feb 22. [3], Primary current generation for a contactless power transfer system using free oscillation and energy injection control, Journal of Power Electronics, vol., no. 3, pp , 2. [3] M. Moghaddami, A. Anzalchi, and A. I. Sarwat, Single-stage threephase ac-ac matrix converter for inductive power transfer systems, IEEE Trans. Ind. Electron., vol. 63, no., pp , Oct 26. [32] M. Moghaddami, A. Moghadasi, and A. I. Sarwat, A single-stage threephase ac-ac converter for inductive power transfer systems, in 26 IEEE Power and Society General Meeting (PESGM), July 26, pp. 5. [33] J. Schneider, Wireless power transfer for light-duty plug-in/ electric vehicles and alignment methodology, SAE International J2954 Taskforce, 26. Arif I. Sarwat (M 8) received his M.S. degree in electrical and computer engineering from University of Florida, Gainesville. In 2 Dr. Sarwat received his Ph.D. degree in electrical engineering from the University of South Florida. He worked in the industry (SIEMENS) for nine years executing many critical projects. Before joining Florida International University as Assistant Professor, he was Assistant Professor of Electrical Engineering at the University at Buffalo, the State University of New York (SUNY). His significant work in energy storage, microgrid and DSM is demonstrated by Sustainable Electric Delivery Systems in Florida. His research areas are smart grids, high penetration renewable systems, power system reliability, large scale distributed generation integration, large scale data analysis, cyber security, and vehicular technology. Masood Moghaddami (S 5) received the B.S. degree from Amirkabir University of Technology, Tehran, Iran, in 29, the M.S. degree from Iran University of Science and Technology, Tehran, Iran, in 22. From 2 to 24, he worked in the industry as an engineer on design and optimization of large power transformers and high-current furnace transformers. He is currently working toward his Ph.D. degree in the Department of Electrical and Computer Engineering, Florida International University, Miami, Florida. He is also a part of FIU-FPL (Florida Power & Light) solar research group. His research interests include finite element analysis, power electronics, contactless power transfer, and renewable energy systems (c) 26 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.