CARLOS A. GALLO FERNANDO L. TOFOLI

Transcription

1 A HIGH POWER FACTOR RECTIFIER ASSOCIATED WITH A ZCZS PWM FULL-BRIDGE INERTER IN A RECTIFIER/INERTER SYSTEM CARLOS A. GALLO Universidade Federal de Uberlândia, Faculdade de Engenharia Mecânica Campus Santa Mônica - Bloco 3M, Uberlândia-MG Brazil gallo@mecanica.ufu.br FERNANDO L. TOFOLI Centro Federal de Educação Tecnológica de Santa Catarina Av. Nereu Ramos, 345-D Bairro Seminário, Chapecó-SC Brazil, CEP 8983-, Fone/Fax: fernandolessa@cefetsc.edu.br ERNANE A. A. COELHO, LUIZ C. DE FREITAS, ALDEIR J. FARIAS, JOÃO BATISTA IEIRA JR. Universidade Federal de Uberlândia, Faculdade de Engenharia Elétrica, Núcleo de Eletrônica de Potência Campus Santa Mônica - Bloco 3N, Fone/Fax: , Uberlândia-MG Brazil Abstract This work presents the operation and design of a high power factor ac-dc-ac power supply rated at kw operating at high switching frequency. Good power factor is obtained using an ac-dc interleaved boost-flyback converter as a preregulator circuit associated with a nondissipative snubber with reduced commutation losses. Commutation losses are practically reduced to zero and EMI emission can also be minimized. The dc-ac conversion performed by a ZCZS PWM full-bridge inverter is also investigated. eywords Ac SMPS, ZCZS, PWM Full-Bridge Inverter. Resumo Este trabalho apresenta o projeto e operação de uma fonte chaveada CA-CC-CA de kw com alto fator de potência operando em alta freqüência. Um valor satisfatoriamente alto para o fator de potência é obtido usando o conversor CA-CC boostflyback entrelaçado como estágio pré-regulador, o qual emprega um snubber não dissipativo para promover operação com menores perdas por chaveamento. Desta forma, pode-se reduzir tais perdas a quase zero, reduzindo também os níveis de interferência eletromagnética. A operação de um conversor CC-CA ZCZS PWM em ponte completa também é investigada. Palavras-chave fontes chaveadas, ZCZS, inversor PWM em ponte completa. Introduction The power supply unit is a very important circuit for all electronic equipment, because it provides the necessary voltages for correct work of the electronic circuits. These electronic circuits are used, usually, to feed professional or domestic equipments, for instance, computer, telecommunication equipments, aviation and military, games, etc. These equipments have become more sophisticated, as it can be seen in [] and [], since reduced size and weight are mandatory requirements, as well as increased efficiency. Generally, these equipments use ac as a primary power energy supply. Therefore, the ac energy must be converted to dc energy, because the majority of these systems require high quality dc power. Linear power supplies are good for low power applications, but are uneconomical and inefficient when more power is required. The alternative is to use switched mode power supplies (SMPS). This kind of power supply presents multi output DC voltages, constant switching frequency, and reduced size and weight when compared with linear power supplies, but the input stages of the switching mode power supplies are well known to be harmonic generators. Recently, there has been great interest about the reduction of input current harmonics and power factor correction (PFC). Moreover, in many singlephase applications, mainly in power supplies, the power level can reach several kilowatts, and in some situations, the input voltage can be quite high too. For these types of application the conventional boost PFC converter has been more used due to its characteristics of dc-voltage gain, lower inductor volume and weight, and losses on the power devices, which will affect converter cost, efficiency, and power density [3]-[5]. It is perfect for pre-regulator applications, but this converter presents appreciable commutation and conduction losses, bringing reduction in the efficiency. Conventional resonant and quasi-resonant converters [6]-[] provide ZCS (Zero-Current Switching) and/or ZS (Zero-oltage Switching) [] [3], and these converters can operate with high-frequency, but these techniques have load limitation, because there are current and/or voltage

2 peaks over the switches and range of frequency control, making difficult the filter components design. The interleaved power conversion technique is a strategy of interconnection of multiple switching cells for which the operation frequency is identical, but for which the internal switching instants are sequentially phased over equal a switching period fractions. This technique can be used with typical PFC IC s [], but in this case the switching cells operate at 4 khz. However with this frequency the volume of filter capacitors is supposed to be greater than that at khz, because the filters must be designed for twice the switching frequency i.e. khz. A good way to reach high-frequency and high-power operation is to use the non-dissipative snubber presented in []-[4]. An ac-dc-ac switching mode power supply provides the conversion of a given ac voltage level to ac voltage. There are several inverters that can be used in this case, but it is necessary that such converters operate in high-frequency and present reduction of switching losses, regulated multiple output, and isolation. A dc-ac ZCZS PWM full-bridge inverter with a nondissipative snubber that can reach high frequency and high-power operation is used in this SMPS with ac output. This converter is suitable for this power level and it provides: - soft switching for full load range; - conduction losses are almost the same as those observed in the hard PWM converter. The main switches turning-on at zero current can reduce significantly the undesirable effects of the parasitic inductances related to the circuit layout. The commutation losses are practically reduced to zero and the EMI emission can also be minimized. The operation of the inverter is analyzed, and design guidelines for the iliary commutation cell are recommended based on this analysis. Experimental results are presented to demonstrate the feasibility of the proposed kw inverter. The Proposed SMPS The ac-dc and dc-ac non-dissipative converters are shown in Figure and Figure, respectively. Figure 3 corresponds to the proposed SMPS. The converters can operate with reduced commutation losses. High power factor is obtained using the average current mode control technique. To simplify the analysis the converters will be presented apart initially. Figure. Ac-dc full-bridge inverter using a soft commutation cell. Figure. Ac-dc interleaved boost-flyback converter associated with the nondissipative snubber. 3 Principle of Operation of The Ac-Dc Interleaved Boost-Flyback Converter Figure shows the interleaved boost converter associated with a nondissipative snubber used as a pre-regulator. Figure 3. High power factor power supply using an interleaved boost-flyback converter and a full-bridge inverter with soft-switching. By definition, the following expressions are valid: o = () CrL o = () C L r r

3 = o3 ( + ) C L L r r I r (3) α = o (4) L C o = (5) dc = (6) = (7) + dc 3 = (8) where: o, o, o3 resonance frequencies; α normalized load current; I o output current; voltage across the flyback stage; dc, dc initial and final values of the voltage across capacitor C r in the fifth stage, respectively. From Figure, one can obtain the waveforms shown in Figure 5. The operation of the converter is divided in six stages, according to Figure 4. A single cell is considered in the analysis due to the inherent analogy between the cells. First Stage (t, t ): This stage begins when switch S is turned on under null current condition, because inductor L r is linearly discharged through the loop involving C o /R o, D 5, and S. The flyback stage maintains the resonance between L and C r. This stage finishes when the voltage across capacitor C r equals the output voltage o, causing diode D 7 to be forward biased. It can be demonstrated that the current through resonant inductor L r is given by: Cr ilr() t = ( dc ) sin L + t (9) arccos Δ t = () Second Stage (t, t ): The voltage across capacitor C r is clamped to o. Resonant inductor L is linearly discharged through the loop formed by D r, D 7, C o /R o, and the flyback stage. This stage is responsible for the PWM characteristics of the converter, and it finishes when switch S is turned off. The current through inductor L is: ( o ) Cr il () t =. t+ ( dc + ) () L L + Δ t = () Third Stage (t, t 3 ): This stage begins when switch S is turned off with null voltage due to resonant capacitor C r, which is linearly discharged through the loop formed by L b, C o /R o, and D 7. C r o Δ t3 = (3) Io Fourth Stage (t 3, t 4 ): There is a resonance between inductor L r and capacitor C r, as the current through L r equals the load current I o. Capacitor C r is negatively charged while inductor L r is fully discharged. 3π Δ t4 = (4) Fifth Stage (t 4, t 5 ): There is a resonance between capacitor C r and inductors L and L r through the loop formed by D 5, C o /R o, D r, and the flyback stage, until the current through L r equals the load current I o. The current through inductor L is: il ( t) = Cr3 ( ) dc sin3t (5) The voltage across capacitor C r is: vcr( t) = - ( -o ) - dc cos t 3 (6) π Δ t5 = (7) 3 Sixth Stage (t 5, t 6 ): The voltage across resonant capacitor C r is clamped to the voltage assumed at the end of the previous resonant stage, until a new switching cycle begins. Boost inductor L b provides energy to the load. Δ t6 = Ts ( Δ t+δ t +Δ t3 +Δ t4 +Δ t5) (8) where T s is the switching period.

4 (a) First stage (t, t ) (b) Second stage (t, t ) (c) Third stage (t, t 3 ) (d) Fourth stage (t 3, t 4 ) (e) Fifth stage (t 4, t 5 ) (f) Sixth stage (t 5, t 6 ) Figure 5. Theoretical waveforms. 4 The Dc-Ac PWM Full-Bridge Inverter The purpose of zero-current-transition (ZCT) is to force the current flowing through a device to decrease to zero before this device is turned on or off. This is accomplished by the addition of an iliary circuit that provides a resonant current to take away the current flowing through the main device before the switching transition. With the ZCT techniques, inverters are expected to achieve a higher switching frequency with reduced switching losses, attenuated acoustic noise, and reduced electromagnetic interference (EMI). Figure shows a typical three-phase ZCT inverter circuit, which was studied in [5] and will not be analyzed in this paper in detail. Historically, Figure 4. Operating stages of the interleaved boost-flyback converter. the research with this circuit configuration goes back to the McMurray inverters for the SCR forced current commutation. For the soft transition research with modern gate-turn-off devices, such as IGBTs, several ZCT control schemes were developed to achieve the zero-current turn-off. In order to provide guidelines for the scheme proposed in [5], which realizes the zero-current and near-zero-voltage switching in high power applications, this paper presents the design considerations of a kw single-phase inverter. 4. Design Example This section presents a design procedure and an example to determine the values regarding the resonant tank elements of the proposed ZCZT commutation for the proposed inverter. The given specifications are presented in Table I. The design procedure consists of four steps, as follows. Table I Design specifications of the single-phase inverter. Parameter alue Output Power P o =W Input oltage E=4 Output oltage o =rms Output frequency f o =6Hz (a) Calculation of the output current peak, given by (9). P I = ( +Δ I) (9)

5 (b) Calculation of the characteristic impedance. To assure the main switches turning-off under ZCS and ZS conditions, during stages 9 and, the peak current taken away from the main switch to the iliary circuit must be larger than the output current peak. From these stages, the current peak through capacitor Cr is given by: I pk E Z = () From (9) and (), a new parameter k can be defined as follows: I pk k = I () where k. If k= is adopted, which is a practical design value to compensate the parasitic losses, the characteristic impedance Z can be given by: Z E ki = () c) Calculation of the resonant frequency. To minimize the reverse recovery of the main diodes, the resonant frequency can be chosen to control the di/dt rate during turn-off, which is then given by (3). di I (3) dt sin k The resonant frequency can be obtained from (3) as follows: di sin dt k = (4) I d) Calculation of the resonant components. If the characteristic impedance and the resonant frequency are given, expressions (5) and (6) can be used to calculate the resonant inductor and capacitor. Z L C = C = (6) Z = L = (5) If a di/dt rate equal to 8A/µs is adopted, the resonant components values L, C, L, and C can be calculated for both legs of the converter, where the first one operates at 3 khz and the second one operates at 6Hz, as shown in Table II. Parameter I Z L C L C Table II Design example. alue 5.4 A 5.3 Ω 5.μH 7.4 nf.6 μh 7. nf 5. Experimental Results A prototype of the proposed switching mode power supply was built using the parameter set in Table III. Table III Design specifications. All switches = IRFP46; C R =4. nf; Diodes = MUR56; C R =5.6 nf; Aux. diodes=apt3d6b; C R = 7.8nF; Full-bridge diodes=hfa5tb6; f S =3 khz; i =/ ; C f =mf; L R =4 μh; C R =7 nf; L b =. mh; P = W; L f =55 μh; o = ; L t =6 μh; I o = 9. A; C =68 μf;. Lf = μh; The experimental results are presented and discussed as follows. As it can be seen the commutation of the switches occurs without losses and power factor is almost unity. Figure 6 shows the power factor correction at rated load, as displacement power factor is.998 and current THD is 5.%. Figure 6. Input voltage and input current at rated load in /div.; I in 5 A/div.; time ms/div. Figure 7 shows soft commutation in the boostflyback converter switch M, which is turned on and off under zero current and zero voltage conditions, respectively. Additionally, there are not high current and voltage peaks when the switch is turned on or off, respectively. Figure 7. Drain-to-source voltage and current waveforms of switch M. Scales: S /div.; I S 4 A/div.; time.8 μs/div. Figure 8 shows the commutation in the inverter switch S, which is turned on and off under zero voltage and zero current conditions, respectively. Figure 9 represents the output voltage and output current waveforms under rated load condition. It can be seen that the control strategy provides nearly sinusoidal output voltage, as the harmonic distortion rate is 4.39%, respectively.

6 Figure shows efficiency as a function of the output power for the rectifier/inverter system, which is above 96% at rated load. quantities, was analyzed theoretically, designed, and implemented. 7 References Efficiency (%) Figure 8. Switch S waveforms. Scales: I s ( /div.); I s (5 A/div.); time ( μs/div.) Figure 9 Output voltage and output current. Scales: o ( /div.); I oi ( A/div.); time (5 ms/div.) A Power (W) Figure. Efficiency as a function of the output power. 6. Conclusions This paper has reported analytical, simulation and experimental developments on a SMPS using the PFC ac-dc boost-flyback converter associated with a nondissipative snubber. It has been demonstrated that the current waveshaping control technique in combination with the nondissipative snubber provides highly efficient power factor correction without commutation losses. The proposed approach allows good performance in high frequency of operation. The second part of this work has presented an active soft switching topology for a PWM ZCZS full-bridge inverter based on the resonance principle. The iliary circuit is bi-directional, operating at ZS and ZCS conditions. The inverter operation and performance are evaluated by experimental tests, which validate the soft switching commutation in ZS and ZCS ways for the main switches. The objective initially proposed was reached, as a switched mode power supply with power factor correction (.998), high efficiency (above 96%) and low harmonic distortions of the input and output [] Staffiere, D., Mankikar, M., Power Technology Roadmap, APEC, PP [] Mankikar, M., Analysis of arious Power Supply business models, APEC, PP [3] Zhang, M.T., Jiang, Y., Lee, Fred, C., Joavanovic, Milam M.; Single-Phase Three-Level Boost Power Factor Correction Converter, IEEE APEC 95, PP [4] Miwa, B. A., Otten, D. M., And Schlecht, M. F., High Efficiency Power Factor Corretion Using Interleaving Techniques, Proceedings of APEC`9 IEEE Catalog n : 9CH389-, pp [5] Corrêa Pinto, J. A.; Pereira, A.A.; de Freitas, L.C.; ieira Jr., J. B.; Farias,.J; "A Power Factor Correction Preregulator AC-DC Interleaved Boost With Soft- Commutation", PESC'97, USA [6] Lee, F. C.; High-Frequency Quasi-Resonant Converter Technologies, Proceedings on the IEEE, ol. 76, no 4, April 988. [7] Barbi, I., Bolacell, J. C. and ieira Jr., J. B., A Forward Pulse-Width Modulated Quasi-Resonant Converter: Analysis, Design and Experimental Results, IEEE IECON 89, Record, pp. -6, Philadelphia, Pennsylvania, USA. [8] Barbi, I., Hey, H. L. and ieira Jr., J. B., A Half- Bridge Pulse-Width Modulated Zero-Current Switched Quasi-Resonant Converter, IEEE IECON 89, Record, pp. 4-47, Philadelphia, Pennsylvania, USA. [9] ieira Jr., J. B., Quasi-Resonant Converters: New topologies, project and analysis, PhD Thesis, UFSC, Florianópolis, SC, Brazil, 99. [] Lee, F. C., Hua, G. and Leu, C. S., Novel Zero-oltage-Transition PWM Converters, IEEE PESC 9, Record, pp. 55-6, Toledo, Spain. [] Barbi, I., Cruz, C. M. T., Unit Power Factor Active Clamping Single Phase Three Level Rectifier, Proceedings of APEC,, pp [] Freitas, L. C., Farias,. J., Caparelli, P. S., ieira Jr., J. B., Hey, H. L., Cruz, D. F. An Optimum ZS-PWM DC - to - DC Converter Family: Analysis, Simulation and Experimental Results. IEEE PESC 9, Record, pp. 9-35, Toledo, Spain, Jul. 99. [3] Lee, F. C., Zhang, J., Sheo, J., Xu, M., Jovanovic, M. M., Evaluation of Input Current in the Critical Mode Boost PFC Converter for Distributed Power Systems, Proceedings of APEC,, pp [4] Lee, F. C., Mao, H., Zhou, X., and Boroyevich, D. Improved Zero Current Transition Converters for High Power Application, in Proc. IEEE IAS Annu. Meeting, 996, pp [5] Gallo, C. A., Tofoli, F. L., Freitas, L. C., Farias,. J., Coelho, E. A. A., ieira Jr., J. B. Proposal of A Soft Commutation Cell Applied To The Single- Phase Full-Bridge Inverter, Congresso Brasileiro de Eletrônica de Potência, 7, pp