A Damping Scheme for switching Ringing of Full SiC MOSFET by Air Core PCB circuit

Transcription

1 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R A Damping Scheme for switching Ringing of Full SiC MOSFET by Air Core PCB circuit Jaesuk Kim, Student Member, IEEE, Dongho Shin, Student Member, IEEE, Seung-Ki Sul, Fellow, IEEE Abstract In this paper a resonant damping circuit to suppress the switching ringing of full SiC MOSFET is proposed. It reveals that the ringing phenomenon caused by parasitic impedances of switching circuit can be damped out using air core PCB transformer which has a properly designed secondary side circuit. The design method for PCB transformer and the secondary circuit is developed considering the physical dimension applied to the PCB transformer inserted between full SiC MOSFET module and snubber capacitor. Experimental results using for V-A full SiC MOSFET module validate the design method and the performance of the proposed air core PCB circuit. Thanks to the damping circuit, the resonant component due to the switching ringing has been reduced to a half compared to that of tem without damping circuit. Conductive, Radiated emission (EMI) Regulation f f L L Possible minimum loss Switching Loss Index Terms SiC MOSFET, switching, ringing, EMI, double pulse test, damping circuit A I. INTRODUCTION S development of the wide-bandgap power semiconductor technology, SiC MOSFET and SiC Schottky diode have been mass produced and commercially available. Thanks to much faster switching speed and virtually no reverse recovery loss compared to conventional Si based power semiconductor, the switching loss of these SiC devices are quite small and the conduction loss of these SiC MOSFET is also small due to its unipolar structure []-[]. And these inherent features of SiC devices enable power converter much more efficient compared to the power converter with Si devices. Because of the higher frequency switching of the converter exploiting reduced switching loss of SiC device, the size of the reactive components used for filter of the converter can be significantly shrunken, thereby the power density of converter can be greatly enhanced. Therefore, despite of their relatively higher cost than Si IGBTs, the SiC MOSFET and Schottky diode have already been applied to some industrial products where power density and efficiency are premium []. However, SiC devices have some drawbacks. Due to the high speed switching characteristics of the SiC MOSFETs, there is a problem of increasing the risk of dielectric breakdown in motor drive application []. It also has been reported that many EMI problems arise when SiC devices are applied [], []-[]. The source of these EMI problems is analyzed as a steep voltage or current slope that occurs during the switching. And the J.-S. Kim, D.-H. Shin and S.-K. Sul are with the Department of Electrical and Computer Engineering, Seoul National University, Seoul -7, Korea ( kimjs77@gmail.com, only@snu.ac.kr, sulsk@plaza.snu.ac.kr). Fig.. Trade off relationship between EMI and switching loss switching ringing occurs due to the parasitic inductance and capacitance of the switching circuit. The voltage or current waveforms of SiC devices with the faster slope and ringing have several tens of MHz frequency components and these components are conducted to adjacent conduction path, coupled to other circuit via coupling impedance of the tem, and radiated. Especially, since the recent SiC MOSFETs have used the existing IGBT package to share the IGBT market and to reduce package cost, it impairs gate stability and worsens EMI issues because of its relatively large leakage component of existing module structure [7]. Accordingly, these EMI due to inherent faster switching of SiC devices has been issued from the dawn of the application of the devices. A simple way to address this EMI issues is to control the switching speed of the SiC devices. By increasing the turn-on, off time of SiC devices, it is possible to reduce the degree of EMI by limiting high frequency components. However, this approach is not appropriate because slowing down the switching speed inevitably increases the switching loss. Thus, it can be noted that the loss and the degree of EMI have a trade-off relationship. And, a proper switching speed to meet the EMI standard for given application has been carefully decided []. Fig. is a conceptual graph to help understanding preciously mentioned trade-off relationship. The generic SiC devices could give curve f which is relationship between the switching loss and EMI, When EMI regulation is indicated by red solid line, the switching loss that has to be selected at best is the L point in the f curve. If a lower level of switching loss is required keeping the same EMI regulation, this curve should be moved to the bottom left corner by EMI reduction techniques like f curve. From this point of view, there is a strong demand for an EMI suppression technique reducing EMI -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

2 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R Fig.. A screw type standard package of a A, V class Full SiC MOSFET module Fig.. (a) Double pulse test circuit considering parasitic components and (b) Simplified circuit of (a) but keeping similar level of the switching loss of the power converter. Many research results have been reported for this purpose []-[]. These researches can be categorized largely as three approaches. One is reducing the source of EMI, another is designing the EMI filter, and the last is improving the parasitic impedance of the switching circuit to reduce EMI. The approach reducing the source of EMI is actively adjusting switching speed (dv/dt) of the devices. Ref. []-[] show that a properly controlled gate driver improves the relationship between EMI and the switching loss. The approach designing the common or differential mode EMI filter (or dv/dt filter) has been introduced in []. Finally, there is an approach improving the parasitic impedance of the switching circuit. This method is used for the purpose of reducing the overshoot and ringing phenomenon appearing at the switching of the device. The ringing phenomenon is caused by resonance of parasitic components, and the magnitude of the current or voltage at such a resonant frequency becomes a key factor in satisfying the conductive or radiative EMI regulation []-[]. In this approach, there are several methods to improve the parasitic impedance of the switching circuit. A typical one is the method optimizing the DC bus bar design [7] and the packaging of the SiC device itself [7], []-[9] by minimizing the parasitic inductance which would move the resonant frequency to higher frequency range. Also, there is another method to suppress resonance by increasing the damping components of the resonance using ferrite bead (RL snubber) or RC snubber [], []. In this paper a noble resonant damping circuit is proposed to reduce the ringing phenomenon at the switching and it finally diminishes the source of EMI. The conventional approach using ferrite bead for the resonant damping is disadvantageous in that the damping performance is deteriorated when the ferrite bead is magnetically saturated by large load current []. And, it is practically difficult to insert the ferrite bead between snubber capacitor and full SiC MOSFET module considering that a package of a few hundred amperes of ~V voltage blocking class is a screw type as shown in Fig.. RC snubber circuit is also difficult to be installed. Even if the circuit is installed, the resistor in series with the snubber capacitor would not be effective due to the parasitic inductance in the snubber circuit itself. The damping component in this paper is made of thin PCB (Printed Circuit Board) and can be easily inserted as a snubber circuit between the power module and busbar/busplate. Also, using the air core, it is free from magnetic saturation and conspicuously reduces the switching ringing by damping out a component at a specific frequency band in voltage/current waveform without increasing the parasitic inductance. II. ANALYSIS OF RESONANCE IN SWITCHING CIRCUIT AND PROPOSED RESONANT DAMPING METHOD A. Modelling of the switching circuit considering parasitic components The ringing phenomenon at the switching is a resonance between the junction capacitance of the power device and parasitic inductance [9], []. This can be depicted as Fig. (a). It is a double pulse test circuit for analyzing switching characteristics of the devices. The capacitor shown in the figure represents the junction capacitance of the SiC MOSFET and Schottky diode. The parasitic inductance includes the inductance in the package, DC bus inductance between snubber capacitor and SiC module, and the ESL (Equivalent Series Inductance) of the snubber capacitor itself. In the standard package shown in Fig,, the parasitic inductance is generally several tens of nh, and the parasitic capacitance varies from several to several hundred nf depending on the drain source voltage. For example, in the case of the SiC module "CASMBM" used in the experiment, L stray = nh and C oss =. nf. Consequently, the resonance frequency due to such parasitic components lies in the range of several to several tens MHz. Fig. (b) is conceptually simplified circuit assuming that the voltage slope which is controlled by the gate driver is constant. It is assumed that the voltage slope across the switch varies in a trapezoidal shape as shown in Fig. and that the slope of trapezoidal varies depending on the gate resistance of the gate driver. The trapezoidal voltage source in Fig. (b) has frequency components shown in Fig., and the resonance appears as the voltage and current ringing in the switching waveform. As the impedance at the resonant frequency is pure resistance, most of the components contributing to the damping are the ESR(Equivalent Series Resistor) of the snubber capacitor, the drain-source turn-on resistor of the SiC MOSFET, and high frequency resistance of the DC busbar. Because the high power SiC module in several hundred amperes, such as the device in Fig., has very low drain-source on-resistance and the commonly used snubber is a film capacitor with a very low ESR, it can be predicted that the damping component of high -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

3 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R -db/dec times times times -db/dec Conductive EMI Radiative EMI IGBT(Tr,Tf : ns, khz) V Tr Tf SiC(Tr,Tf : ns, khz) SiC(Tr,Tf : ns, khz) t i + - di EMF M dt di EMF M dt Fig.. The structure of Rogowski coil (a) without return line, (b) with return line i + - Hz Fig.. Equivalent circuit of double pulse test circuit M Passive component circuit N PCB Passive component circuit + - L C R Fig.. Equivalent circuit of double pulse test circuit with resonant damping circuit power converter becomes very small and it is difficult to properly damp out the resonant current. Therefore, generally accepted method is to minimize the parasitic inductance and raise the resonant frequency to a higher frequency band. As shown in Fig., the frequency component of the voltage waveform decreases at the rate of -db/dec. as frequency increases, and the resonance component can be reduced accordingly as the resonant frequency increases. It can be seen that the switching speed and switching frequency of the devices decide the frequency spectrum of the voltage at the switching. With slower switching and less switching frequency in the case of Si based IGBT the voltage spectrum is at least db lower compared to the spectrum of faster switching and higher switching frequency in the case of SiC device. However, minimizing the parasitic inductance is not easy to accomplish unless the snubber capacitor and the device package are restructured because these two parts account for the majority of the total parasitic inductance of the switching circuit. B. Principle of the proposed resonant damping circuit The principle of resonant damping proposed in this paper is to adjust the impedance of the switching circuit by using air core transformer (or mutual inductor). In Fig., the concept of the principle is depicted as a circuit schematic. In Fig., it is shown that the switching circuit can be represented as an equivalent LCR circuit coupled to the proposed damping circuit. The resonant circuit on the primary side and the additional damping circuit on the secondary side are coupled with an air-core transformer. Since the air-core transformer does not have ferromagnetic material, there is no Passive component circuit Fig. 7. Assembly of proposed PCB transformer and power module magnetic saturation caused by the load current and only AC component can be transmitted to the secondary side. This proposed circuit can be easily fabricated as a form of PCB transformer and it is similar to the structure of the PCB Rogowski coil which measures the current on the primary side using magnetic field around the conductor []-[]. In a similar principle, the PCB transformer can have effect of adding the impedance of the primary side by inserting passive element into the Rogowski coil structure such as shown in Fig.. The inserted impedance can be freely implemented by using various SMD (Surface Mounted Devices) resistor and capacitor components on the PCB transformer. Because this circuit can be implemented thin and small, it can be easily inserted into a power module as shown in Fig. 7, where the PCB is inserted between power module and snubber capacitor. III. DESIGN OF RESONANT DAMPING CIRCUIT The aforementioned PCB transformer can be modeled as a single phase coupled inductor. And, it is shown briefly in Fig.. The voltage-current relationship of Fig. can be expressed as (). di di v L M dt dt () di di v M L dt dt The above equation can be rewritten by using the impedance of the passive circuit Z(s) in the Laplace domain as shown in (). M s V LsI I L s Z() s () () PCB L si Z s I -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

4 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R M i i Passive component circuit Z(s) L v v L a b Fig.. Equivalent circuit of Assembly of proposed PCB transformer and power module As shown in (), the impedance of inserted PCB circuit seen from the primary side is Z PCB(s). Based on this equation, the PCB circuit impedance Z(s) can be designed to increase the damping component of the equivalent LCR circuit of the primary side. Firstly, as a case study, the physical range of mutual inductance M and self-inductance L values can be calculated. Based on this, the optimal design of the PCB circuit for resonance damping can be achieved. A. Mutual and Self Inductance of PCB Transformer A design of the mutual inductor is the most essential part of PCB transformer. To do that, mutual and self- inductance of the secondary side, denoted as M and L respectively, can be represented based on the actual dimension of the coil. The targeted SiC MOSFET module is CREE's CASMBM with IGBT type standard package. Because two SiC switches are placed in a module, two damping circuits should be used per module as shown in Fig. 7. And two circuits can be placed at the same PCB. Fig. 9 is a full-size circular PCB coil based on the dimensions of a given package. As mentioned before, the structure of this coil is very similar to that of the Rogowski coil. In case of the circular coil as above, mutual inductance between the primary and secondary sides and self-inductance can be expressed as (). Nh b M ln ( H) a () N h b L ln ( H) a In (), a stands for inner radius of coil, b for outer radius of coil, and h for height of each turn of coil. Dimensions a, b, and h are terms related to the cross-sectional area of the coil, and N is the number of turn of the circular coil. Each dimension is specified in Fig. 9. Substituting the dimension of the circular PCB coil shown in Fig. 9 into (), M and L can be expressed as () in terms of number of turn of the coil. M k N ( H) () L k N ( H) where k is. -9 and it is decided by the cross sectional area of each coil turn. Although the core shown in Fig. 9 is designed as a symmetrical circular shape for the convenience of calculation Fig. 9. A full-size circular PCB coil based on the dimensions of a given package; a=.mm, b=.7mm, h=.mm, N=9 of M and L, the shape of the coil can be modified to fit the actual structure like Fig. 7 or to maximize cross-sectional area (or k ) for getting larger mutual inductance. In addition, it is possible to design a coil in which one side is open like the structure of an actual PCB Rogowski coil sensor shown in Fig. (b) and it will increase the usability of the assembly of the PCB. In this paper, a perfect circular shaped coil has been used. In the following session, a passive elements to be inserted into the PCB coil based on the above equation is derived. B. Design of PCB Circuit impedance As shown in Fig., when the impedance of the primary side is equivalent to the LCR resonance circuit, the total impedance with the PCB transformer can be expressed as (). M s Ztotal () s Ls R C s L s Z() s () Z ( s) Z ( s) PCB In (), L, C, and R denote lumped sum of inductive, capacitive, and resistive parasitic components associated to each switches as depicted in Fig.. Based on (), there are two design methods of impedance of the PCB circuit for suppressing the resonance of the primary side. One is moving the resonant frequency to a higher frequency by inserting an impedance in the PCB circuit, and the other is increasing the resistive component of the primary side while the resonance frequency remains unchanged. Firstly, there is a method with reducing the inductive or capacitive component of the primary side to increase the resonance frequency of the switching circuit. For this purpose, Z PCB(s) can be simulated with a negative capacitor or inductor to reduce the total capacitance or inductance of the primary side. Since Z PCB (s) has s in the numerator, it is hard to simulate it as the capacitor, but easy to make negative inductance component when Z(s) =. If Z(s) =, Z PCB(s) can be expressed as (), which means that the negative inductance component can be added to the primary side. () M s ZPCB s k s () L -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

5 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R This negative inductor is expressed as a function of k irrespective of the number of turns N of the PCB transformer. Therefore, the cross-sectional area of the PCB coil should be increased to have a noticeable effect of () to the primary circuit. However, considering that the k value calculated in Section.A is very small, it is almost impossible to get visible effect under the limited dimensions of the coil. As a next option, there is a method for increasing the damping at the resonance frequency. When a resistor is inserted into the PCB coil (Z(s)=R ), the impedance appeared on the primary side using () can be calculated as (7). s Z () PCB s k s RL, where ω RL is R /L. The resistance R for maximum resistive component at the resonance frequency (ω ) can be derived as () from (7). R L () In this case, the available maximum resistance to be inserted into the primary side can be expressed by (9). k Re ZPCB ( j ) (9). According to (9), the coil can additionally insert a resistance, up to mω. Because the high frequency resistance component of switching circuit is about to mω, it is hard to expect noticeable difference since the inserted resistance only increases the total resistance by about to %. Another method is using an LC resonance circuit. The impedance of the LCR resonant circuit on the primary side (Z (s)) has an inverted triangular shape impedance on the Bode plot, which is minimized at the resonant frequency. If the Z PCB(s) has a triangular impedance characteristic with the maximum impedance at the resonance frequency, the total impedance at the resonance frequency could be significantly increased. When a capacitors and resistors are inserted in series, /C (s) + R, in a PCB circuit, the above-mentioned triangular shape Z PCB(s) can be implemented. If /C (s) + R is substituted to Z(s), () can be obtained with M k L. s ZPCB () s k s s LC LC LC (7) (), where LC LC, LC R C L. In the same way, the impedance of the primary side can be derived as (). s s Z () s L () s, where L C, R C L. The total impedance on the primary side can be given as (). Z ( s) Z ( s) Z ( s) total PCB L s s s k s s LCLCs LC () It can be easily seen that the resonance frequency of the PCB coil should be matched to the resonance frequency of the tem because it is desired to insert the maximum impedance at the resonance frequency. In this case, the only design parameter of the PCB coil is the damping coefficient or, equivalently, resistive component R. LC The total impedance Z total(s) is shown in Fig. when different resistances are used. In Fig., as the resistance decreases, the impedance at the resonant frequency increases but two new resonant frequencies arise with a phase of at both sides of the original resonant frequency of the tem. Therefore, it can be seen that the damping coefficient LC should be properly selected. It is not easy to obtain the analytical solution of () to maximize impedance without incurring resonance. In this paper, a numerical solution was used to obtain the optimal resistance. From (), () can be derived under the condition () that Z total(s) has one resonance frequency where it has zero phase in the limited range of the resistance, R. : Im Z ( j), for uniqe () LC total k LC () L In practice, optimal is at near the lower limit in () LC because a solution is at the boundary where two or more resonances likely occur. Assuming that the solution is roughly set as (), the primary impedance obtained by of the PCB transformer is given by (). Z PCB k LC () L k k ( j ) LC k L () As the result, the added resistive component at primary side is up to times larger in case of Z(s)=R. Using k in (), the LC Ztotal ( s) : R Ztotal ( s) : R Z ( s) : R total Z () s Z ( ) : PCB s R Z ( ) : PCB s R Z ( ) : PCB s R Fig.. Bode plot of ZPCB(s), Z(s), Ztotal(s) when Z(s)=/C(s)+R -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

6 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R Voltage probe Passive component circuit Ids ( A / div) Shunt resistor For current measurement Full SiC MOSFET Module Snubber capacitor + - V ds I ds Fig.. A double pulse test set-up and measurement devices Passive component circuit ( a) Vds ( V / div) ns/ div Vds ( V / div) Copper Spacer for PCB circuit Ids ( A / div) ns/ div Fig.. 9mm copper spacers for PCB transformer value of () can be set as approximately.7 and the impedance given by () would be times larger than the value in (9). Therefore, it can be concluded that by adding RC circuit to the terminal of the secondary side of the PCB transformer, much higher resistance at the primary side can be implemented to damp out the resonance. A. Experimental set-up IV. EXPERIMENT To verify the effectiveness of the proposed PCB circuit, an experimental test set-up for a double pulse test, which is generally used to evaluate the switching characteristics of the switching devices, was constructed. Since the resonance frequency is tens of MHz, the current was measured by using a coaxial shunt resistor and the voltage was measured using a high-voltage passive probe. The delays of all measuring devices were de-skewed before the test. CREE s driver product cgdhbp was used for SiC MOSFET gate driving. Fig. is a photograph of experimental set-up for the double pulse test. A uh air core inductor was used as a load. In the figure, voltage and current measurement points for switching loss calculation have been marked. In principle, the PCB embedding transformer should be inserted into the space between the + or - terminal of the module and the lug type snubber capacitor as shown in Fig. 7. However, it was difficult to apply this principle in Fig. 7 directly to the double pulse test set-up because the test set-up itself was implemented on PCB board as shown in Fig.. Thus, 9mm copper spacers were used like Fig. to secure the space to insert the PCB transformer. Because of the copper spacers, the inductance increases by nh and the resonant frequency decreases by about %. Fig.. A switching waveforms when the gate resistance is Ω at A load current; (a) Turn on (b) Turn off B. Characteristics of resonant component of full SiC MOSFET module The switching characteristics of SiC MOSFETs were measured with the copper spacers. The gate resistor was changed from,.7,, to Ω, and the turn on and off waveforms were recorded when the load current was changed to,, and A. Fig. is the switching waveform when the gate resistance is Ω and the load current is A. It can be seen that the resonance frequency is about MHz and the time constant of the attenuation of the resonance is relatively large. The waveforms in Fig. was stored and FFT (Fast Fourier Transform) was performed. The results are shown in Fig. (a), (b) and it can be seen that the resonance component at MHz exists in large quantity. In an EMC test, it cannot pass the test even if one frequency component exceeds the reference value. Therefore, it is important that the maximum magnitude of the voltage and current at a certain frequency should be minimized. In Fig. the magnitude of maximum frequency component within the range of to MHz is shown according to different gate resistances and load currents along with the switching loss. Also, in order to represent the total value of the resonance frequency component itself, all frequency components from to MHz are root sum squared to get RMS value. In Fig., this value of the resonant frequency components and the switching for different gate resistances and load currents are shown. From Fig. and, it can be seen that smaller gate resistance to decrease the switching loss results in larger magnitude of voltage and current at and near the resonant frequency. This is similar to the situation in Fig. mentioned in the introduction part of this paper. -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

7 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R 7 Vfft max(v) Ω A.7Ω A Ω A Ifft max(a) C. Effect of resonant damping PCB circuit.... The L of the switching circuit is estimated by using the resonant frequency and the attenuation ratio shown in Fig. 7 [7]. The self-inductance of the PCB circuit was measured by an impedance analyser (Fig. ), and the result was almost consistent with (). In order to insert the maximum resistive component in the switching circuit, a circuit consisted with series connected resistor and capacitor is inserted in the PCB circuit as described in Section.B, and the value of capacitor C is calculated by (7) using the measured resonance frequency. C (7) L Since the two coils exist in one PCB board and they are inserted in the + and - terminals of SiC module as shown in Fig. 7, the second term of () is added two times. Therefore, the effect of doubling k is shown, and the value of k is substituted into () which is the range of the damping coefficient to get the optimal resistance value, R. Numerical solution was derived in the range of (), and the optimum resistance R was calculated by about Ω. In order to verify the analysis in Session.B, the tendency of attenuation of the switching ringing by changing R from,,, to Ω has been investigated. Fig. 7 is the waveform at the switching ( Ω gate resistance case) when the resonance frequency of the PCB coil is matched to ω and the R resistance varies. From the figure, it can be seen that there are two major frequency components in the.. Ω A.7Ω A Ω A Fig.. The magnitude of maximum (a) Voltage and (b) Current frequency component within the range of to MHz and switching loss for different gate resistances and load currents. Vtotal(Vrms) Ω A.7Ω A Ω A.. Itotal (Arms) Ω A.7Ω A Ω A Fig.. Root sum square of (a) Voltage and (b) Current frequency component from to MHz and switching loss for different gate resistances and load currents. PCB coil(self inductance) Equivalent circuit R =.Ω L =.99H C =.pf Impedance(Ω) PCB Coil impedance(measurement, Ω) magnitude(maesurement) EQV_ circuit(calculation).... Fig.. The result of measuring the impedance of the PCB coil in Fig. 9 using the impedance analyzer (HP9A) and calculated impedance of PCB coil s equivalent circuit(l =.99μH, R =.ohm and C =.pf). TABLE I SWITCHING LOSSES WITH AND WITHOUT PCB COIL (C, R ) Experimental Configuration Without PCB Coil With PCB Coil (pf, Ω) With PCB Coil (pf, Ω) With PCB Coil (pf, ) With PCB Coil (pf, Ω) waveforms when using a low R resistance ((e)~(h) of Fig. 7) and that the ringing attenuates very quickly when the R resistance is Ω ((i) and (j) of Fig. 7). The frequency domain comparison is shown in Fig.. It reveals that the analysis in Section.B is correct in qualitative sense. Too small R resistance in Z(s) can greatly reduce the main resonance frequency component, but it may incur new resonances on both sides of the original resonance frequency. However, this tendency disappears as the R resistance value in Z(s) increases, and switching ringing is attenuated to an appropriate level. Therefore, the final waveform using Ω in Fig. (i), (j) is appropriate and it reduces the resonance component by about %. This effect was also observed under different operating conditions (A, A), as shown in Fig. 9. In Table, the switching losses with and without PCB Coil (C, R ) is tabulated. Since the position of the drain source voltage measurement includes the PCB coil as shown in Fig., the loss shown in Table includes the loss of the PCB coil itself. The switching loss was calculated based on the SEMIKRON s switching loss calculation method []. From the table, it can be seen that the loss variation due to PCB coil is about %. However, since a loss of less than % (J) easily comes from the measurement error, from Table it can be said that the power loss due to PCBs is negligible. MHz E on(mj) E off(mj) E total(mj) E total(%) (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

8 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R A Turn off(w/o PCB) A Turn on(w/o PCB) Frequency spectrum of Voltage(w/o PCB) Frequency spectrum of Current(w/o PCB) A Turn off(w/ PCB; R= Ω) A Turn on(w/ PCB; R= Ω) Frequency spectrum of Voltage(w/ PCB; R= Ω) Frequency spectrum of Current(w/ PCB; R= Ω) () c ( d) - () c ( d) A Turn off(w/ PCB; R= Ω) A Turn on(w/ PCB; R= Ω) Frequency spectrum of Voltage(w/ PCB; R= Ω) Frequency spectrum of Current(w/ PCB; R= Ω) () e ( f ) A Turn off(w/ PCB; R= ) A Turn on(w/ PCB; R= ) - () e Frequency spectrum of Voltage(w/ PCB; R= ) Frequency spectrum of Current(w/ PCB; R= ) ( f ) ( g ) ( h) A Turn off(w/ PCB; R= Ω) A Turn on(w/ PCB; R= Ω) ( g) Frequency spectrum of Voltage(w/ PCB; R= Ω) Frequency spectrum of Current(w/ PCB; R= Ω) ( h) () i ( j) Fig. 7. Experimental results of double pulse test(turn on, off waveform) using Ω gate resistor; (a),(b) without PCB coil; (c),(d) with PCB coil(z(s): pf, Ω) (e),(f) with PCB coil(z(s): pf, Ω) (g),(h) with PCB coil(z(s): pf, Ω) (i),(j) with PCB coil(z(s): pf, Ω) In Fig. and the magnitude of the resonance frequency components with and without the PCB coil is depicted according to gate resistor (,.7,, Ω) with switching losses when the load current is A. As shown in Fig., the magnitude of the highest frequency component is diminished by about % regardless of gate resistor value when PCB coils are used. Therefore, it can be seen that the case of Ω gate resistor with PCB coil and the case of Ω without PCB coil have similar resonant frequency components. This means that, when using Ω gate resistor with a PCB coil, the switching loss is reduced to a quarter, but the level of EMI generated in the - - resonant frequency band is similar to that of Ω gate resistor without PCB coil case. These results show that the trade-off relationship shown in Fig. is enhanced by inserting PCB coils. In conclusion it can be said that thanks to the PCB circuit the curve, f, moves to f in the figure. V. CONCLUSION () i ( j) Fig.. Experimental results of double pulse test(voltage, current FFT results) using Ω gate resistor; (a),(b) without PCB coil; (c),(d) with PCB coil(pf, Ω) (e),(f) with PCB coil(pf, Ω) (g),(h) with PCB coil(pf, Ω) (i),(j) with PCB coil(pf, Ω) The power converter using SiC power devices can reduce the switching loss significantly compared to the converter using the conventional Si based devices. However, high-speed switching -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

9 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R 9 - A Turn off(w/o PCB) A Turn on(w/o PCB) A Turn off(w/ PCB; R= Ω) A Turn on(w/ PCB; R= Ω) 7 7 Vtotal(Vrms) Before PCB After PCB Ω.7Ω Ω Itotal (Arms) Before PCB After PCB.7Ω 7 9 Fig.. Comparison of experimental results with and without PCB coil (Z(s): pf, Ω) ; Root sum square of (a) Voltage and (b) Current frequency component from to MHz and switching loss for different gate resistances and A load currents. Ω Ω () c ( d) A Turn off(w/o PCB) () e ( f ) A Turn off(w/ PCB; R= Ω) A Turn on(w/ PCB; R= Ω) ( g ) ( h) Fig. 9. Experimental results of double pulse test(voltage, current waveform) using Ω gate resistor; (a),(b) A without PCB coil; (c),(d) A with PCB coil(pf, Ω); (e),(f) A without PCB coil; (g),(h) A with PCB coil(pf, Ω) A Turn on(w/o PCB) Vfft max(v) Before PCB After PCB Ω.7Ω Ω Ifft max(a) Before PCB After PCB.7Ω 7 9 Fig.. Comparison of experimental results with and without PCB coil (pf, Ω) ; The magnitude of maximum (a) Voltage and (b) Current frequency component within the range of to MHz and switching loss for different gate resistances and A load currents. based on SiC devices can be reduced by % without increasing the EMI of the converter. Specifically, it is shown that the resonance frequency component in the switching waveform has been reduced to a half when the same gate resistance is used in the gate driver circuit of SiC MOSFET. Therefore, it is expected that the proposed circuit would be helpful for reducing EMI of power converter using SiC power devices without impairing the potential of the switching loss of the converter. Ω Ω 7 9 of SiC devices increases the high frequency component on the switching waveform, which causes various EMI problems. In particular, it may lead to a conclusion that lower switching speed at the expense of increased switching losses can meet the EMC regulations in product development. However, this situation is undesirable to exploit full potential of SiC devices. In this paper, a novel PCB damping circuit has been proposed to improve the trade-off between switching loss and EMI generation. The proposed PCB damping circuit has no ferromagnetic core and can be made very thin. The air-core transformer in this paper is similar in structure to the PCB Rogowski coil, so it can be easily inserted into standard SiC power modules of several hundred amperes. The proposed PCB circuit can add resistive component to a specific resonance frequency band and reduce switching ringing phenomenon conspicuously. In addition, by switching loss measurement, it is shown that the switching loss due to the proposed circuit is negligible. Experimental results have shown that thanks to the proposed PCB transformer the switching loss of the power converter ACKNOWLEDGMENT This work was supported by the Seoul National University Electric Power Research Institute. REFERENCES [] N. Oswald, P. Anthony, N. McNeill, and B. H. Stark, An experimental investigation of the trade off between switching losses and EMI generation with hard-switched All-Si, Si-SiC, and All-SiC device combinations, IEEE Trans. Power Electron., vol. 9, no., pp. 9 7,. [] M. Nawaz and K. Ilves, On the comparative assessment of.7 kv, a full SiC-MOSFET and Si-IGBT power modules, in Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp. 7,. [] K. Yamaguchi, Design and evaluation of SiC-based high power density inverter, 7kW/liter, kw/kg, Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp [] P. Nayak, Y. Sukhatme and K. Hatua, "Passive damping of device current and motor terminal voltage in a SiC MOSFET based inverter fed induction motor drive," IEEE International Conference on Power Electronics, Drives and Energy Systems (PEDES), Trivandrum,, pp (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

10 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R [] H. Tanaka, K. Suzuki, W. Kitagawa and T. Takeshita, "Design for conducted noise reduction on AC/DC converter using SiC-MOSFET," 9th International Conference on Electrical Machines and Systems (ICEMS), Chiba,, pp. -. [] O. Alatise, N. -a. Parker-Allotey, D. Hamilton, and P. Mawby, The Impact of Parasitic Inductance on the Performance of Silicon Carbide Schottky Barrier Diodes, IEEE Trans. Power Electron., vol. 7, no., pp.,. [7] D. P. Sadik, K. Kostov, J. Colmenares, F. Giezendanner, P. Ranstad, and H. P. Nee, Analysis of Parasitic Elements of SiC Power Modules with Special Emphasis on Reliability Issues, IEEE J. Emerg. Sel. Top. Power Electron., vol., no., pp. 9 99,. [] S. Yin, K. J. Tseng, C. F. Tong, R. Simanjorang, C. J. Gajanayake, and A. K. Gupta, A novel gate assisted circuit to reduce switching loss and eliminate shoot-through in SiC half bridge configuration, Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp.. [9] H. Huang, X. Yang, Y. Wen, and Z. Long, A switching ringing suppression scheme of SiC MOSFET by Active Gate Drive, IEEE th Int. Power Electron. Motion Control Conf. IPEMC-ECCE Asia,, pp. 9. [] Y. Lobsiger and J. W. Kolar, Closed-Loop di/dt and dv/dt IGBT Gate Driver, IEEE Trans. Power Electron., vol., no., pp. 7, Jun.. [] N. F. Oswald, B. H. Stark, and N. McNeill, IGBT gate voltage profiling as a means of realising an improved trade-off between EMI generation and turn-on switching losses, Power Electron. Mach. Drives (PEMD ), th IET Int. Conf., pp.,. [] K. Yamaguchi, Y. Sasaki, and T. Imakubo, Low loss and low noise gate driver for SiC-MOSFET with gate boost circuit, Ind. Electron. Soc. IECON - th Annu. Conf. IEEE, pp. 9 9,. [] D. Han, C. Morris, W. Lee, and B. Sarlioglu, Determination of CM choke parameters for SiC MOSFET motor drive based on simple measurements and frequency domain modelling, in Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp. 7. [] Q. Liu, Modular Approach for Characterizing and Modelling Conducted EMI Emissions in Power Converters Modular Approach for Characterizing and Modelling Conducted EMI Emissions in Power Converters, Ph.D. dissertation, Dept. Elect. Eng., Virginia Tech Univ., Blacksburg, VA, USA,. [] Jih-Sheng Lai, Xudong Huang, E. Pepa, Shaotang Chen, and T. W. Nehl, Inverter EMI modeling and simulation methodologies, IEEE Trans. Ind. Electron., vol., no., pp. 7 7, Jun.. [] D. Han and B. Sarlioglu, Comprehensive Study of the Performance of SiC MOSFET-Based Automotive DC-DC Converter under the Influence of Parasitic Inductance, IEEE Trans. Ind. Appl., vol., no., pp.,. [7] N. Zhang, S. Wang, and H. Zhao, Develop Parasitic Inductance Model for the Planar Busbar of an IGBT H Bridge in a Power Inverter, IEEE Trans. Power Electron., vol., no., pp. 9 9,. [] Y. Ren, X. Yang, F. Zhang, L. Tan and X. Zeng, "Analysis of a low-inductance packaging layout for Full-SiC power module embedding split damping," in Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp. -7. [9] S. Guo, L. Zhang, Y. Lei, X. Li, W. Yu, and A. Q. Huang, Design and application of a V ultra-fast integrated Silicon Carbide MOSFET module, Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp. 7. [] T. Kim, D. Feng, M. Jang, and V. G. Agelidis, Common Mode Noise Analysis for Cascaded Boost Converter With Silicon Carbide Devices, IEEE Trans. Power Electron., vol., no., pp. 97 9, Mar. 7. [] I. Josifović, J. Popović-Gerber, and J. A. Ferreira, Improving SiC JFET switching behaviour under influence of circuit parasitic, IEEE Trans. Power Electron., vol. 7, no., pp.,. [] K. Hasegawa, S. Takahara, S. Tabata, M. Tsukuda, and I. Omura, A New Output Current Measurement Method With Tiny PCB Sensors Capable of Being Embedded in an IGBT Module, IEEE Trans. Power Electron., vol., no., pp. 77 7, Mar. 7. [] D. Bortis, J. Biela, and J. W. Kolar, Active Gate Control for Current Balancing of Parallel-Connected IGBT Modules in Solid-State Modulators, IEEE Trans. Plasma Sci., vol., no., pp. 7, Oct.. [] T. Tao, Z. Zhao, W. Ma, Q. Pan, and A. Hu, Design of PCB Rogowski Coil and Analysis of Anti-interference Property, IEEE Trans. Electromagn. Compat., vol., no., pp.,. [] J. Wang, Z. Shen, C. Dimarino, R. Burgos, and D. Boroyevich, Gate driver design for.7kv SiC MOSFET module with Rogowski current sensor for shortcircuit protection, Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, no. 97, pp.. [] N. Langmaack, G. Tareilus, and M. Henke, Novel highly integrated current measurement method for drive inverters, Conf. Proc. - IEEE Appl. Power Electron. Conf. Expo.,, pp [7] S. Ogasawara and H. Akagi, Modelling and damping of high-frequency leakage currents in PWM inverter-fed AC motor drive tems, IEEE Trans. Ind. Appl., vol., no., pp., 99. [] Application Note AN. Determining Switching Losses of SEMIKRON IGBT Modules., [Online]. Available: Jaesuk Kim (S') was born in Busan, Korea, in 97. He received B.S double degrees in mechanical engineering and electrical engineering and M.S. degree in electrical engineering in and, respectively, from Seoul National University, Seoul, Korea, where he is currently working toward the Ph.D. degree in electrical engineering. His current research interests include power electronic control of electrical machines and power conversion circuit. Dong-Ho Shin (S'??) was born in Korea in 99. He received the B.S. degree in electrical engineering from Seoul National University, Seoul, Korea, in, where he is currently pursuing M.S degree in electrical engineering and computer science. He was involved in SiC device based motor drive and designing gate driver. His current research interests include circuit breaker, especially, solid state circuit breaker. Seung-Ki Sul (S'7, M'7, SM'9, F') He received the B.S., M.S., and Ph.D. degrees in electrical engineering from Seoul National University, Seoul, Korea, in 9, 9, and 9, respectively. From 9 to 9, he was an Associate Researcher with the Department of Electrical and Computer Engineering, University of Wisconsin, Madison. From 9 to 99, he was a Principal Research Engineer with LG Industrial Systems Company, Korea. Since 99, he has been a member of faculty of School of the Electrical and Computer Engineering, Seoul National University, where he is currently a -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

11 This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI.9/TPEL.7.7, IEEE TPEL-Reg-7--.R Professor. He was promoted as a fellow of IEEE with the contribution to PWM technology. He has published over IEEE journal papers and a total of more than international conference papers in the area of power electronics. He was the program chair of IEEE PESC and general chair of IEEE ECCE-Asia, ICPE,. He holds U.S.A patents, 7 Japanese patents, Korean patents, and granted Ph.Ds under his supervision. For year, he was the president of KIPE. He was recipient of IEEE Transaction st and nd paper awards on Industrial Application, simultaneously. He was also recipient of Outstanding Achievement Award of IEEE Industrial Application Society and the recipient of 7 Newell award of IEEE PELS. His current research interests include power electronic control of electrical machines, electric/hybrid vehicles and ship drives, High Voltage DC transmission based on MMC, and power-converter circuits for renewal energy sources. -99 (c) 7 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

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