BALUNS ARE A key component of double-balanced

Transcription

1 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 47, NO. 9, SEPTEMBER Design and Characterization of Multilayer Spiral Transmission-Line Baluns Yeong J. Yoon, Yicheng Lu, Member, IEEE, Robert C. Frye, Member, IEEE, Maureen Y. Lau, Peter R. Smith, Lou Ahlquist, and Dean P. Kossives Abstract We discuss the design of coupled spiral transmissionline baluns modeled after the Marchand type. The balun structure consists of a pair of coupled spiral conductors vertically offset across intervening polyimide layers. The baluns are fabricated on various substrates (glass and high- and low-resistivity silicon). The characteristics such as return loss, insertion loss, and output signal imbalance are measured. The center frequencies of 3-dB bandwidths (BW s), primarily determined by their conductor lengths, range from 1.2 to 3.5 GHz. The 3-dB BW normalized by the center frequency is 1.48 in all cases. We observe an optimum BW for better performance. Return losses at the center frequencies range from 13 to 18 db. Amplitude imbalance distributes in the range of db, depending on the sizes of devices and substrates. The minimum insertion loss is 0.55 db for the balun on a glass substrate with 100-m-wide conductors. The devices fabricated on glass and high resistivity (> cm) silicon show remarkably similar behaviors despite the large difference in dielectric constant. This technique is applicable to monolithic microwave integrated circuits. Index Terms Balun, MMIC, multilayer, passive devices, transmission line. I. INTRODUCTION BALUNS ARE A key component of double-balanced mixer and push pull amplifier designs in wireless systems. They provide balanced outputs from an unbalanced input. Balanced outputs for the applications require half the input signal amplitude at the two output terminals, which are 180 out of phase with each other. Conventional transformers have been used as baluns. Many such transformers, however, have an upper frequency limit of several hundred megahertz because magnetic flux leakage and capacitance between the windings limit their higher frequency response. Since active baluns [1] generally consume large dc power, passive baluns are preferred for the reduction of power consumption in wireless systems. In passive baluns, there are several types, such as the 180 hybrid type, lumped-element filter type, and Marchand type. 180 hybrids or ring resonators are often impractical for monolithic integration due to their large size. Filter-type baluns have drawbacks, such as poor balance at Manuscript received February 6, 1999; revised May 20, Y. J. Yoon was with the Department of Electrical and Computer Engineering, Rutgers University, Piscataway, NJ USA. He is now with Lucent Technologies, Murray Hill, NJ USA. Y. Lu is with the Department of Electrical and Computer Engineering, Rutgers University, Piscataway, NJ USA. R. C. Frye, M. Y. Lau, P. R. Smith, L. Ahlquist, and D. P. Kossives are with Bell Laboratories, Lucent Technologies, Murray Hill, NJ USA. Publisher Item Identifier S (99) their outputs and complicated layouts [2]. In recent years, compensated Marchand baluns [3] have been revisited because they provide excellent balanced outputs over a wide frequency range. Marchand-type baluns have been intensively studied. They can be made of coupled microstrip lines [4], Lange couplers [5], or spiral coils [6], [7]. At low gigahertz frequencies, Marchand baluns of the microstrip or Lange-coupler types have relatively large geometry, making them difficult to be integrated. The spiral type has the advantages of compact layouts and increased mutual coupling, leading to shorter metal length for a given operating frequency. However, the reported characteristics from spiral-type baluns display large output signal imbalance, and details of measurement results are often not available [6], [7]. Engels et al. [8] have reported simulations predicting good balun characteristics; however, these results have not been confirmed by experiments. Theoretical analyses based on coupled transmission-line theory have also been reported and compared with full-wave simulations [9], [10]. Measurement results with excellent balance of the type have been reported by the authors [11], [12], however, these reports do not include design concept, extensive experimental results, and the substrate effect. In this paper, we report multilayer spiral transmission-line baluns of Marchand type fabricated on various substrates. The balun has two metal layers that are vertically offset by an intervening dielectric layer to form a coupled transmissionline pair. The design concepts for the case are addressed. The design parameters to provide good performance are found by lumped model simulation [13]. The baluns designed with different geometries on various substrates are characterized in terms of return loss, insertion loss, and output balance. II. PRINCIPLE OF OPERATION The original compensated Marchand balun [3] was a complicate structure consisting of coaxial transmission lines. Later, it was simplified by Roberts [14], as shown schematically in Fig. 1. According to Roberts representation, the input impedance at the point, consists of an open-circuit stub of characteristic impedance in series with the parallel combination of a short-circuit stub of characteristic impedance (formed by the shields) and a load resistance at and. Thus, (1) /99$ IEEE

2 1842 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 47, NO. 9, SEPTEMBER 1999 Fig. 1. Equivalent representation of a Marchand-type balun. Roberts nomenclature for coaxial-cable balun [14]. Microstrip transmission-line version. where is the electrical length. At frequencies for which the electrical lengths of the transmission lines approach, the impedance of the short-circuit stub becomes infinite while the open-circuit stub impedance becomes very small. Consequently, the input impedance at point converges to. Note that the operating frequency range of these devices is centered about their resonant frequency. When frequencies are off the center of bandwidth (BW), is still close to since the open-circuit stub impedance is small and the short-circuit stub impedance is large in a certain frequency range. This explains why this type of balun has broad-band characteristics. Larger values of in Fig. 1 make the balun less sensitive to frequency variation. Fig. 1 shows the concept of a coupled coaxial line balun applied to a monolithic planar structure. This is a direct mapping of the coaxial balun, with one stripline corresponding to the inner conductor of the coaxial line and the other corresponding to the shield. The striplines connected to ground serve as parts of the input transmission line and open-circuit stub and as short-circuit stub simultaneously. Unlike Roberts coaxial design, for the microstrip-line version the characteristic impedance of the lines and may not be identical with source impedance. For input matching, the impedance at point should be transformed back to by the quarterwavelength-long transmission line at the input side. III. DESIGN AND STRUCTURE A simple microstrip-type balun, like the one shown in Fig. 1, has large dimensions when designed to operate in the low gigahertz range because the length of each line at those frequencies should be equal to one-quarter wavelength (typically several centimeters). When the transmission lines are coiled into a spiral configuration, several advantages are expected. Turning the lines back on themselves results in an increase in the mutual capacitance and inductance, both between the lines as well as between individual segments within the same line. Consequently, for the same lengths of conductor, the resonant frequencies of the spiral devices are significantly lower than those for simple straight lines. For Fig. 2. Conductor arrangement for the designed balun. Return and insertion losses as a function of the ratio of the conductor width to the spiral pitch W=(W + S). baluns designed to operate in a particular frequency range, the spiral structure is more compact, and its resistance is lower because it requires less overall length of wire. A previous investigation of a Marchand balun in a spiral configuration used a pair of vertically stacked spiral lines [8], where the centerlines of the conductors lie on top of the other. The characteristic impedance of the stacked structure is determined mainly by the ratio of the dielectric thickness to linewidth. This makes it difficult to match the balun input impedance with the characteristic impedance of an external 50- line. Matching in this case requires either a relatively thick dielectric, unsuitable to thin-film technology, or a very narrow width of conductor with large resistive loss. We have employed a structure that is better suited to multilayer thin-film technologies. In this structure, the upper conductor is centered above the gap in the lower conductor, offset vertically by a dielectric layer. Each of the two spirals is identical, but one is rotated 180 with respect to the other, as shown in Fig. 2. This structure allows the center-tocenter spacing of the lines (which primarily determines their mutual inductance) and the overlaid degree (which primarily determines their mutual capacitance) to be independently varied. Since the structure is close to coplanar, characteristic impedance is mainly determined by the lateral dimensions of the conductors and is nearly independent of dielectric thickness. For a fixed inner radius, the variations in the number of turns and the center-to-center distance between the adjacent lines (pitch), as well as in linewidth result in a different amount of overlap between the two spirals while maintaining a fixed overall conductor length. This changes the mutual capacitance between the coils, whereas only slightly changing the self-inductance. Mutual inductance is primarily determined

3 YOON et al.: DESIGN AND CHARACTERIZATION OF MULTILAYER SPIRAL TRANSMISSION-LINE BALUNS 1843 TABLE I DESIGN PARAMETERS FOR SPRIAL TRANSMISSION-LINE BALUNS by center-to-center distance and it is relatively insensitive to this variation of the overlap. This is equivalent to altering the characteristic impedances and in Fig. 1. Increasing the coil length, either by adding number of turns or by enlarging inner radius, is equivalent to increasing the electrical length, which determines the center frequency of the balun. By analogy with the coupled transmission-line theory, the band center occurs at the quarter-wavelength resonant frequency of the individual segments. Simulation of the lumped model [13] enables us to find the variation in return and insertion loss with 50- single-ended source and 100- balanced load when changing the conductor width. Fig. 2 shows an example of this variation with width for a fixed pitch ( m). It can be seen that the best input matching and insertion loss in this example is obtained when the ratio of width to pitch lies between Over this range, the balun characteristics are less sensitive to the exact value of the ratio. Well-matched design of these structures is greatly simplified by the observation that this ratio works well over the range of linewidths ( m) that we studied. We have verified the designs from the lumped model simulations by using full-wave simulation [15]. The geometric parameters for the designs considered in this study are listed in Table I. Note that in all of these devices, the width-to-pitch ratio is roughly 0.4. Fig. 3 shows a micrograph of a balun. One of the spiral pairs is rotated by 180 with respect to the other. The layout ensures that the directions of current flow in the two coils are the same, further enhancing the mutual inductive coupling. The ground for the device is provided by a coplanar ring surrounding the structure. Notice that the layout of the spiral coils in the design is symmetric to the ground connections. This minimizes imbalance between outputs due to asymmetric connections to ground [12]. Fig. 3 illustrates the cross section of one spiral coil pair. We fabricated the baluns on several different substrate materials: glass, high-resistivity silicon ( 4000 cm), and low-resistivity silicon ( 20 cm). The silicon substrates have a 1- m-thick thermally grown silicon dioxide layer for added electrical isolation. The first 3- m-thick aluminum layer serves as the primary spiral coil. A4- m-thick polyimide layer provides insulation between Al layers. Another 3- m-thick layer of Al for the secondary coil is sputtered and etched, followed by coating of the second layer of polyimide. The third metal layer (also used for solder attachment in the technology) is a 1- m-thick multilayer of Fig. 3. Micrograph of a spiral transmission-line balun. Cross-sectional view of a multilayer spiral pair in the balun. Cr/CrCu/Cu/Au, which forms the interconnections between the inner and outer parts of the spirals. IV. RESULTS AND DISCUSSION A. Measurement Methods When measuring -parameters of multiport devices by using a general two-port network analyzer, the remaining ports should be terminated by auxiliary loads. According to Tippet et al. [16], the loads may be arbitrary. In this measurement, the idle loads consisted of two parts: 20-dB attenuator and 50- load. 1 After measurements of all ports by the method [16], the true -parameter matrices for 50- characteristic impedance at all ports can be obtained by the renormalization equation [17] where and are the corrected and measured -parameter matrices, respectively, and is the identity matrix. is the reflection-coefficient matrix for the auxiliary loads. The reflection coefficient of the load can be found using a one-port measurement method. For our measurements, the idle load was directly attached to a Cascade ground signal ground probe and connected to a measurement probe by using a through of a standard impedance substrate. Since the separation between the probes on the through is less than 100 m, the phase angle change of the reflection coefficient due to the reference plane shift is negligibly small, i.e., in the low gigahertz frequency range. Typically, insertion loss of a discrete balun is measured by connecting two devices back-to-back to form a single-ended input and output. For accurate insertion-loss measurement, the two baluns should have similar topologies and only real (not complex) port impedances. It is important to note that the backto-back configuration for the measurement does not account 1 20-dB calibrated attenuator: Omni Spectra Model and 50- load: Hewlett-Packard part #HP909D. (2)

4 1844 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 47, NO. 9, SEPTEMBER 1999 for any loss resulting from phase imbalance. A formula considering both path loss and phase imbalance loss is given as [18] (3) where is absolute phase imbalance, defined below in (5). The first term in (3) accounts for loss by amplitude attenuation, and the second term accounts for phase imbalance loss. When the phase imbalance is small, the second term can be neglected, and we get Both phase and amplitude imbalances between the outputs are important properties for defining balun characteristics, and are evaluated by the equations (4) and (5) (6) Fig. 4. Typical measured S-parameters of a balun (Design C: 3.5 turns, glass substrate), where the BW and center frequency are defined by a conventional 3-dB method. Insertion loss and imbalance characteristics illustrating the OBW. B. Balun BW The typical measured -parameter response of the baluns is plotted in Fig. 4, and Fig. 4 shows the phase and amplitude imbalance from (5) and (6) and insertion loss from (3). Flat output response, good balance between and, and good input matching throughout the BW can be seen in Fig. 4. The frequencies at the edge of the BW are determined by the conventional 3-dB method. The center frequency of the band is averaged with the lower corner frequency and the upper corner frequency, as shown in the figure. Fig. 5 and shows the center frequency as a function of spiral conductor length. [The length is for a single spiral, e.g., Fig. 1]. Fig. 5 shows the case of 30- m-conductor width for both substrates of glass and high-resistivity silicon, and Fig. 5 shows similar curves for 50- m-wide conductor. The center frequency decreases with the length of coil since the electrical length increases as the length of coil increases. It is found that when the 3-dB BW is normalized by the center frequency, the normalized BW for this type of balun structure is commonly 1.48 for all of the different metal widths and substrates that we examined. Thus, actual 3-dB frequencies and can be readily calculated. Interestingly, the center frequency difference between the baluns on glass and high-resistivity silicon is only approximately GHz for both metal width cases, and is even smaller for longer coils. This is a smaller difference than expected, given the difference in the dielectric constants for the glass (3.6) and silicon (11.9). Apparently, the mutual capacitive coupling between the coils mostly occurs in the polyimide layer [refer to the cross sectional view of the baluns, as described in Fig. 3]. Since the mutual capacitance is Fig. 5. Center frequency versus coil length for: 30- and 50-m-wide conductor on glass and high-resistivity silicon substrates. much larger than the individual capacitance to ground or the self-interwinding capacitance of either coil, it primarily determines the BW of the baluns. This results in center frequencies that are relatively insensitive to the different substrates when baluns are designed with this configuration of the conductors. Near the band edges, all important balun characteristics such as return loss, amplitude and phase balance, and insertion loss degrade severely. Thus, it is useful to define

5 YOON et al.: DESIGN AND CHARACTERIZATION OF MULTILAYER SPIRAL TRANSMISSION-LINE BALUNS 1845 TABLE II OBW NORMALIZED BY CENTER FREQUENCY Fig. 7. Insertion loss at center frequency versus single coil length for 30- and 50-m-wide conductor on glass and high-resistivity silicon substrates. Fig. 6. Return loss at center frequency versus single coil length for: 30- and 50-m-wide conductor on glass and high-resistivity silicon substrates. an optimum bandwidth (OBW) for practical applications, as shown in Fig. 4. The upper frequency bound of OBW is set by a phase imbalance less than 4. The lower one,, is determined by insertion loss at the center frequency plus 0.5 db because added loss places greater demands on the noise figures of successive stages. Thus, when baluns operate in the OBW, excellent amplitude and phase balance at the outputs, high return loss, and low insertion loss are guaranteed. The measured OBW frequencies and normalized by are listed in Table II. Note that, like the center frequency, these band limits are relatively insensitive to the substrate material. C. Return Loss Measured return loss at the center frequency was found to be between db, as illustrated in Fig. 6 and. Lumped model simulations, as seen in Fig. 2, also predict the values being around 13 db. The return loss is mainly determined by the effective characteristic impedance of the coupled-line structure. Since mutual capacitance is the dominant value in the total capacitance, the characteristic impedances of the spiral baluns do not vary significantly when the geometry of the conductors is fixed. Thus, return loss is relatively independent of substrate materials for the given structure. Maximum return losses over the OBW are almost similar to the losses at. D. Insertion Loss Insertion losses of spiral baluns at for different metal width on both substrates are plotted in Fig. 7 and. Increased wire length results in more loss, suggesting that metal resistance plays an important role. The insertion losses of baluns on high-resistivity silicon are slightly higher than those on glass. This difference is caused by additional dissipation in the silicon substrate itself. (The glass is generally considered to be a nearly lossless material in this frequency range.) Insertion loss for baluns on glass with the conductors of 100- m width and 6- m thickness (design H in Table I) are measured and found to be 0.55 db. The length for these devices is cm and the center frequency is 1.45 GHz. These devices show the lowest insertion loss in this study. E. Low-Resistivity Substrate Effects The measured characteristics of the baluns (design A, 3.5 turns) fabricated on low-resistivity ( 20 cm) silicon substrates are shown in Fig. 8. For the device built on lowresistivity silicon, the performance severely degrades so that the device no longer behaves as a balun when compared with Fig. 4. Loss in the substrates deteriorates propagation along the spiral transmission lines. Consequently, the magnitude of is much lower than. The measured results indicate that components, such as baluns, which utilize electromagnetic coupling between structures should avoid lossy substrate materials. F. Output Amplitude Balance When the baluns operate in the OBW, the worst-case amplitude imbalances within the OBW for 50- m-conductor-width baluns are plotted in Fig. 9. The imbalances for glass substrates are generally higher than those for the high-resistivity silicon substrates, and the difference becomes more prominent

6 1846 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 47, NO. 9, SEPTEMBER 1999 Fig. 8. The measured S-parameters of a balun (Design A : 3.5 turns) on low-resistivity (20 1 cm). silicon wafer. We define criteria for an OBW in which the balun operates with better performances in terms of return loss, insertion loss, and amplitude and phase balance at outputs. The upper frequency bound of the OBW is almost the same with the center frequency. The OBW is located around the second quarter of the 3-dB BW. The minimum insertion loss of 0.55 db is achieved for baluns fabricated on glass with 100- m-wide metal. Amplitude imbalance within OBW is less than 0.9 db for glass substrate, and less than 0.5 db for highresistivity silicon substrate, respectively. The devices built on 20- cm resistivity silicon substrates do not exhibit the balun characteristics, indicating the critical importance of low-loss substrates for this type of device. ACKNOWLEDGMENT The authors thank F. Hrycenko and T. Gabara for photomask preparation. REFERENCES Fig. 9. Amplitude imbalance within OBW for 50-m-wide conductor on different substrates. when the length gets shorter. From the lumped model simulation [13], we find that capacitance to ground plays an important role for the amplitude balance. Larger capacitance to ground gives rise to better balance. This capacitance is formed mostly through the substrate, explaining the superior performance on silicon substrates and the superior performance for longer lines. This observation suggests that improvements in the balance characteristics could be obtained by modification of the shape of the surrounding ground ring. Furthermore, when the response near in Fig. 4 is carefully examined, it can be seen to have a slight peak, which dominates the maximum imbalance within the OBW. Thus, by careful selection of, the imbalance over the OBW can be further minimized. V. CONCLUSION We have presented design methods for spiral transmissionline baluns consisting of multilayered metal and polyimide, and characterized the baluns fabricated on various substrates. These structures, which use vertically offset spiral coils, differ from the previously reported vertically stacked devices. The simulation results indicate that when the ratio of conductor width to pitch falls between , these baluns have good return-loss characteristics for 50- source and 50- loads, good amplitude and phase balance, and low insertion loss. The center frequencies of the 3-dB BW range from 1.2 to 3.5 GHz. The relative BW normalized by the center frequency is commonly 1.48 in all cases. Since mutual capacitive coupling between the lines occurs mostly in the insulator, i.e., polyimide, the frequency response of the baluns is relatively insensitive to the dielectric constant of the substrates. Return losses at center frequencies are in the range of db. [1] K. W. Kobayashi, A novel HBT active transformer balanced Schottky diode mixer, in IEEE MTT-S Int. Microwave Symp. Dig., 1996, pp [2] H. Chiou, H. Lin, and C. Chang, Lumped-element compensated high/low-pass balun design for MMIC double-balanced mixer, IEEE Microwave Guided Wave Lett., vol. 7, pp , Aug [3] N. Marchand, Transmission line conversion transformers, Electron., vol. 17, p. 142, Dec [4] A. M. Pavio and A. Kikel, A monolithic or hybrid broadband compensated balun, in IEEE MTT-S Int. Microwave Symp. Dig., 1990, pp [5] M. C. Tsai, A new compact wide-band balun, in IEEE Microwave Millimeter-Wave Monolithic Circuits Symp. Dig., 1993, pp [6] T. Chen, K. W. Chang, S. B. Bui, H. Wang, G. Samuel, L. C. T. Lui, T. S. Lin, and W. S. Titus, Broad-band monolithic passive baluns and monolithic double-balanced mixer, IEEE Trans. Microwave Theory Tech., vol. 39, pp , Dec [7] T. Gokdemir, S. B. Economides, A. Khalid, A. A. Rezazadeh, and I. D. Robertson, Design and performance of GaAs MMIC CPW baluns using over-laid and spiral couplers, in IEEE MTT-S Microwave Symp. Dig., 1997, pp [8] M. Engels and R. H. Jansen, A novel compact balun structure for multilayer MMIC s, in Proc. 26th European Microwave Conf., 1996, p [9], Design of integrated compensated baluns, Microwave Opt. Technol. Lett., vol. 14, no. 2, p. 75, [10] R. Schwindt and C. Nguyen, Computer-aided analysis and design of a planar multilayer Marchand balun, IEEE Trans. Microwave Theory Tech., vol. 42, pp , July [11] Y. J. Yoon, Y. Lu, R. C. Frye, and P. R. Smith, A monolithic spiral transmission line balun, in IEEE 7th Topical Meeting Elect. Performance Electron. Packag., 1998, pp [12], A silicon monolithic spiral transmission line balun with symmetrical design, IEEE Electron Device Lett., vol. 20, pp , Apr [13], Modeling of monolithic RF spiral transmission-line balun, submitted for publication. [14] W. K. Roberts, A new wide-band balun, Proc. IRE, vol. 45, pp , Dec [15] J. Zhao, S. Kapur, D. E. Long, and W. W. Dai, Efficient threedimensional extraction based on static and full-wave layered Green s functions, in 35th Design Automation Conf., June 1998, p [16] J. C. Tippet and R. A. Speciale, A rigorous technique for measuring the scattering matrix of a multiport device with a two-port network analyzer, IEEE Trans. Microwave Theory Tech., vol. MTT-30, pp , May [17] H. Dropkin, Comments on a rigorous technique for measuring the scattering matrix of a multiport device with a two-port network analyzer, IEEE Trans. Microwave Theory Tech., vol. MTT-31, pp , Jan [18] B. E. Willcox, Determine the loss of discrete baluns, Microwave RF, p. 103, Jan

7 YOON et al.: DESIGN AND CHARACTERIZATION OF MULTILAYER SPIRAL TRANSMISSION-LINE BALUNS 1847 Yeong J. Yoon received the B.S. and M.S degrees from Yonsei University, Seoul, South Korea, in 1982 and 1984, respectively, and the M.S.E.E. and Ph.D. degrees in electrical and computer engineering from Rutgers University, Piscataway, NJ, in 1993 and 1999, respectively. He has several years of semiconductor industrial experience in Korea. Since 1997, he has been with the Wireless Research Laboratory, Bell Laboratories, Lucent Technologies, Murray Hill, NJ. His research interests include design and modeling of high-frequency devices, high-frequency mixer design, and passive/active device integration on chip. Maureen Y. Lau received the B.S. degree in biochemistry from the State University of New York at Binghamton, in 1982, and the M.S. degree in material science at Stevens Institute of Technology, Hoboken, NJ, in In 1982, she joined Bell Laboratories, Lucent Technologies, Murray Hill, NJ, where she is currently a Member of Technical Staff, involved in the area of high-speed silicon nmos and bipolar CMOS integrated-circuit technology research and development. Since 1988, she has been involved in advanced electronics packaging research. Yicheng Lu (M 90) received the B.S degree in applied physics from Jiao Tong University, Shanghai, China, in 1982, and the Ph.D. degree in electrical engineering from the University of Colorado at Boulder, in In 1988, he joined the faculty of the Department of Electrical and Computer Engineering, Rutgers University, Piscataway, NJ, where he is currently an Associate Professor. His main research includes multiquantum-well light modulators, MOCVD growth of oxide and wide-bandgap nitride films, and new piezoelectric materials for RF applications. These research projects have generated 110 referred papers, and over 120 conference presentations. He holds four U.S. patents. Robert C. Frye (M 90) received the B.S. and Ph.D. degrees in electrical engineering from the Massachusetts of Technology (MIT), Cambridge, in 1973 and 1980, respectively. From 1973 to 1975, he was with the Central Research Laboratories, Texas Instruments Incorporated, where he as involved with charge-coupled devices for analog signal processing. Since 1980, he has been with Bell Laboratories, Lucent Technologies, Murray Hill, NJ, where his research activities have included thin-film semiconductor devices and neural-network implementation and applications. More recently, his work has focused on advanced electronic interconnection technology, multichip modules, and integrated passive components for RF applications. He is currently a Distinguished Member of Technical Staff in the Design Principles Research Department. Dr. Frye is a member of the International Microelectronics and Packaging Society (IMAPS). Peter R. Smith received the B.S.E.E. degree from Northeastern University, Boston, MA, in 1972, and the M.S.E.E. and electrical engineer (professional) degrees from Columbia University, New York, in 1975 and 1977, respectively. He is currently a Member of Technical Staff in the Electronics Packaging Research Department, Lucent Technologies, Bell Laboratories, Murray Hill, NJ. His research interests include the application of microwave techniques to characterization and modeling of devices in packages. Lou Ahlquist has been with Bell Laboratories, Lucent Technologies, Murray Hill, NJ, for 20 years as a Process Engineer in the area of integrated circuit, laser platforms for photonic subsystems, and flip-chip technologies for digital and RF modules. He has developed wide expertise in lithography, thin-film deposition, and etching. He has one patent pending for a novel lift-off process. Dean P. Kossives received the B.S. degree in chemistry from the University of Iowa, Iowa City, in He is currently a Consultant to the Electronic Material Research and Packaging Department, Bell Laboratories, Lucent Technologies, Murray Hill, NJ, where he has developed and patented processes integral to the production of multichip modules. In 1996, he joined the CDI Corporation, Philadelphia, PA, but has continued working with Bell Laboratories providing expertise in the area of photolithography, wet chemistry, solder bumping assembly, and cost control.

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