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1 Zhang Yun an Shi Jilong an Zhou Lei an Li Jing an Sumner Mark an Wang Ping an Xia Changliang (7) Wie input-voltage range boost three-level DC- DC converter with quasi-z source for fuel cell vehicles. IEEE Transactions on Power Electronics 3 (9). pp ISSN Access from the University of Nottingham repository: %Three-Level%DC-DC%Converter%with%Quasi-Z%Source%for%Fuel %Cell%Vehicles.pf Copyright an reuse: The Nottingham eprints service makes this work by researchers of the University of Nottingham available open access uner the following conitions. This article is mae available uner the University of Nottingham En User licence an may be reuse accoring to the conitions of the licence. For more etails see: A note on versions: The version presente here may iffer from the publishe version or from the version of recor. If you wish to cite this item you are avise to consult the publisher s version. Please see the repository url above for etails on accessing the publishe version an note that access may require a subscription. For more information please contact eprints@nottingham.ac.uk

2 change prior to final publication. Manuscript ID:TPEL-Reg R.> Wie Input-Voltage Range Boost Three-Level DC-DC Converter with Quasi-Z Source for Fuel Cell Vehicles Yun Zhang Member IEEE Jilong Shi Lei Zhou Jing Li Member IEEE Mark Sumner Senior Member IEEE Ping Wang an Changliang Xia Senior Member IEEE Abstract: To solve the problem of the mismatche voltage levels between the ynamic lower voltage of the fuel cell stack an the require constant higher voltage (4V) of the DC link bus of the inverter for fuel cell vehicles a Boost three-level DC-DC converter with a ioe rectification quasi-z source (BTL-DRqZ) is presente in this paper base on the conventional flying-capacitor Boost three-level DC-DC converter. The operating principle of a wie range voltage-gain for this topology is iscusse accoring to the effective switching states of the converter an the multi-loop energy communication characteristic of the DRqZ source. The relationship between the quasi-z source net capacitor voltages the moulation inex an the output voltage is euce an then the static an ynamic self-balance principle of the flying-capacitor voltage is presente. Furthermore a Boost three-level DC-DC converter with a synchronous rectification quasi-z source (BTL-SRqZ) is aitionally propose to improve the conversion efficiency. Finally a scale-own. kw BTL-SRqZ prototype has been create an the maximum efficiency is improve up to 95.66% by using synchronous rectification. The experimental results valiate the feasibility of the propose topology an the correctness of its operating principles. It is suitable for the fuel cell vehicles. Keywors: Boost three-level DC-DC converter fuel cell vehicles Quasi-Z source synchronous rectification wie range of voltage-gain. I. INTRDUCTIN Non-renewable energy sources continue to be consume an fossil fuel relate emissions continue to increase pollution [~3]. Manuscript receive June 8 6. Accepte for publication November 6. This work was supporte in part by the National Natural Science Founation of China uner Grants an 574 an in part by the Research Program of Application Founation an Avance Technology of Tianjin China uner Grant 5JCQNJC39. Yun Zhang Jilong Shi Lei Zhou an Ping Wang are with the School of Electrical Engineering an Automation Tianjin University Tianjin 37 China (fax: ; zhangy@tju.eu.cn jilong_shi@63.com Luxuszl@63.com an pingw@tju.eu.cn). Jing Li is with the Department of Electrical an Electronic Engineering University of Nottingham Ningbo China ( jing.li@nottingham.eu.cn). Mark Sumner is with the Department of Electrical an Electronic Engineering University of Nottingham UK ( mark.sumner@nottingham.ac.uk). Changliang Xia is with the School of Electrical Engineering an Automation Tianjin University an also with the Tianjin Key Laboratory of Avance Technology of Electrical Engineering an Energy Tianjin Polytechnic University China ( motor@tju.eu.cn). With regar to transport the evelopment of clean-energy vehicles can have a major impact on improving air quality (especially in cities) as well as reucing other fossil fuel relate problems [4~6]. The fuel cell vehicle is an important type of the clean-energy vehicle an its obvious avantage is that it provies clean propulsion power with zero emission as well as higher energy utilization [7~9]. However the fuel cell usually has a current source characteristic with low output voltage an high output current. In aition it is ifficult to use it to supply an inverter to rive a vehicle ue to its soft output characteristic [~]. Therefore it must be interface to the DC link bus of the inverter through a step-up DC-DC converter with a wie range of voltage-gain. The wie gap in voltage levels between the fuel cell stack an the DC link bus can be matche an stable DC link bus voltage can also be obtaine. Usually the conventional Boost two-level DC-DC converter is employe ue to its simple structure [3 4] but it suffers from isavantages incluing limite voltage-gain an high voltage stress for its power semiconuctors. To alleviate the problem of mismatche voltage levels the rate voltage of the fuel cell stack has to be increase (increasing the ifficulty of assembling the fuel cell stack). At the same time power semiconuctors with higher rate blocking voltage nee to be employe an consequently the conuction losses can be improve. In orer to reuce the high voltage stress of power semiconuctors Boost three-level DC-DC converters have been propose an then the voltage stress can be reuce by half [5~7]. However there remain two essential problems concerning the interface between the fuel cell stack an the DC-link bus namely the same limite voltage-gain with that of the Boost two-level converter an the complicate control require for the flying-capacitor voltage balance of the Boost three-level converter especially the voltage imbalance of the flying-capacitor in the transient state [8] - this latter may cause power semiconuctor failure. It is therefore necessary to solve these problems for fuel cell vehicles which use the Boost three-level DC-DC converter with a flying capacitor. As to the non-isolate step-up DC-DC converters with high voltage-gain the voltage multiplier circuits are aopte to exten the voltage-gain [9]. The switche-inuctor structures for step-up DC-DC converters can also obtain high voltage-gain as well as the switche-capacitor DC-DC converters [ ]. However these step-up DC-DC converters with high voltage-gain are too complex to reuce their cost an size. The quaratic Boost DC-DC converter can also achieve a high

3 change prior to final publication. Manuscript ID:TPEL-Reg R.> voltage-gain []. However the power semiconuctors of the output sie (the high voltage sie) suffer from high voltage stresses (ue to the high output voltage) an create a high v/t uring switching. Although a large conversion ratio interleave Boost DC-DC converter using two stages in parallel an one series multiplier stage can convert 4V to V [3] there still two ioes in the multiplier stage which suffer from the full output voltage stress. A family of ioe-couple-wining Boost DC-DC converters with a high voltage-gain can perform better than their active-clamp counterparts ue to recycle leakage energy [4] achieving a maximum efficiency about 9.7%. Base on [3] an [4] a high voltage-gain interleave Boost DC-DC converter magnetically couple to a voltage-ouble circuit was propose in [5]. In aition another high voltage-gain Boost DC-DC converter can obtain higher efficiency which is base on the three-state commutation cell with aitional two transformers (six winings) [6]. Z source net has been applie in the traitional step-up DC-DC converters to achieve the higher voltage-gain [7] but their input an output sies on't share the common groun which may result in maintenance safety an EMI problems. In aition the output ioe can be replace by an inuctor in the Z source DC-DC converters [8] but the voltage-gain is reuce unexpectely. The ioe rectification quasi-z (DRqZ) source circuit is another moifie energy storage circuit structure which has been propose for the combination of a low voltage DC source an an inverter [9 3]. It can also be use in the step-up DC-DC converters with the features of lower capacitor voltages an the common groun [3] but its voltage-gain is the same as the conventional Z source DC-DC converters an the voltage stress of the power switch is still as high as the output voltage. The couple inuctor base Z source DC-DC converters can achieve high voltage-gain by setting the turn ratio of the couple inuctor [3]. However the spike voltage of the power switches may be very large ue to the leakage inuctor of the couple inuctor. In [33] a common groune Z source DC-DC converter with high voltage-gain is presente by changing the connection way of the grouning the input source an the loa are locate on the same sie of the Z source instea of being locate on both sies of the Z source. It is analyze in [33] that the voltage stress of the power semiconuctors is reuce in the range of half of the output voltage to nearly the output voltage when increasing the uty cycle (voltage-gain). In aition the current stress of the power switch is several times as high as the output current while increasing the uty cycle (voltage-gain). In this paper a wie input-voltage range Boost three-level DC-DC converter with a ioe rectification quasi-z source (BTL-DRqZ) is propose as a solution which can reuce the voltage stress of all semiconuctors to half of the output voltage; it also has a common groun for the input an output by using the flying-capacitor three-level structure an operates well with a high voltage-gain proper uty cycles (.5<=<.75) an balancing of the voltage of the flying capacitor without aitional harware. Although one more power switch an ioe are employe compare to the conventional quasi-z source Boost DC-DC converter the lower rate voltage semiconuctors with lower on-resistance can replace the higher rate voltage evices. In aition the equivalent frequency of the inuctor current an the capacitor voltage ripple in the propose topology is ouble the switching frequency ue to using one aitional power switch ioe an flying capacitor achieve by using the flying-capacitor three-level structure with two phase-shifte 8 egree gate riving signals. These features are beneficial to improve efficiency. In orer to improve the efficiency of the propose converter further the Boost three-level DC-DC converter with a synchronous rectification quasi-z source (BTL-SRqZ) is aitionally propose base on the BTL-DRqZ. This paper is organize as follows: in Section II the topology of the BTL-DRqZ for fuel cell vehicles is presente. The operation principles of the converter topology with a synchronous rectification quasi-z source are iscusse in Section III. In Section IV the parameters of all components are esigne an the losses of the propose topology are analyze. Then the experimental results measure from the prototype are analyze in Section V. Finally the conclusion is elivere in Section VI. II. TPLGY F DRQZ SURCE CNVERTER In orer to wien the step-up voltage gain of the Boost DC-DC converter the DRqZ source net "L -L -D -C -C " has been investigate. The input of the converter is comprise of the voltage source of the fuel cell U FC=U in an its associate reverse blocking ioe D FC. A three-level DC-DC converter with flying-capacitor is aopte to halve the voltage stress on the power evices an also allow U in an the DC link bus to have a common groun. The resulting BTL-DRqZ for a fuel cell vehicle is shown in Fig.. i L i C i D L D L D D 3 D FC U FC=U in C i C C i L S S p i D Q Q n C fly i D3 C o I U DC link Fig. Propose Boost three-level DC-DC converter with ioe rectification quasi-z source (BTL-DRqZ) for fuel cell vehicles. III. PERATIN PRINCIPLES A. peration states Accoring to Fig. there are four switching states "S S " in a switching perio i.e. S S ={ } where "" represents the power switches Q Q "N" an "" represents Q Q "FF". L an L are storing energy while C an C are ischarging energy when S S =. In the other switching states L an L ischarging energy whereas C an C are charging. In aition the sequence of the switching states in a switching perio is relate to the uty cycle ranges of the power switches Q Q. For example Sequence I "----" appears within the range of < = <.5 while Sequence II "----" can be obtaine by the range of.5< = < where an ( = ) are the corresponing uty cycles for Q an Q in a Boost three-level DC-DC converter. However the inuctors L an L only ischarge in Sequence I ue to the absence of switching state S S =. Therefore it is likely that

4 change prior to final publication. Manuscript ID:TPEL-Reg R.> 3 the propose converter operates within the range of.5< = <. In the active switching states the energy flow paths between the fuel cell stack source inuctors an capacitors are shown in Fig. an the PWM moulation strategy an important waveforms are illustrate in Fig. 3. In Fig. (a) there are three energy flow loops when S S =: in loop- L is ischarging at the same time C is charging through D. The inuctor current i L an the capacitor voltage U C are shown in Fig. 3(e f); in loop- L an U in in series are ischarging while C is charging through D FC an D. Thus the inuctor current i L an the capacitor voltage U C can be illustrate in Fig. 3( g); in loop-3 L L an U in in series are ischarging while the flying-capacitor C fly is charging through D FC D D an Q. Hence the corresponing voltage an current waves are shown in Fig. 3( e h j k m). In aition the instantaneous PWM voltage of the converter U pn (S S =) is simply the voltage across C fly namely U pn=u Cfly as shown in Fig. 3(n). When S S = there are also three energy flow paths as shown in Fig. (b). It can be seen that the ifference between S S = an S S = is the ischarging/charging state of the flying-capacitor C fly e.g. C fly U in L an L are in a series connection an ischarge to supply the DC link sie through D FC D Q an D 3. The corresponing voltage an current waveforms are shown in Fig. 3( e h i l m). At the same time the instantaneous PWM voltage of the converter U pn (S S =) is escribe as U pn=u U Cfly rather than the voltage across C fly as shown in Fig. 3(n). In another active switching state S S = D is FF ue to the reverse voltage of L. As a result two energy flow paths are left as shown in Fig. (c). In loop- C (which stays in a series connection with U in) is ischarging while L is charging through D FC Q an Q ; similarly C is transferring energy to L through Q an Q in loop-. Consequently the instantaneous PWM voltage of the converter U pn= (S S =) can be obtaine as shown in Fig. 3(n). DFC Uin DFC Uin i L i C i D C loop- L D L loop- i L i C i D C loop- i C C L D L i C loop- C (a) (b) i L S S i L S S p p i D D loop-3 Q Q n D Q Q n Cfly loop-3 Cfly i D3 D3 Co D3 Co I U I U DC link DC link i C i L L D L i L i C D FC U in C loop- C loop- S S p Q Q n D C fly D 3 C o I U DC link (c) Fig. Energy flow paths among the voltage source inuctors an capacitors in effective switching states. (a) SS= (D is N). (b) SS= (D is N). (c) SS= (D is FF). B. peration with wie range of voltage-gain In orer to simplify the explanation it is assume the capacitance of the capacitors in Fig. is infinite as well as the inuctance of the inuctors. Therefore capacitors C C are seeme to be constant voltage sources an L L can be consiere as constant current sources. In aition the flying-capacitor voltage is half of the output voltage U e.g. U Cfly=U /. When S S = or S S = L an L are ischarging. Thus i L an i L are ientical in Fig. (a b) an the voltages across L an L are also equal (): () u L_is = u L_is By means of Fig. (a b) an KVL (Kirchhoff s Voltage Laws) the voltage balance equations can be obtaine as follows U U in ul_is ul_is ul_is UC () Uin ul_is UC When S S = L an L are charging their voltages an u L_ch u L_ch can be escribe as follows from Fig. (c) an KVL Uin UC ul_ch (3) UC ul_ch Accoring to () an () the ischarging voltage across L can be written as (4) U Uin u L_is (4) while the charging voltage of L is obtaine by virtue of ()~(4) U Uin u L_ch (5) Regaring the charging/ischarging time of L when S S = an S S = the ischarging time t L_is of L is escribe as follows by means of the PWM moulation strategy shown in Fig. 3(a~c) tl_is [( ) ( )] T (6) m while the charging time t L_ch of L is written t [ ( )] T (7) L_ch where = = are the uty cycles of Q an Q respectively m is the moulation inex an T is the carrier perio.

5 change prior to final publication. Manuscript ID:TPEL-Reg R.> 4 m.5 S S i L i L U C U C i D i Q i Q i D i D3 U Cfly U pn carrier T/ - carrier Uo/ (a) T T/ T (n) Fig. 3 PWM moulation strategy an important waveforms. In current continuous moe the voltage-secon balance equation for L can be establishe as follows by means of the equal charging an ischarging energy in each carrier perio u t u t (8) Uo/ L_is L_is L_ch L_ch As a result the step-up voltage-gain M of the BTL-qZ can be obtaine by the combination of (4)~(8) M U (9) U 3 4 where In aition the capacitor voltages across C an C can also be gaine by virtue of () (4) an (9) U C (.5) U () U C ( ) U By means of (9) the propose topology in Fig. has a wier step-up voltage-gain range especially the uty cycles of Q an Q are kept within the range of Consequently the conventional Boost three-level DC-DC converter s ilemma between the high voltage-gain an the non-extreme uty cycles can be solve by the propose topology. In Fig. 4 it is shown the comparison of voltage-gain M via uty cycles among the conventional Boost three-level converter the interleave converter in [3] the common groun converter in [33] an the propose one. Therefore the propose converter in Fig. has a wier range of voltage-gain than those previously presente. Even if it operates with lower voltage-gain (i.e. M=) the more proper uty cycles.5.75 will appear rather than the extreme low uty cycles in [3] an [33]. in (b) (c) () (e) (f) (g) (h) (i) (j) (k) (l) (m) M Fig. 4 Comparison of voltage-gain M via uty cycles among conventional Boost three-level converter interleave converter in [3] common groun converter in [33] an propose one. C. Self-balance of flying-capacitor voltage Accoring to Fig. (a b) L is ischarging an its voltage u L_is is just the voltage across C ul_is UC ( SS ) () When S S = D an Q are N as shown in Fig. (a) so the flying-capacitor voltage U Cfly_ across C fly can be escribe as follows by () UCfly_ UC UC () Similarly when S S = Q an D 3 are N as shown in Fig. (b) the flying-capacitor voltage U Cfly_ can also be obtaine U U ( U U ) (3) Cfly_ C C While S S = Q an Q are N but D an D 3 are FF as shown in Fig. (c). Consequently the flying-capacitor voltage U Cfly_ is maintaine. Accoring to () an(3) it is conclue the flying-capacitor voltage U Cfly irectly epens on the sum of U C an U C from the DRqZ source net. Furthermore the obvious relationship between U Cfly an the output voltage U is euce from () U UCfly (4) From the analysis above it can also be further conclue that the flying-capacitor voltage U Cfly is clampe by the sum of U C an U C from the DRqZ source net an U Cfly can follow half the output voltage U by this self-balance characteristic both in the converter's static an ynamic states. Therefore extra balance controls for the flying-capacitor voltage can be remove an the voltage stress of all power semiconuctors can still be constant at half the output voltage. D. Synchronous rectification operation for quasi-z source Accoring to (4) an Fig. the voltage stress of the power semiconuctors Q Q D an D 3 is half the output voltage. Regaring the voltage stress of D from the DRqZ source system its blocking voltage is just the sum of U C an U C when S S = as shown in Fig. (c). Therefore it is also half the output voltage (). These avantages above are beneficial to reucing the conuction losses by using appropriate semiconuctors which are of lower on-resistance or lower voltage rop. The other cause of the conuction losses is the current flowing through the ioes i.e. D ~D 3 shown in Fig.. The instantaneous currents i D i D through D an D can be escribe as follows when S S = by means of Fig. (a) an KCL (Kirchhoff s Current Laws).

6 change prior to final publication. Manuscript ID:TPEL-Reg R.> 5 i i i i id il ic D L L D ( SS ) (5) where i C> is the instantaneous current flowing through C i L an i L are the instantaneous currents of L an L as shown in Fig. (a). Similarly the instantaneous currents i D an i D3 through D an D 3 can also be written as follows when S S = id il il id3 ( SS ) (6) id3 il ic whilst D ~D 3 are FF when S S =. Therefore when S S ={ } the relationships of i D~i D3 to i L can be obtaine as follows by means of (5)~(6) the referre relations of i C> an i L= i L id il ( SS ) id3 il ( SS ) (7) id il ( SS ) Consequently it is conclue that the instantaneous currents flowing D D 3 of the propose converter are smaller than the corresponing input current of the voltage source. But the instantaneous current flowing in D from the DRqZ source network is larger than the corresponing input current of the voltage source. As a result the conuction loss of D must be the largest among D ~D 3. In aition D can be replace by the synchronous rectification MSFET Q SR (D SR is its anti-parallel boy ioe) which is of lower on-resistance. This propose BTL-SRqZ for the fuel cell vehicles is shown in Fig. 5. The voltage stress of Q SR is also half the output voltage as follows when S S = U UQSR UC UC (8) U in L Q SR D SR C C L D FC D D 3 S SR S S p Q Q n C fly C o U o DC link Fig. 5 Propose Boost three-level DC-DC converter with synchronous rectification quasi-z source (BTL-SRqZ) for fuel cell vehicles. As to the gate riving signal S SR for the synchronous rectification power switch Q SR it can be obtaine from "Exclusive R" logic combining S an S epicte in Fig. 6(a~). In orer to avoi conuction behavior of Q SR uring the state of S S = the ea time t must be ae to the ieal gate riving signal of Q SR by the principle of "FF in avance an N with elay" as shown in Fig. 6(b~). For instance Q SR must be turne off ahea of time by t before the switching state changes to S S = an turne on with elaye time t after S S is change to or. In aition t is etermine by the ea time moulation inex m an carrier perio T easily as follows shown in Fig. 6(a ) T t m (9) The anti-parallel boy ioe D SR conucts when Q SR is turne off in avance an the current flows through D SR instea of Q SR. As a result the voltage stress of Q SR is just the forwar voltage rop of D SR i.e. Q SR is turne off with near Zero-Voltage Switching (ZVS) as shown in Fig. 6(~f). Similarly Q SR is turne on with ZVS. carrier carrier (m+m ) m.5 S S S SR i DSR U QSR t shift T/ t shift (a) T (b) U o / T/ (f) ZVS ZVS trun-off trun-on T Fig. 6 Gate riving signals of synchronous rectification power switches ea time an zero-voltage switching. IV. CMPNENT PARAMETERS DESIGN A. Power switches an ioes From () an (4) it is shown that the voltage U Cfly of the flying capacitor C fly is half of the output voltage U as well as the total voltage of C an C. The voltage stress of the power switches an ioes employe in the propose topology can be euce in terms of the energy flow paths among the voltage source inuctors an capacitors uring their effective switching states as shown in Fig.. When S S = Q an D 3 are in the FF state as shown in Fig. (a). Therefore the blocking voltages of Q an D 3 are U Cfly an (U U Cfly) respectively. When S S = Q an D are turne off as shown in Fig. (b). So the voltage stresses of Q an D are clampe by (U U Cfly) an U Cfly respectively. When S S = D ~D 3 are in the FF state as shown in Fig. (c). As a result the blocking voltages of D ~D 3 are (U C+U C) U Cfly an (U State thus the voltage stresses of all semiconuctors are obtaine as follows U UQ UCfly = U UQ U U Cfly U U D U C U C () U UD UCfly U U D3 U U Cfly With regar to current stresses (namely average currents in the N state) of the semiconuctors Q Q D ~D 3 they can be obtaine as () using the ampere-secon equations of the capacitors C fly an C base on the energy flow paths among the (c) () (e) U Cfly) respectively.

7 change prior to final publication. Manuscript ID:TPEL-Reg R.> 6 voltage source inuctors an capacitors in the effective switching states as shown in Fig.. 4 IQ I IQ I ID ( ) I () 3 4 I ID I ID3 where I Q I Q an I D~I D3 are average currents of Q Q an D ~D 3 when they are in the N state respectively an I is the output loa current. In aition the current stress of D FC is the average current of the inuctor L namely IDFC IL I () 3 4 It is note that when S S = an the current stresses of Q an Q are lower (they are the same as the current stresses of D 3 an D respectively as escribe in ()) while they are as high 4 as ouble the average currents of the inuctors i.e. 3 4 I when S S =. B. Inuctors an capacitors Accoring to the charging an ischarging states of the inuctors L an L as shown in Fig. 3(b~e) L an L are in the charging state when S S =. The inuctances of L an L can be euce as (3) UC Uin L ( ) il fs (3) UC L ( ) il fs where an are the current fluctuations of L an L an f s is the switching frequency. Combining (3) with (9) an () the inuctances of L an L can be obtaine as (4) which relates the output voltage U the inuctor current fluctuations an the switching frequency f s an the uty cycle i L i L i L i L U L ( )( ) il fs (4) U L ( )( ) il fs When S S = C an C are in the ischarging state the capacitances of C an C can be euce as (5) in terms of Fig. (c) an Fig. 3(b~g) ( ) I C (3 4 ) U C fs (5) ( ) I C (3 4 ) U C fs where UC an UC are the capacitor voltage fluctuations of C an C. Regaring the flying capacitor C fly it is ischarge when S S = as shown in Fig. (b) an the capacitance of C fly can be obtaine as I C fly U f (6) where U Cfly Cfly is the capacitor voltage fluctuation of C fly that is not relate with the uty cycle of power switches. In terms of Fig. (b) the output capacitor C is only charge when S S =; the capacitance of C can be euce as I C (7) U f where U is the capacitor voltage fluctuation of C. C. Comparisons with other step-up solutions Accoring to the euce above the comparisons can be rawn between the propose an the other step-up solutions as shown in TABLE I. The conventional Boost an three-level Boost DC-DC converters nee one inuctor respectively but their ieal voltage-gain of /(-) is limite ue to the effects of parasitic resistance an extreme uty cycles. It is note that the voltage stress of four semiconuctors in the three-level Boost DC-DC converter can be reuce a half comparing with that of the conventional one ue to using two aitional semiconuctors an one flying capacitor. The high voltage-gain step-up DC-DC converters in [3] an [33] nee two inuctors respectively. Although six semiconuctors are employe in the converter without the snubber circuit in [3] there still exist two ioes with the voltage stress of U an its maximum conversion efficiency is about 9.6%. While a maximum conversion efficiency of the converter in [33] is improve to 94% three semiconuctors an three capacitors are neee. However the voltage stress of all the semiconuctors is between U / an U e.g. 3U /4 rather than U /. Regaring the propose converter the number of main components is between those of the converters in [3] an [33] the voltage stress of all the semiconuctors is U / an its maximum conversion efficiency can be 95.66% which is higher than those in [3] an [33]. V. EXPERIMENTAL RESULTS AND ANALYSIS In orer to verify the feasibility an effectiveness of the propose BTL-SRqZ for fuel cell vehicles a scale-own. kw BTL-SRqZ converter prototype was constructe as shown in Fig. 7. In the experiment the fuel cell stack source U FC=U in is replace by an ajustable DC voltage source with a range of U in=6~5v an the converter voltage loop is controlle by a TMS3F8335 DSP. The power circuit IXTKN3P MSFETs (its rate voltage is 3V an its rate current is A while the output voltage of the converter is U =4V) an DSEC6-3A Schottky Barrier Dioes are use. In aition the switching frequency is f s= khz the ea time is t = μs the initial values of the qz source inuctors are L =8 μh an L =5 μh respectively the loa resistor is R L=33~4 an the reference output voltage is 4V. The main experimental parameters of the propose converter are shown in TABLE II. s s

8 change prior to final publication. Manuscript ID:TPEL-Reg R.> 7 Step-up Solutions Conventional Boost Three-level Boost Converter without snubber in [3] Converter in [33] Propose converter Voltage Gain (< <) (< <) (< <) ( ) (< <.5) 3 4 (.5<= <.75) TABLE I Comparisons between propose an other step-up solutions. Amount of Amount of Amount of Voltage Semiconuctors Inuctors Capacitors Stress U U U U U ( ) U Current Stress.5 I I I.5 I ( ) I I I I (3 4 )( ) I I Maximum Efficiency % 94% 95.66% TABLE II Main experimental parameters of propose converter. Parameters an components Values (units) Rate power Pn.kW Input c voltage Uin 6~5V utput c voltage U 4V Switching frequency fs khz Dea time t μs Inuctor L 8 μh Inuctor L 5 μh Capacitors C C Cfly 45V/66 μf Capacitor C 45V/44 μf Loa RL 33~4 MSFETs Q Q QSR IXTKN3P (3V/A) Dioes D D3 DFC DSEC6-3A (3V/6A) Fig. 7 Experimental prototype. Even when the input voltage is U in=4v the experimental PWM voltage U pn is shown in Fig. 8 an the frequency of U pn is ouble of the switching frequency. Although the step-up voltage-gain (U /U in) is the actual uty cycles (= = =-.3=.7) are about.7 instea of the actual extreme value of the typical boost converter which is more than.9 uner the action of the voltage control loop. Furthermore the amplitue of U pn is V (alternating with the flying-capacitor voltage U Cfly an U U Cfly) namely half the output voltage. Thus it verifies U Cfly=U / in the steay state an the flying-capacitor voltage self-balances well without any extra controls. U pn (5V/iv) U Cfly =V U -U Cfly =V U in =4V (V/iv) ( ) T 3μs T / 5μs Half switching perio t(μs/iv) Fig. 8 utput PWM voltage when input voltage Uin=4V an M=. The experimental results of the synchronous rectification ZVS for the SRqZ source system are shown in Fig. 9. Because of the ea time t = μs Q SR is boun to be turne on with a elay an the anti-parallel boy ioe D SR is conucte uring the ea time. It is notice that the voltage stress of Q SR changes from the forwar voltage rop of D SR to half the output voltage uring the ea time. Therefore Q SR can be turne off with ZVS as shown in Fig. 9. Similarly the voltage stress of Q SR changes from half the output voltage to the forwar voltage rop of D SR uring the ea time. Thus Q SR can be turne on with ZVS. ZVS Turn-off U GS (5V/iv) U DS (5V/iv) ZVS Turn-on t(4μs/iv) Fig. 9 Experimental results of synchronous rectification ZVS. As to the applicability of the propose converter for the fuel

9 change prior to final publication. Manuscript ID:TPEL-Reg R.> 8 cell vehicles the experimental results in which the input voltage U in is change graually from the wie range of V to 4V over ozens of secons are shown in Fig. (a). It is seen that the output voltage U nearly stays aroun the reference voltage 4V uner the action of the voltage control loop an the wie step-up voltage-gain (U /U in) range changes from 3.3 to. In fact the actual voltage-gain in the voltage control loop is more than 3.3 to ue to the losses compensation of the converter's operation. Corresponingly the input current (i L) increases graually with the wie-range change input voltage (from V to 4V) as shown in Fig. (b) when the loa is constant. V U o =4V (V/iv) U in =V~4V (V/iv) (a) 4V t(4s/iv) V V U (V/iv) U Cfly (5V/iv) In ynamic State I V In static state 4V In ynamic State II t(4s/iv) Fig. Dynamic flying capacitor voltage corresponing to the variable output voltage U=~4V in open loop. qz source are charge an ischarge twice uring each switching perio. Compare with the converter in [33] there are one aitional active power switch an two more ioes in the propose converter. However the equivalent switching frequency of the propose converter is ouble the one of the converter in [33]. All the volumes of capacitors an inuctors in the quasi-z-source can be reuce by almost a half compare with those of the converter in [33]. In aition the quasi-z-source capacitor voltage stresses are lower than those of the converter in [33]. Therefore the volume of the propose converter can be significantly reuce compare to that of the converter in [33]. V U in =V~4V (V/iv) U /=V S S = S S = U pn (5V/iv) 4V i L (5A/iv) t(4s/iv) T μs Switching perio i L (A/iv) t(μs/iv) (b) Fig. utput voltage an inuctor current with wie-range change input voltage from V to 4V in ynamic state. (a) utput an input voltages. (b) Input current an voltage. In Fig. the flying-capacitor voltage U Cfly is change accoring to the output voltage U (between V an 4V in the open loop) in the static an ynamic states. It is notice that the flying-capacitor voltage U Cfly still keeps at half of the output voltage U especially in the ynamic states I an II. Because the voltage across the flying-capacitor is clampe by the total voltages of the qz source capacitors whose voltages are relate to the corresponing real-time uty cycles an the output voltage U. Uner the voltage control loop the propose BTL-SRqZ converter operates well in conitions of the output voltage U =4V an the output power P =. kw. The output PWM voltage U pn an the inuctor current i L are shown in Fig. (a). The inuctor L is charge when the instantaneous PWM voltage of U pn is zero (S S =). Then the inuctor L is ischarge when U pn stays at U /=V (S S = or ). In aition the current i L of the inuctor L is nearly the same as that of L as shown in Fig. (b). Therefore the inuctors of the (a) Switching perio T μs i L (A/iv) i L (A/iv) t(μs/iv) (b) Fig. utput PWM voltage an inuctor currents. (a) utput PWM voltage an inuctor current. (b) Inuctor currents. In orer to valiate the ynamic behavior of the propose converter an experiment was carrie out which use a step change of loa between 33Ω an Ω an the output voltage an inuctor current are shown in Fig. 3. The inuctor currents (e.g. i L) have corresponing responses between 8A an A an the output voltage U nearly keeps at constant 4V with

10 change prior to final publication. Manuscript ID:TPEL-Reg R.> 9 the voltage loop. It can be seen that i L changes to A from 8A over ms with the loa step-change from Ω to 33Ω an it recovers from A to 8A over ms with the loa step-change from 33Ω to Ω. U (V/iv) i L (5A/iv) Loa step-change from Ω to 33Ω Loa step-change from 33 Ω to Ω t(ms/iv) Fig. 3 utput voltage an inuctor current when loa step-change between 33Ω an Ω. For the wie input-voltage range operation of the propose converter the conversion efficiencies relate to the variable input voltages (e.g. 6V 8V 4V 5V) an the ifferent output powers (e.g. 4W 8W W) are measure by a Power Analyzer (Yokogawa-WT3). Then the relationship between the efficiency the variable input voltages an the ifferent output powers in SR operation are illustrate in Fig. 4. It is notice that the maximum measure efficiency in SR operation is about 95.66% as shown in Fig. 4. In aition when the output power is constant an the input voltage eclines the efficiency ecreases corresponingly ue to the increasing losses cause by the growing input current. In the same conitions above the efficiencies in DR operation are also measure an the SR efficiency is higher than that of DR. The minimum efficiency ifference area appears aroun the meium input voltage (U in=v) an its average efficiency ifference is about.6%. While the maximum efficiency ifference area exists aroun the lower an higher input voltage (U in=8v an 5V) areas an its average value is near.85%. Efficiency (%) U in (V) Fig. 4 Relationship between efficiency variable input voltages an ifferent output powers in SR operation. The calculate loss istributions for the experiment when U in=5v an P =W are shown in Fig. 5. In DR operation the total losses of the converter are 57.6W an the loss istribution is shown in Fig. 5(a). The turn-on an turn-off (switching) an conuction losses of Q an Q account for 39.87% of the total losses. The conuction losses of all ioes D -D 3 an D FC account for 4.57% of the total losses which is a little more than the switching an conuction losses of Q an Q ue to the higher conuction loss of D (in the quasi-z-source). However the total losses of the converter are reuce to 49.6W in the SR operation an the loss istribution is shown in Fig. 5(b). The switching an conuction losses of Q an Q account for 46.9% of the total losses an the conuction losses of D D 3 D FC an Q SR are reuce to 3.3% of the total losses ue to the SR operation of Q SR instea of D in the quasi-z-source. Conuction losses of ioes (4.57%) 3.7W Copper losses (.%) 6.4W 3.47W Conuction losses of Q an Q (6.8%) (a) Capacitor losses (4.75%) Conuction losses of Q an Q (7.4%) Capacitor losses (4.%) Core losses.34w.85w (3.4%) Switching losses of Q an Q (33.79%) 9.8W Core losses (3.75%) Copper losses.34w.85w (.99%) 6.4W Conuction losses of ioes (8.4%) 3.99W 9.8W 3.47W.93W Switching losses of Q an Q (39.5%) Conuction losses of QSR (3.9%) (b) Fig. 5 Calculate loss istributions for experiment when Uin=5V an P=W. (a) In DR operation. (b) In SR operation. VI. CNCLUSIN The topology of the BTL-SRqZ is propose in this paper. It has the avantages of lower voltage stress for the power semiconuctors an the common groun between the input an output sies as well as the wier range of the voltage-gain with moest uty cycles.5.75 for the power switches. In aition the voltage of the flying-capacitor can be clampe well at half the output voltage by the capacitor voltages of the quasi-z source net in both the static an ynamic states. At the same time the synchronous rectification power switch operates with ZVS turn-on an turn-off an the losses of the quasi-z source circuit can be reuce by the synchronous rectification operation. Therefore it is suitable to vehicles powere by a fuel cell stack which has a soft output characteristic. REFERENCES [] C. Jin X. Sheng an P. Ghosh ptimize electric vehicle charging with intermittent renewable energy sources IEEE Journal of Selecte Topics in Signal Processing vol. 8 no. 6 pp Dec. 4. [] B. Zeng J. Zhang X. Yang J. Wang J. Dong an Y. Zhang Integrate planning for transition to low-carbon istribution system with renewable energy generation an eman response IEEE Trans. Power Syst. vol. 9 no. 3 pp May 4. [3] A. Soroui R. Caire N. Hajsai an M. Ehsan Probabilistic ynamic multi-objective moel for renewable an non-renewable istribute generation planning IET Gener. Transm. Distrib. vol. 5 no. pp May. PQ PD Pcu PC Pfe P PQ PDSR Pcu PC Pfe P PQSR

11 change prior to final publication. Manuscript ID:TPEL-Reg R.> [4] K. Li T. Chen Y. Luo an J. Wang Intelligent environment-frienly vehicles: concept an case stuies IEEE Trans. Intelligent Transportation Systems vol. 3 no. pp Mar.. [5] A. T-Raissi an D. L. Block Hyrogen: automotive fuel of the future IEEE Power & Energy Magazine vol. no. 6 pp Nov. 4. [6] A. S. Samosir an A. H. M. Yatim Implementation of ynamic evolution control of biirectional DC DC converter for interfacing ultracapacitor energy storage to fuel-cell system IEEE Trans. In. Electron. vol. 57 no. pp ct.. [7] G. Fontes C. Turpin an S. Astier A Large-signal an ynamic circuit moel of a H / PEM fuel cell: escription parameter ientification an exploitation IEEE Trans. In. Electron. vol. 57 no. 6 pp Jun.. [8] A. Askarzaeh an A. Rezazaeh An innovative global harmony search algorithm for parameter ientification of a PEM fuel cell moel IEEE Trans. In. Electron. vol. 59 no. 9 pp Sep.. [9] J. Morales-Morales I. Cervantes an U. Cano-Castillo n the esign of robust energy management strategies for FCHEV IEEE Trans. Veh. Technol. vol. 64 no. 5 pp May 5. [] G. Su an L. Tang A reuce-part triple-voltage DC DC converter for EV/HEV power management IEEE Trans. Power Electron. vol. 4 no. pp ct. 9. [] U. R. Prasanna an A. K. Rathore Dual three-pulse moulation-base high-frequency pulsating DC link two-stage three-phase inverter for electric/hybri/fuel cell vehicles applications IEEE Journal of Emerging an Selecte Topics in Power Electronics vol. no. 3 pp ct. 4. [] J. Jia G. Wang Y. T. Cham Y. Wang an M. Han Electrical characteristic stuy of a hybri PEMFC an ultracapacitor system IEEE Trans. In. Electron. vol. 57 no. 6 pp Dec.. [3] N. D. Benavies an P. L. Chapman Mass-optimal esign methoology for DC-DC converters in low-power portable fuel cell applications IEEE Trans. Power Electron. vol. 3 no. 3 pp May 8. [4] G. Dotelli R. Ferrero P. G. Stampino S. Latorrata an S. Toscani PEM fuel cell rying an flooing iagnosis with signals injecte by a power converter IEEE Trans. Instrum. Meas. vol. 64 no. 8 pp Aug. 5. [5] S. Dusmez A. Hasanzaeh an A. Khaligh Comparative analysis of biirectional three-level DC DC converter for automotive applications IEEE Trans. In. Electron. vol. 6 no. 5 pp May 5. [6] H. Chen an J. Liao Design an implementation of sensorless capacitor voltage balancing control for three-level Boosting PFC IEEE Trans. Power Electron. vol. 9 no. 7 pp Jul. 4. [7] X. Ruan B. Li Q. Chen S. Tan an C. K. Tse Funamental consierations of three-level DC DC converters: topologies analyses an control IEEE Trans. Circuits Syst. I Reg. Papers vol. 55 no. pp Dec. 8. [8] L. Shi B. P. Baipaiga M. Ferowsi an M. L. Crow Improving the ynamic response of a flying-capacitor three-level Buck converter IEEE Trans. Power Electron. vol. 8 no. 5 pp May 3. [9] Y. J. A. Alcazar D. S. liveira Jr. F. L. Tofoli an R. P. Torrico-Bascopé DC DC nonisolate Boost converter base on the three-state switching cell an voltage multiplier cells IEEE Trans. In. Electron. vol. 6 no. pp ct. 3. [] Y. Tang D. Fu T. Wang an Z. Xu Hybri switche-inuctor converters for high step-up conversion IEEE Trans. In. Electron. vol. 6 no. 3 pp Mar. 5. [] W. Qian D. Cao J. G. Cintrón-Rivera M. Gebben D. Wey an F. Z. Peng A switche-capacitor DC DC converter with high voltage gain an reuce component rating an count IEEE Trans. In. Appl. vol. 48 no. 4 pp Jul./Aug.. [] A. M. Jorge L. Rorigo P. Elvia an L. G. Jorge Moelling an control of a DC DC quaratic boost converter with R P IET Power Electron. vol. 7 no. pp. May 4. [3] R. Gules L. L. Pfitscher an L. C. Franco An interleave Boost DC-DC converter with large conversion ratio ISIE'3 pp Jun. 3. [4] Q. Zhao an F. C. Lee High-efficiency high step-up DC DC converters IEEE Trans. Power Electron. vol. 8 no. pp Jan. 3. [5] G. A. L. Henn R. N. A. L. Silva P. P. Praça L. H. S. C. Barreto an D. S. liveira Interleave-Boost converter with high voltage gain IEEE Trans. Power Electron. vol. 5 no. pp Nov.. [6] P. P. Praça G. A. L. Henn D. S. liveira L. H. S. C. Barreto an R. N. A. L. Silva High voltage gain single stage DC-DC converter base on three-state commutation cell APEC'3 pp Mar. 3. [7] S. J. Amoeo H. G. Chiacchiarini an A. R. liva High-performance control of a DC DC Z-source converter use for an excitation fiel river IEEE Trans. Power Electron. vol. 7 no. 6 pp Jun.. [8] V. P. Galigekere an M. K. Kazimierczuk Analysis of PWM Z-source DC-DC converter in CCM for steay state IEEE Trans. Circuits Syst. I Reg. Papers vol. 59 no. 4 pp Apr.. [9] S. Yang F. Z. Peng Q. Lei R. Inoshita an Z. Qian Current-fe quasi-z-source inverter with voltage Buck Boost an regeneration capability IEEE Trans. In. Appl. vol. 47 no. pp Mar./Apr.. [3]. Ellabban H. Abu-Rub an B. Ge A quasi-z-source irect matrix converter feeing a vector controlle inuction motor rive IEEE Journal of Emerging an Selecte Topics in Power Electronics vol. 3 no. pp Jun. 5. [3] K. Patiar an A. C. Umarikar High step-up pulse-with moulation DC DC converter base on quasi-z-source topology IET Power Electron. vol. 8 no. 4 pp [3] B. Poorali A. Torkan an E. Aib High step-up Z-source DC DC converter with couple inuctors an switche capacitor cell IET Power Electron. vol. 8 no. 8 pp [33] H. Shen B. Zhang D. Qiu an L. Zhou A common groune Z-source DC-DC converter with high voltage gain IEEE Trans. In. Electron. vol. 63 no. 5 pp Yun Zhang (M 3) was born in Jiangsu China in 98. He receive the B.S. an M.S. egrees in electrical engineering from the Harbin University of Science an Technology Harbin China in 3 an 6 respectively an the Ph.D. egree in electrical engineering from the Harbin Institute of Technology Harbin in. He is currently an Associate Professor at the School of Electrical Engineering an Automation Tianjin University Tianjin China. His current research interests inclue topologies moulation an control strategies of power converters for microgri an electric vehicles. Jilong Shi was born in Shanong China. He receive his B.S. egree in Electrical Engineering from the Changchun University f Science an Technology Changchun China in 4. He starte pursing his M.S. egree in Electrical Engineering from Tianjin University Tianjin China in 4. His current research interests inclue DC-DC converters an fuel cell vehicles. Lei Zhou was born in Ningxia China. He receive his B.S. egree in Electrical Engineering from the Tianjin University Tianjin China in 5. He starte pursing his M.S. egree in Electrical Engineering from Tianjin University Tianjin China in 5. His current research interests inclue DC-DC converters moeling an analysis of DC-DC converters.

12 change prior to final publication. Manuscript ID:TPEL-Reg R.> Jing Li (M 5) receive the B.Eng. (Hons.) an M.Sc. (Distinction) egrees from the Beijing Institute of Technology Beijing China in 999 an respectively an the Ph.D. egree from the University of Nottingham Nottingham U.K. in. She was a Research Fellow with the Power Electronic Machine an Control Group University of Nottingham. She is currently a Lecturer at the Department of Electrical an Electronic Engineering University of Nottingham Ningbo China. Her research interests inclue conition monitoring for motor rive systems an power istribution systems an avance control an esign of motor rive systems. Mark Sumner (SM 5) receive the B.Eng. egree in electrical an electronic engineering from Lees University Lees U.K. in 986 an the Ph.D. egree in inuction motor rives from the University of Nottingham Nottingham U.K. in 99. He was with Rolls Royce Lt. Ansty U.K. He was a Research Assistant with the University of Nottingham where he became a Lecturer in ctober 99 an is currently a Professor of electrical energy systems. His research interests inclue control of power electronic systems incluing sensorlessmotor rives iagnostics an prognostics for rive systems power electronics for enhance power quality an novel power system fault location strategies. Ping Wang was born in Tianjin China in 959. She receive the B.S. M.S. an Ph.D.egrees in Electrical Engineering from the Tianjin University Tianjin China in an 5 respectively. Since 98 she has been a Teacher an a Researcher at Tianjin University where she is currrently a Professor. Her current research interests inclue power electronic control of renewable energy sources PWM converters an intelligent etection an control. Changliang Xia (M 8-SM ) was born in Tianjin China in 968. He receive the B.S. egree from Tianjin University China in 99 an the M.S. an Ph.D. egrees from Zhejiang University China in 993 an 995 respectively all in electrical engineering. He is currently a Professor in the School of Electrical Engineering an Automation Tianjin University an also in Tianjin Key Laboratory of Avance Technology of Electrical Engineering an Energy Tianjin Polytechnic University. In 8 he became Yangtze Fun Scholar Distinguishe Professor an is currently supporte by National Science Fun for Distinguishe Young Scholars. His research interests inclue electrical machines an their control systems power electronics an control of win generators.