FEATURES DESCRIPTIO APPLICATIO S TYPICAL APPLICATIO LTC1562 Very Low Noise, Low Distortion Active RC Quad Universal Filter

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1 FEATRES Continuous Time No Clock Four nd Order Filter Sections, khz to khz Center Frequency ±.% Typical Center Frequency Accuracy ±.3% Typical Center Frequency Accuracy (A Grade) Wide Variety of Response Shapes Lowpass, Bandpass and Highpass Responses 3dB Typical S/N, ±V Supply (Q = ) 7dB Typical S/N, Single V Supply (Q = ) db Typical S/(N + THD) at ±V Supply, khz Input Rail-to-Rail Input and Output Voltages DC Accurate to 3mV (Typ) Zero-Power Shutdown Mode Single or Dual Supply, V to V Total Resistor-Programmable f O, Q, Gain APPLICATIO S High Resolution Systems (4 Bits to Bits) Antialiasing/Reconstruction Filters Data Communications, Equalizers Dual or I-and-Q Channels (Two Matched 4th Order Filters in One Package) Linear Phase Filtering Very Low Noise, Low Distortion Active RC Quad niversal Filter DESCRIPTIO The LTC is a low noise, low distortion continuous-time filter with rail-to-rail inputs and outputs, optimized for a center frequency (f O ) of khz to khz. nlike most monolithic filters, no clock is needed. Four independent nd order filter blocks can be cascaded in any combination, such as one th order or two 4th order filters. Each block s response is programmed with three external resistors for center frequency, Q and gain, using simple design formulas. Each nd order block provides lowpass and bandpass outputs. Highpass response is available if an external capacitor replaces one of the resistors. Allpass, notch and elliptic responses can also be realized. The is designed for applications where dynamic range is important. For example, by cascading nd order sections in pairs, the user can configure the IC as a dual 4th order Butterworth lowpass filter with 4dB signal-to-noise ratio from a single V power supply. Low level signals can exploit the built-in gain capability of the. Varying the gain of a section can achieve a dynamic range as high as db with a ±V supply. Other cutoff frequency ranges can be provided upon request. Please contact LTC Marketing. Replacing LC Filter Modules, LTC and LT are registered trademarks of Linear Technology Corporation. TYPICAL APPLICATIO Dual 4th Order khz Butterworth Lowpass Filter R IN k V R IN3 k R Q,.k R, k R3, k R Q3,.k 3 R IN, k INV B V B V B INV C V C V C V + V SHDN V A V A INV A AGND V D V D INV D R IN4, k R Q, 3k R, k 3 R4, k R Q4, 3k TA V OT V V OT SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V SEE TYPICAL APPLICATIONS FOR OTHER CTOFF FREQENCIES DC ACCRATE, NONINVERTING, NITY-GAIN, RAIL-TO-RAIL INPT AND OTPTS. PEAK SNR db WITH ±V SPPLIES k k FREQENCY (Hz) TA3b M fa

2 ABSOLTE AXI RATI GS W W W (Note ) Total Supply Voltage (V + to V )... V Maximum Input Voltage at Any Pin...(V.3V) V (V + +.3V) Storage Temperature Range... C to C W PACKAGE/ORDER I FOR ATIO PACKAGE/ORDER INFORMATION W Operating Temperature Range C... C to 7 C I... 4 C to C Lead Temperature (Soldering, sec)... 3 C INV B V B V B V * V + SHDN V * V A V A INV A TOP VIEW INV C V C V C V * V AGND V * V D V D INV D G PACKAGE -LEAD PLASTIC SSOP *G PACKAGE PINS 4, 7, 4, 7 ARE SBSTRATE/SHIELD CONNECTIONS AND MST BE TIED TO V T JMAX = C, θ JA = 3 C/W ORDER PART NMBER CG ACG IG AIG INV B V B V B V + SHDN V A V A INV A TOP VIEW INV C V C 4 V C 3 V AGND V D V D INV D N PACKAGE -LEAD PDIP T JMAX = C, θ JA = C/W ORDER PART NMBER CN Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The denotes the specifications that apply over the full operating temperature range, otherwise specifications are at T A = C. V S = ±V, outputs unloaded, SHDN pin to logic low, unless otherwise noted. AC specs are for a single nd order section, R IN = R = R Q =k ±.%, f O = khz, unless noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX NITS V S Total Supply Voltage 4.7. V I S Supply Current V S = ±.37V, R L = k, C L = 3pF, Outputs at V 7.3. ma V S = ±V, R L = k, C L = 3pF, Outputs at V. ma V S = ±.37V, R L = k, C L = 3pF, Outputs at V 3. ma V S = ±V, R L = k, C L = 3pF, Outputs at V. ma Output Voltage Swing V S = ±.37V, R L = k, C L = 3pF V P-P V S = ±V, R L = k, C L = 3pF.3. V P-P V OS DC Offset Magnitude, V Outputs V S = ±.37V, Input at AGND Voltage 3 mv (Lowpass Response) V S = ±V, Input at AGND Voltage 3 mv DC AGND Reference Point V S = Single V Supply. V Center Frequency (f O ) Error (Note ) (SSOP) V S = ±V, V Output Has R L = k, C L = 3pF.. % A (SSOP) V S = ±V, V Output Has R L = k, C L = 3pF.3. % (PDIP) V S = ±V, V Output Has R L = k, C L = 3pF.. % H L LP Passband Gain (V Output) V S = ±.37V, f IN = khz, db V Output Has R L = k, C L = 3pF H B BP Passband Gain (V Output) V S = ±.37V, f IN = f O, db V Output Has R L = k, C L = 3pF fa

3 ELECTRICAL CHARACTERISTICS The denotes the specifications that apply over the full operating temperature range, otherwise specifications are at T A = C. V S = ±V, outputs unloaded, SHDN pin to logic low, unless otherwise noted. AC specs are for a single nd order section, R IN = R = R Q =k ±.%, f O = khz, unless noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX NITS Q Error V S = ±.37V, LP Output Has R L = k, C L = 3pF +3 % Wideband Output Noise, V S = ±.37V, BW = khz, Input AC GND 4 µv RMS Lowpass Response (V Output) V S = ±V, BW = khz, Input AC GND 4 µv RMS Input-Referred Noise, Gain = BW = khz, f O = khz, Q =, Input AC GND 4. µv RMS THD Total Harmonic Distortion, f IN = khz,.v P-P, V and V Outputs Have db Lowpass Response (V Output) R L = k, C L = 3pF f IN = khz,.v P-P, V and V Outputs Have 7 db R L = k, C L = 3pF Shutdown Supply Current SHDN Pin to V +. µa SHDN Pin to V +, V S = ±.37V. µa Shutdown-Input Logic Threshold. V Shutdown-Input Bias Current SHDN Pin to V µa Shutdown Delay SHDN Pin Steps from V to V + µs Shutdown Recovery Delay SHDN Pin Steps from V + to V µs Inverting Input Bias Current, Each Biquad pa Note : Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note : f O change from ±V to ±.37 supplies is.% typical, f O temperature coefficient, 4 C to C, is ppm/ C typical. TYPICAL PERFOR A CE CHARACTERISTICS W fo ERROR (%) f O Error vs Nominal f O (V S = ±V) Q = Q =. Q = NOMINAL f O (khz) f O ERROR (%) f O Error vs Nominal f O (V S = ±.V) Q = Q =. Q = NOMINAL f O (khz) Q ERROR (%) 3 3 Q Error vs Nominal f O (V S = ±V) T A = 7 C T A = C R IN = R Q Q = NOMINAL f O (khz) Q = Q =. Q = G G G3 fa 3

4 TYPICAL PERFOR A CE CHARACTERISTICS W Q ERROR (%) 3 3 Q Error vs Nominal f O (V S = ±.V) T A = 7 C T A = C R IN = R Q Q = Q = Q =. Q = PEAK BP Peak BP Gain vs Nominal f O (V S = ±V) (Figure 3, V Output) R IN = R Q T A = 7 C T A = C Q = Q = Q =. Q = PEAK BP Peak BP Gain vs Nominal f O (V S = ±.V) (Figure 3, V Output) R IN = R Q T A = 7 C T A = C Q = Q = Q =. Q = NOMINAL f O (khz) NOMINAL f O (khz) NOMINAL f O (khz) G4 G G NOISE (µvrms) LP Noise vs Nominal f O (V S = ±V, C) (Figure 3, V Output) (R IN = R) Q = Q =. Q = NOMINAL f O (khz) BP NOISE (µvrms) BP Noise vs Nominal f O (V S = ±V, C) (Figure 3, V Output) (R IN = R Q ) Q = Q =. Q = NOMINAL f O (khz) THD (AMPLITDE BELOW FNDAMENTAL) (db) k Distortion vs External Load Resistance (V S = ±V, C) (Figure ) nd ORDER LOWPASS f O = khz Q =.7 OTPT LEVEL V RMS (.3V P-P ) ±V SPPLIES f IN = khz f IN = khz k k EXTERNAL LOAD RESISTANCE (Ω) k G7 G G PI F CTIO S Power Supply Pins: The V + and V pins should be bypassed with capacitors to an adequate analog ground or ground plane. These capacitors should be connected as closely as possible to the supply pins. In the -lead SSOP package, the additional pins 4, 7, 4 and 7 are internally connected to V (Pin ) and should also be tied to the same point as Pin for best shielding. Low noise linear supplies are recommended. Switching supplies are not recommended as they will lower the filter dynamic range. Analog Ground (AGND): The AGND pin is the midpoint of an internal resistive voltage divider, developing a potential halfway between the V + and V pins, with an equivalent series resistance nominally 7kΩ. This serves as an internal ground reference. Filter performance will reflect the quality of the analog signal ground and an analog ground plane surrounding the package is recommended. The analog ground plane should be connected to any digital ground at a single point. For dual supply operation, the AGND pin should be connected to the ground plane 4 fa

5 PIN FNCTIONS (Figure ). For single supply operation, the AGND pin should be bypassed to the ground plane with at least a capacitor (at least µf for best AC performance) (Figure ). These figures show -pin package connections. The same principles apply to the -pin package with allowance for its different pin numbers. The -pin ANALOG GROND PLANE V SINGLE-POINT SYSTEM GROND -PIN SSOP V DIGITAL GROND PLANE (IF ANY) package does not have the four substrate pins (Pins 4, 7, 4, 7 in the -pin package). Shutdown (SHDN): When the SHDN input goes high or is open-circuited, the enters a zero-power shutdown state and only junction leakage currents flow. The AGND pin and the amplifier outputs (see Figure 3) assume a high impedance state and the amplifiers effectively disappear from the circuit. (If an input signal is applied to a complete filter circuit while the is in shutdown, some signal will normally flow to the output through passive components around the inactive op amps.) A small pull-up current source at the SHDN input defaults the to the shutdown state if the SHDN pin is left floating. Therefore, the user must connect the SHDN pin to a logic low (V for ±V supplies, V for V total supply) for normal operation of the. (This convention permits true zero-power shutdown since not even the driving logic must deliver current while the part is in shutdown.) With a single supply voltage, use V for logic low do not connect SHDN to the AGND pin. F Figure. Dual Supply Ground Plane Connection (Including Substrate Pins 4, 7, 4, 7) /4 src* *R AND C ARE PRECISION INTERNAL COMPONENTS C ANALOG GROND PLANE V + 7 -PIN SSOP 4 3 µf V R Z IN INV R Q V F V + / REFERENCE + SINGLE-POINT SYSTEM GROND DIGITAL GROND PLANE (IF ANY) Z IN TYPE R C RESPONSE AT V BANDPASS HIGHPASS RESPONSE AT V LOWPASS BANDPASS IN EACH CASE, f kω O = (khz) ( R ) Q = RQ R ( ) khz f O F Figure. Single Supply Ground Plane Connection (Including Substrate Pins 4, 7, 4, 7) Figure 3. Equivalent Circuit of a Single nd Order Section (Inside Dashed Line) Shown in Typical Connection. Form of Z IN Determines Response Types at the Two Outputs (See Table) fa

6 PIN FNCTIONS INV A, INV B, INV C, INV D: Each of the INV pins is a virtualground summing point for the corresponding nd order section. For each section, external components Z IN, R, R Q connect to the INV pin as shown in Figure 3 and described further in the Applications Information. Note that the INV pins are sensitive internal nodes of the filter and will readily receive any unintended signals that are capacitively coupled into them. Capacitance to the INV nodes will also affect the frequency response of the filter sections. For these reasons, printed circuit connections to the INV pins must be kept as short as possible, less than one inch (.cm) total and surrounded by a ground plane. V A, V B, V C, V D: Output Pins. Provide a bandpass, highpass or other response depending on external circuitry (see Applications Information section). Each V pin also connects to the R Q resistor of the corresponding nd order filter section (see Figure 3 and Applications Information). Each output is designed to drive a nominal net load of kω and 3pF, which includes the loading due to the external R Q. Distortion performance improves when the outputs are loaded as lightly as possible. Some earlier literature refers to these outputs as BP rather than V. V A, V B, V C, V D: Output Pins. Provide a lowpass, bandpass or other response depending on external circuitry (see Applications Information section). Each V pin also connects to the R resistor of the corresponding nd order filter section (see Figure 3 and Applications Information). Each output is designed to drive a nominal net load of kω and 3pF, which includes the loading due to the external R. Distortion performance improves when the outputs are loaded as lightly as possible. Some earlier literature refers to these outputs as LP rather than V. BLOCK DIAGRA W Overall Block Diagram Showing Four 3-Terminal nd Order Sections INV V V INV V V A B C C V + V V + SHTDOWN SWITCH + + R ND ORDER SECTIONS R SHDN V SHTDOWN SWITCH AGND D + C C + C INV V V V V BD fa

7 APPLICATIONS INFORMATION Functional Description W The contains four matched, nd order, 3-terminal universal continuous-time filter blocks, each with a virtual-ground input node (INV) and two rail-to-rail outputs (V, V). In the most basic applications, one such block and three external resistors provide nd order lowpass and bandpass responses simultaneously (Figure 3, with a resistor for Z IN ). The three external resistors set standard nd order filter parameters f O, Q and gain. A combination of internal precision components and external resistor R sets the center frequency f O of each nd order block. The is trimmed at manufacture so that f O will be khz ±.% (±.% typical for PDIP package) if the external resistor R is exactly k. However, lowpass/bandpass filtering is only one specific application for the nd order building blocks in the. Highpass response results if the external impedance Z IN in Figure 3 becomes a capacitor C IN (whose value sets only gain, not critical frequencies) as described below. Responses with zeroes are available through other connections (see Notches and Elliptic Responses). Moreover, the virtual-ground input gives each nd order section the built-in capability for analog operations such as gain (preamplification), summing and weighting of multiple inputs, handling input voltages beyond the power supplies or accepting current or charge signals directly. These Operational Filter TM frequency-selective building blocks are nearly as versatile as op amps. The user who is not copying exactly one of the Typical Applications schematics shown later in this data sheet is urged to read carefully the next few sections through at least Signal Swings, for orientation about the, before attempting to design custom application circuits. Also available free from LTC, and recommended for designing custom filters, is the general-purpose analog filter design software FilterCAD TM for Windows. This software includes tools for finding the necessary f, Q and gain parameters to meet target filter specifications such as frequency response. Setting f O and Q Each of the four nd order sections in the can be programmed for a standard filter function (lowpass, bandpass or highpass) when configured as in Figure 3 with a resistor or capacitor for Z IN. These transfer functions all have the same denominator, a complex pole pair with center frequency ω O = πf O and quality parameter Q. (The numerators depend on the response type as described below.) External resistors R and R Q set f O and Q as follows: k fo = = khz C R R Ω R ( ) π ( ) Q RQ RQ RQ khz = = = ( R) R ( kω) R R f R = k and C = pf are internal to the while R and R Q are external. A typical design procedure proceeds from the desired f O and Q as follows, using finite-tolerance fixed resistors. First find the ideal R value for the desired f O : khz R( Ideal)= k ( Ω) f O Then select a practical R value from the available finitetolerance resistors. se the actual R value to find the desired R Q, which also will be approximated with finite tolerance: RQ = Q ( kω) R The f O range is approximately khz to khz, limited mainly by the magnitudes of the external resistors required. As shown above, R varies with the inverse square of f O. This relationship desensitizes f O to R s Operational Filter and FilterCAD are trademarks of Linear Technology Corporation. Windows is a registered trademark of Microsoft Corporation. O fa 7

8 APPLICATIONS INFORMATION W tolerance (by a factor of incrementally), but it also implies that R has a wider range than f O. (R Q and R IN also tend to scale with R.) At high f O these resistors fall below k, heavily loading the outputs of the and leading to increased THD and other effects. At the other extreme, a lower f O limit of khz reflects an arbitrary upper resistor limit of MΩ. The s MOS input circuitry can accommodate higher resistor values than this, but junction leakage current from the input protection circuitry may cause DC errors. The nd order transfer functions H LP (s), H BP (s) and H HP (s) (below) are all inverting so that, for example, at DC the lowpass gain is H L. If two such sections are cascaded, these phase inversions cancel. Thus, the filter in the application schematic on the first page of this data sheet is a dual DC preserving, noninverting, rail-to-rail lowpass filter, approximating two straight wires with frequency selectivity. Figure 4 shows further details of nd order lowpass, bandpass and highpass responses. Configurations to obtain these responses appear in the next three sections. Basic Lowpass When Z IN of Figure 3 is a resistor of value R IN, a standard nd order lowpass transfer function results from to V (Figure ): () s HLω O = HLP() s = IN() s s + O / Q s ( ω ) + ωo V V The DC gain magnitude is H L = R/R IN. (Note that the transfer function includes a sign inversion.) Parameters ω O (= πf O ) and Q are set by R and R Q as above. For a nd order lowpass response the gain magnitude becomes QH L R IN R Q INV V nd ORDER V /4 R F V OT Figure. Basic Lowpass Configuration BANDPASS RESPONSE LOWPASS RESPONSE HIGHPASS RESPONSE GAIN (V/V) H B.77 H B GAIN (V/V) H P H L.77 H L GAIN (V/V) H P H H.77 H H f L f O f H f (LOG SCALE) f P f C f (LOG SCALE) f C f P f (LOG SCALE) fo Q = fo= flfh fh f ; L fl = fo + Q Q + fh = fo + Q Q + fc= fo Q + Q + fp = fo Q HP = H L Q 4Q fc = fo Q + Q + fp = f O Q HP = H H Q 4Q Figure 4. Characteristics of Standard nd Order Filter Responses fa

9 APPLICATIONS INFORMATION W at frequency f O, and for Q >.77, a gain peak occurs at a frequency below f O, as shown in Figure 4. Basic Bandpass There are two different ways to obtain a bandpass function in Figure 3, both of which give the following transfer function form: H BP () s HB( ωo/ Q) s = s + ( ωo/ Q) s+ ω O ω O = πf O and Q are set by R and R Q as described previously in Setting f O and Q. When Z IN is a resistor of value R IN, a bandpass response results at the V output (Figure a) with a gain parameter H B = R Q /R IN. Alternatively, a capacitor of value C IN gives a bandpass response at the V output (Figure b), with the same H BP (s) expression, and the gain parameter now H B = (R Q /kω)(c IN /pf). This transfer function has a gain magnitude of H B (its peak value) when the frequency equals f O and has a phase shift of at that frequency. Q measures the sharpness of the peak (the ratio of f O to 3dB bandwidth) in a nd order bandpass function, as illustrated in Figure 4. R IN R Q INV V nd ORDER V /4 R (a) Resistive Input V OT C IN R Q INV V nd ORDER V /4 (b) Capacitive Input Figure. Basic Bandpass Configurations R V OT F Basic Highpass When Z IN of Figure 3 is a capacitor of value C IN, a highpass response appears at the V output (Figure 7). Hs H + O / Q s ( ω ) + ωo V () s = HHP() s = VIN() s s Parameters ω O = πf O and Q are set by R and R Q as above. The highpass gain parameter is H H = C IN /pf. For a nd order highpass response the gain magnitude at frequency f O is QH H, and approaches H H at high frequencies (f >> f O ). For Q >.77, a gain peak occurs at a frequency above f O as shown in Figure 4. The transfer function includes a sign inversion. C IN R Q INV V nd ORDER V /4 R F7 V OT Figure 7. Basic Highpass Configuration Signal Swings The V and V outputs are capable of swinging to within roughly mv of each power supply rail. As with any analog filter, the signal swings in each nd order section must be scaled so that no output overloads (saturates), even if it is not used as a signal output. (Filter literature often calls this the dynamics issue.) When an unused output has a larger swing than the output of interest, the section s gain or input amplitude must be scaled down to avoid overdriving the unused output. The can still be used with high performance in such situations as long as this constraint is followed. For an section as in Figure 3, the magnitudes of the two outputs V and V, at a frequency ω = πf, have the ratio, V( jω) ( khz) V( jω ) = f regardless of the details of Z IN. Therefore, an input frequency above or below khz produces larger output amplitude at V or V, respectively. This relationship can guide the choice of filter design for maximum dynamic range in situations (such as bandpass responses) where there is more than one way to achieve the desired frequency response with an section. fa

10 APPLICATIONS INFORMATION Because nd order sections with Q have response peaks near f O, the gain ratio above implies some rules of thumb: f O < khz V tends to have the larger swing f O > khz V tends to have the larger swing. The following situations are convenient because the relative swing issue does not arise. The unused output s swing is naturally the smaller of the two in these cases: Lowpass response (resistor input, V output, Figure ) with f O < khz Bandpass response (capacitor input, V output, Figure b) with f O < khz Bandpass response (resistor input, V output, Figure a) with f O > khz Highpass response (capacitor input, V output, Figure 7) with f O > khz The -, a higher frequency derivative of the, has a design center f O of khz compared to khz in the. The rules summarized above apply to the - but with khz replacing the khz limits. Thus, an - lowpass filter section with f O below khz automatically satisfies the desirable condition of the unused output carrying the smaller signal swing. R IN k R Q.k INV V nd ORDER V /4 W R k C L 3pF V OT R L (EXTERNAL LOAD RESISTANCE) F Figure. khz, Q =.7 Lowpass Circuit for Distortion vs Loading Test Low Level or Wide Range Input Signals The contains a built-in capability for low noise amplification of low level signals. The Z IN impedance in each nd order section controls the block s gain. When set for unity passband gain, a nd order section can deliver an output signal more than db above the noise level. If low level inputs require further dynamic range, reducing the value of Z IN boosts the signal gain while reducing the input referred noise. This feature can increase the SNR for low level signals. Varying or switching Z IN is also an efficient way to effect automatic gain control (AGC). From a system viewpoint, this technique boosts the ratio of maximum signal to minimum noise, for a typical nd order lowpass response (Q =, f O = khz), to db. Input Voltages Beyond the Power Supplies Properly used, the can accommodate input voltage excursions well beyond its supply voltage. This requires care in design but can be useful, for example, when large out-of-band interference is to be removed from a smaller desired signal. The flexibility for different input voltages arises because the INV inputs are at virtual ground potential, like the inverting input of an op amp with negative feedback. The fundamentally responds to input current and the external voltage appears only across the external impedance Z IN in Figure 3. To accept beyond-the-supply input voltages, it is important to keep the powered on, not in shutdown mode, and to avoid saturating the V or V output of the nd order section that receives the input. If any of these conditions is violated, the INV input will depart from a virtual ground, leading to an overload condition whose recovery timing depends on circuit details. In the event that this overload drives the INV input beyond the supply voltages, the could be damaged. The most subtle part of preventing overload is to consider the possible input signals or spectra and take care that none of them can drive either V or V to the supply limits. Note that neither output can be allowed to saturate, even if it is not used as the signal output. If necessary the passband gain can be reduced (by increasing the impedance of Z IN in Figure 3) to reduce output swings. The final issue to be addressed with beyond-the-supply inputs is current and voltage limits. Current entering the virtual ground INV input flows eventually through the output circuitry that drives V and V. The input current magnitude ( / Z IN in Figure 3) should be limited by design to less than ma for good distortion performance. On the other hand, the input voltage appears across the fa

11 APPLICATIONS INFORMATION W external component Z IN, usually a resistor or capacitor. This component must of course be rated to sustain the magnitude of voltage imposed on it. Lowpass T Input Circuit The virtual ground INV input in the Operational Filter block provides a means for adding an extra lowpass pole to any resistor-input application (such as the basic lowpass, Figure, or bandpass, Figure a). The resistor that would otherwise form Z IN is split into two parts and a capacitor to ground added, forming an R-C-R T network (Figure ). This adds an extra, independent real pole at a frequency: fp = π RC P T where C T is the new external capacitor and R P is the parallel combination of the two input resistors R INA and R INB. This pair of resistors must normally have a prescribed series total value R IN to set the filter s gain as described above. The parallel value R P can however be set arbitrarily (to R IN /4 or less) which allows choosing a convenient standard capacitor value for C T and fine tuning the new pole with R P. A practical limitation of this technique is that the C T capacitor values that tend to be required (hundreds or thousands of pf) can destabilize the op amp in Figure 3 if R INB is too small, leading to AC errors such as Q enhancement. For this reason, when R INA and R IN B are unequal, preferably the larger of the two should be placed in the R INB position. Highpass T Input Circuit A method similar to the preceding technique adds an extra highpass pole to any capacitor-input application (such as the bandpass of Figure b or the highpass of Figure 7). This method splits the input capacitance C IN into two series parts C INA and C INB, with a resistor R T to ground between them (Figure ). This adds an extra st order highpass corner with a zero at DC and a pole at the frequency: fp = π RC T P where C P = C INA + C INB is the parallel combination of the two capacitors. At the same time, the total series capacitance C IN will control the filter s gain parameter (H H in Basic Highpass). For a given series value C IN, the parallel value C P can still be set arbitrarily (to 4C IN or greater). R INA R INB C INA C INB C T R Q R R T R Q R INV V nd ORDER V /4 F INV V nd ORDER V /4 F Figure. Lowpass T Input Circuit Figure. Highpass T Input Circuit The procedure therefore is to begin with the target extra pole frequency f P. Determine the series value R IN from the gain requirement. Select a capacitor value C T such that R P = /(πf P C T ) is no greater than R IN /4, and then choose R INA and R INB that will simultaneously have the parallel value R P and the series value R IN. Such R INA and R INB can be found directly from the expression: R R 4R R ± ( ) IN IN IN P The procedure then is to begin with the target corner (pole) frequency f P. Determine the series value C IN from the gain requirement (for example, C IN = H H (pf) for a highpass). Select a resistor value R T such that C P = /(πr T f P ) is at least 4C IN, and select C INA and C INB that will simultaneously have the parallel value C P and the series value C IN. Such C INA and C INB can be found directly from the expression: C C 4C C ± ( ) P P IN P fa

12 APPLICATIONS INFORMATION W This procedure can be iterated, adjusting the value of R T, to find convenient values for C INA and C INB since resistor values are generally available in finer increments than capacitor values. Different f O Measures Standard nd order filter algebra, as in Figure 4 and the various transfer-function expressions in this data sheet, uses a center frequency parameter f O (or ω O, which is πf O ). f O can also be measured in practical ways, including: The frequency where a bandpass response has phase shift The frequency where a bandpass response has peak gain The geometric mean of the 3.dB gain frequencies in a bandpass ( ƒ L ƒ H in Figure 4) An ideal mathematical nd order response yields exactly the same frequency by these three measures. However, real nd order filters with finite-bandwidth circuitry show small differences between the practical f O measures, which may be important in critical applications. The issue is chiefly of concern in high-q bandpass applications where, as the data below illustrate, the different f measurements tend to converge anyway for the. At low Q the bandpass peak is not sharply defined and the 3dB frequencies f L and f H are widely separated from this peak. The s f O is trimmed in production to give an accurate phase shift in the configuration of Figure a with resistor values setting f = khz and Q =. Table below shows typical differences between f O values measured via the bandpass criterion and f O values measured using the two other methods listed above (Figure a, R IN = R Q ). Table f O Q = Q = Q = Q = (BP ) BP-PEAK f O ƒ L ƒ H f O BP-PEAK f O ƒ L ƒ H f O khz +.3% +.3% +.% +.% khz +.% +.% +.% +.% 4kHz +.% +.% +.% +.% Demo Board The demo board is assembled with an or A in a -pin SSOP package and power supply decoupling capacitors. Jumpers on the board configure the for dual or single supply operation and power shutdown. Pads for surface mount resistors and capacitors are provided to build application-specific filters. Also provided are terminals for inputs, outputs and power supplies. fa

13 TYPICAL APPLICATIONS (Basic) Quad 3rd Order Butterworth Lowpass Filter, Gain = 3 R INA R IN3A R INB C IN V R IN3B R Q R R3 R Q3 V OT 3 INV B V B V B V + V SHDN V A V A INV A INV C V C V C AGND V D V D INV D V OT 3 R Q R R4 R Q4 V R INB R IN4B R INA C IN R IN4A f 3dB = khz C IN3 V OT3 V OT4 SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V C IN4 TAa k k FREQENCY (Hz) TAb M Quad 3rd Order Butterworth f 3dB f 3dB f 3dB f 3dB f 3dB f 3dB f 3dB Lowpass Filters khz 4kHz khz khz khz khz 4kHz C IN pf pf pf pf pf pf pf R INA 44.k 4.3k 3.k.43k.k.7k.k R INB k 7.k 4.3k 3.k.k.k 3.4k R Q 4k.k 7.4k.4k.k.k.k R 4k.k 7.4k.4k.k.k.k All four sections have identical R INA, R INB and C IN values. All resistor values are ±% fa 3

14 TYPICAL APPLICATIONS (Basic) Dual 4th Order Lowpass Filters R IN V R IN3 R Q R R3 R Q3 R IN 3 INV B V B V B INV C V C V C V + V SHDN V A V A INV A AGND V D V D INV D R IN4 3 R Q R R4 R Q4 TA3a SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V OT V V OT k k FREQENCY (Hz) BTTERWORTH f 3dB = khz TA3b M Quick Design Formulas for Some Popular Response Types: Butterworth (Maximally Flat Passband) for f C khz to 4kHz Chebyshev (Equiripple Passband) for f C khz to khz Bessel (Good Transient Response) for f C khz to 7kHz R, R3, R IN, R IN3 = k khz 4.4k khz 3.k khz R Q, R Q3 =.4k khz 7.k khz.k khz R, R4, R IN, R IN4 = k khz 7.7k khz 4.k khz R Q, R Q4 = 3.7k khz 7.3k khz 3.7k khz Notes: f C is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at f C. For Chebyshev filters with ±.db passband ripple up to. f C, use A grade. Example: Butterworth response, f C = khz. from the formulas above, R = R3 = R IN = R IN3 = k(khz/khz) = 4k. R Q = R Q3 =.4k(kHz/kHz) =.k. R = R4 = R IN = R IN4 = k(khz/khz) = 4k. R Q = R Q4 = 3.7k(kHz/kHz) =.4k. se nearest % values. TA3 TABLE 4 fa

15 TYPICAL APPLICATIONS th Order Lowpass Filters (Basic) R IN V R Q R R3 R Q3 3 R IN INV B INV C V B V C V B V C V + V SHDN V A V A INV A AGND V D V D INV D 3 R Q R V R4 R Q CHEBYSHEV f C = khz R IN4 R IN3 SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V OT TA4a k k k FREQENCY (Hz) TA4b Quick Design Formulas for Some Popular Response Types: Butterworth (Maximally Flat Passband) for f C khz to 4kHz R = R IN = k khz Chebyshev (Equiripple Passband) for f C khz to khz R = 7.k khz, R IN =.R* Bessel (Good Transient Response) for f C khz to 7kHz R = R IN =.k khz R Q =.k khz R Q =.3k khz khz + khz R Q = 3.3k khz R = R IN = k khz khz R = R ƒ IN = 4.k C R = R IN =.7k khz R Q = k khz R Q = 7.k khz khz + 44kHz R Q =.k khz R3 = R IN3 = k khz khz R3 = R ƒ IN3 = 7.k C R3 = R IN3 =.k khz R Q3 =.k khz R Q3 =.k khz khz + 3kHz R Q3 = 3.k khz R4 = R IN4 = k khz R4 =.7k khz, R IN4 = R4*. R4 = R IN4 = 3.4k khz R Q4 =.3k khz R Q4 =.7k khz R Q4 =.4k khz Notes: f C is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at f C. For Chebyshev filters with ±.db passband ripple up to. f C, use A grade. *The resistor values marked with an asterisk (*) in the Chebyshev formulas (R and R4) should be rounded to the nearest standard finite-tolerance value before computing the values dependent on them (R IN and R IN4 respectively). Example: Chebyshev response, f C = khz. The formulas above give R = 7.k, nearest standard % value 7.k. sing this % value gives R IN =.k, already a standard % value. R Q =.7k, nearest % value.k. R = R IN = 4.k, nearest % value k. R Q =.k, nearest % value k. R3 = R IN3 = 7.k, already a standard % value. R Q3 =.7k, nearest % value.7k. R4 =.7k, already a standard % value. This gives R IN4 =.4k, nearest % value.k. R Q4 =.7k, nearest % value.k. TA4 TABLE fa

16 TYPICAL APPLICATIONS (Basic) th Order Bandpass Filter, Single V Supply, Center Frequency 3dB Bandwidth = R IN C IN R Q INV B INV C V B V C R Q R 3 R V B V C V V + V µf SHDN AGND R3 3 R4 V A V D R Q3 V A INV A V D INV D R Q4 C IN3 R IN4 SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V OT TA7a f CENTER = khz FREQENCY (khz) TA7b Quick Design Formulas for Center Frequency f C (Recommended Range 4kHz to 4kHz): R = R3 =.k khz R = R4 =.7k khz R Q = R Q3 = 4.k khz R Q = R Q4 = 43.k khz khz + 3kHz khz + 4kHz C IN = C IN3 = pf k R Q R IN = R IN4 = RR Q C IN (k)(.pf) khz + khz Notes: R Q, R and C IN should be rounded to the nearest standard finite-tolerance value before using these values in the later formulas. Example: Center frequency f C of khz. The formulas give R = R3 =.k, nearest standard % value.k. R Q = R Q3 =.k, nearest % value.k. R = R4 =.k, nearest % value k. R Q = R Q4 = 47.k, nearest % value 47.k. C IN = C IN = 3.pF using.k for R Q, nearest standard % capacitor value 33pF. This and the % value R = k also go into the calculation for R IN = R IN4 =.k, nearest % value 4.k. TA7 TABLE fa

17 TYPICAL APPLICATIONS (Basic) th Order Bandpass Filter, Single V Supply, Center Frequency db Bandwidth = R IN R IN INV B INV C R Q V B V C R Q R 3 R V B V C V V + V µf SHDN AGND R3 3 R4 V A V D V A V D R Q3 INV A INV D R Q4 R IN4 R IN3 SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V OT TAa f CENTER = khz FREQENCY (khz) TAb Quick Design Formulas for a Center Frequency f C (Recommended Range khz to khz): R = R3 =.7k khz + 73kHz R ƒ IN = R IN3 = C khz R. R Q = R Q3 =.k khz khz + khz R = R4 =.k khz R IN = R IN4 = + 34kHz khz R Q 4.3 R Q = R Q4 =.k khz khz + 3kHz Notes: R and R Q should be rounded to the nearest standard finite-tolerance value before using these values in the later formulas. For f C < khz, the maximum peak-to-peak passband input level is (f C /khz)v. se A for minimum variation of passband gain. Example: Center frequency f C of khz. The formulas give R = R3 =.7k, nearest standard % value.k. This value gives R IN = R IN3 =.4k, nearest % value.k. R Q = R Q3 =.k, nearest % value 4.k. R = R4 =.k, already a standard % value. This gives R IN = R IN4 = 3.4k (again already a standard % value). R Q = R Q4 = 3.4k, nearest % value 3.4k. If A is used, resistor tolerances tighter than % will further improve the passband gain accuracy. TA TABLE fa 7

18 TYPICAL APPLICATIONS (Basic) th Order Bandpass (High Frequency) Filter 3dB Bandwidth = Center Frequency, Gain = R IN V + R Q R R3 R Q3 R IN 3 INV B V B V B INV C V C V C V + V SHDN V A V A INV A AGND V D V D INV D R IN4 R IN3 3 R Q R R4 R Q4 SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V V OT TAa 3 f CENTER = khz FREQENCY (khz) TAb th Order Bandpass Filter 3dB BW = f CENTER f, Gain = CENTER f CENTER f CENTER f CENTER f CENTER f CENTER f CENTER khz khz khz khz khz 3kHz 4kHz Side B R IN 4.4k.3k.34k.k.k.4k.k R Q 4.4k.3k 4.k 3.3k 34.k 3.4k 3.k R.4k.4k.k.k.k.k.k Sides A, C, D R IN, R IN3, R IN4 4.4k.3k 4.k 3.3k 34.k 3.4k 3.k R Q, R Q3, R Q4 4.4k.3k 4.k 3.3k 34.k 3.4k 3.k R, R3, R4.4k.4k.k.k.k.k.k All resistor values are ±% fa

19 TYPICAL APPLICATIONS C IN pf (Basic) th Order Wideband Bandpass Filter f CENTER = khz, 3dB BW 4kHz to khz R IN.k INV B INV C R Q k R Q 4.7k V B V C R.k 3 R 34.k V B V C V + V + V V µf SHDN AGND 3 V A V D R3 3.4k R4.7k V A V D R Q3.k R Q4 k INV A INV D C IN3 7pF C IN4 47pF V OT TAa 3 4 FREQENCY (khz) TAb SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V th Order Highpass.dB Ripple Chebyshev Filter f CTOFF = 3kHz C IN C IN pf V C IN3 pf R Q,.k INV B INV C V B V C R, 3.7k 3 V B V C V + V R3, 7k R Q3, 4.k SHDN V A V A INV A AGND V D V D INV D 3 R Q,.k R,.k R4, 7k R Q4,.k SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V C IN pf V C IN4 pf V OT TAa k k k M FREQENCY (Hz) TAb TOTAL OTPT NOISE = 4µV RMS nd Order 3kHz Highpass Cascaded with th Order 3kHz Lowpass C IN pf R IN,.3k INV B INV C R Q, 3.k R Q,.k V B V C R, k 3 R,.3k V B V C V V + V V 3 SHDN AGND 4 3 V A V D R3,.3k R4,.3k V A V D R Q3, 4k R Q4, 3.74k INV A INV D 7 R IN3,.k R IN4, 3.4k V OT 4 TAa FREQENCY (khz) SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. TAb PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V ALL RESISTORS = % METAL FILM fa

20 APPLICATIONS INFORMATION Notches and Elliptic Responses W The basic (essentially all-pole) circuit techniques described so far will serve many applications. However, the sharpest-cutoff lowpass, highpass and bandpass filters include notches (imaginary zero pairs) in the stopbands. A notch, or band-reject, filter has zero gain at a frequency f N. Notches are also occasionally used by themselves to reject a narrow band of frequencies. A number of circuit methods will give notch responses from an Operational Filter block. Each method exhibits an inputoutput transfer function that is a standard nd order bandreject response: H BR () s = s H s N( + ω N) + ( ω / Q) s+ ω O O with parameters ω N = πf N and H N set by component values as described below. (ω = πf and Q are set for the Operational Filter block by its R and R Q resistors as described earlier in Setting f and Q). Characteristically, the gain magnitude H BR (jπf) has the value H N (f N /f ) at DC (f = ) and H N at high frequencies (f >> f N ), so in addition to the notch, the gain changes by a factor: High Frequency Gain DC Gain = ƒ ƒ N The common principle in the following circuit methods is to add a signal to a filtered replica of itself having equal gain and phase difference at the desired notch frequency O f N. The two signals then cancel out at frequency f N. The notch depth (the completeness of cancellation) will be infinite to the extent that the two paths have matching gains. Three practical circuit methods are presented here, with different features and advantages. Examples and design procedures for practical filters using these techniques appear in a series of articles attached to this data sheet on the Linear Technology web site ( Also available free is the analog filter design software, FilterCAD for Windows, recommended for designing filters not shown in the Typical Applications schematics in this data sheet. Elementary Feedforward Notches A textbook method to get a phase difference at frequency f N for a notch is to dedicate a bandpass nd order section (described earlier under Basic Bandpass), which gives phase shift at the section s center frequency f O (Figure, with C IN = ), so that f N = f O. The bandpass section of Figure a, at its center frequency f O, has a phase shift of and a gain magnitude of H B = R Q /R IN. A notch results in Figure if the paths summed into virtual ground have the same gains at the frequency (then I O = ). This requires a constraint on the resistor values: RIN RQ = R R FF IN C IN R IN R Q R R IN I O R GAIN INV V nd ORDER V /4 VIRTAL GROND V OT R FF + F Figure. Feedforward Notch Configuration for f N f O fa

21 APPLICATIONS INFORMATION W Note that the depth of the notch depends on the accuracy of this resistor ratioing. The virtual-ground summing point in Figure may be from an op amp as shown, or in a practical cascaded filter, the INV input of another Operational Filter block. The transfer function in Figure with C IN = is a pure notch (f N = f ) of the H BR (s) form above, and the parameters are: ƒ H N N = ƒ R = O GAIN RFF Because f N = f in this case, the gain magnitude both at DC and at high frequencies (f >> f N ) is the same, H N (assuming that the op amp in Figure adds no significant frequency response). Figure shows this. Such a notch is inefficient as a cascaded part of a highpass, lowpass or bandpass filter (the most common uses for notches). Variations of Figure can add a highpass or lowpass shape to the notch, without using more Operational Filter blocks. The key to doing so is to decouple the notch frequency f N from the center frequency f of the Operational Filter block (this is shown in Figures 3 and ). The next two sections summarize two variations of Figure with this highpass/ lowpass shaping, and the remaining section shows a different approach to building notches. Feedforward Notches for f N > f When C IN in Figure, the notch frequency f N is above the center frequency f and the response has a lowpass shape as well as a notch (Figure 3). C IN contributes phase lead, which increases the notch frequency above the center frequency of the nd order Operational Filter block. The resistor constraint from the previous section also applies here and the H BR (s) parameters become: ƒ H N N = ƒ O R C IN IN RQC O N = R GAIN ƒ R FF ƒ C is the internal capacitor value in the Operational Filter block (in the, pf). The configuration of Figure is most useful for a stopband notch in a lowpass filter or as an upper stopband notch in a bandpass filter, since the two resistors R IN and R FF can replace the input resistor R IN of either a lowpass section (Figure ) or a resistor-input bandpass section (Figure a) built from a second Operational Filter block. 4 f N f O DC GAIN = H N ( ) HIGH FREQ GAIN = H N f N = f O = khz H N = Q = FREQENCY (khz) AN4 TA 4 f O = khz f N = khz Q = DC GAIN = db FREQENCY (khz) F3 Figure. Notch Response with f N = f O Figure 3. Notch Response with f N > f O fa

22 APPLICATIONS INFORMATION W The configuration is robust against tolerances in the C IN value when f N approaches f (for f N /f.4, as a rule of thumb) which is attractive in narrow transition-band filters, because of the relative cost of high accuracy capacitors. Further application details appear in Part of the series of articles. Feedforward Notches for f N < f Just as feedforward around an inverting bandpass section yields a notch at the section s f (Figure with C IN = ), feedforward around an inverting lowpass section causes a notch at zero frequency (which is to say, a highpass response). Moreover, and this is what makes it useful, introducing a capacitor for phase lead moves the notch frequency up from DC, exactly as C IN in Figure moves the notch frequency up from the center frequency f. In Figure 4, the inverting lowpass output (V) of the Operational Filter block is summed, at a virtual ground, with a fed-forward input signal. Capacitor C IN shifts the resulting notch frequency, f N, up from zero, giving a low frequency notch with a highpass shape (Figure ). The H BR (s) response parameters are now: ƒ H N N = ƒ = O R R GAIN FF R C R R C R Q IN IN The constraint required for exact cancellation of the two paths (i.e., for infinite notch depth) becomes: R R IN FF RQ C = RC IN R and C are the internal precision components (in the, k and pf respectively) as described above in Setting f and Q. The configuration of Figure 4 is most useful as a lower stopband notch in a bandpass filter, because the resistors R IN and R FF can replace the input resistor R IN of a bandpass section made from a second Operational Filter block, as in Figure a. The configuration is robust against tolerances in the C IN value when f N approaches f (for f / f N.4, as a rule of thumb) which is attractive in narrow transition-band filters, because of the relative cost of high accuracy capacitors. Further application details appear in Part of the series of articles. 4 k f N f O DC GAIN = H N ( ) f O = khz f N = khz Q = HIGH FREQ GAIN = db k FREQENCY (Hz) HIGH FREQ GAIN = H N F M Figure. Notch Response with f N < f C IN R IN R Q R R IN I O R GAIN INV V nd ORDER V /4 VIRTAL GROND V OT R FF + F4 Figure 4. Feedforward Notch Configuration for f N < f O fa

23 APPLICATIONS INFORMATION R-C niversal Notches A different way to get phase shift for a notch is to use the built-in phase difference between the two Operational Filter block outputs along with a further from an external capacitor. This method achieves deep notches independent of component matching, unlike the previous techniques, and it is convenient for cascaded highpass as well as lowpass and bandpass filters. The V output of an Operational Filter block is a timeintegrated version of V (see Figure 3), and therefore lags V by over a wide range of frequencies. In Figure, a notch response occurs when a nd order section drives a virtual-ground input through two paths, one through a capacitor and one through a resistor. Again, the virtual ground may come from an op amp as shown, or from another Operational Filter block s INV input. Capacitor C N adds a further to the difference between V and V, producing a wideband phase difference, but frequency-dependent amplitude ratio, between currents I R and I C. At the frequency where I R and I C have equal magnitude, I O becomes zero and a notch occurs. This gives a net transfer function from to V OT in the form of H BR (s) as above, with parameters: ƒ N = π RCRC H N N N = RGAIN CN R C IN W RGAIN R DC Gain = R R IN ƒo High Frequency Gain RC N N ƒ = DC Gain = RC N N R and C are the internal precision components (in the, k and pf respectively) as described above in Setting f and Q. nlike the notch methods of Figures and 4, notch depth from Figure is inherent, not derived from component matching. Errors in the R N or C N values alter the notch frequency, f N, rather than the degree of cancellation at f N. Also, the notch frequency, f N, is independent of the section s center frequency f, so f N can freely be equal to, higher than or lower than f (Figures, 3 or, respectively) without changing the configuration. The chief drawback of Figure compared to the previous methods is a very practical one the C N capacitor value directly scales H N (and therefore the high frequency gain). Capacitor values are generally not available in increments or tolerances as fine as those of resistors, and this configuration lacks the property of the previous two configurations that sensitivity to the capacitor value falls as f N approaches f. nlike the previous notch circuits, this one is also noninverting at DC. R IN R Q R R N I R I O R GAIN INV V nd ORDER V /4 C N I C VIRTAL GROND + V OT F Figure. The R-C niversal Notch Configuration for an Operational Filter Block fa 3

24 TYPICAL APPLICATIONS th Order khz Lowpass Elliptic Filter with db Stopband Attenuation (Advanced) C IN 4pF R IN 37.4k V R IN 4.7k R Q 3.k R 3.k R3 3.k R Q3 34k R IN3 3.k 3 INVB VB VB V + SHDN VA VA INVA INVC VC VC V AGND VD VD INVD 3 R Q 3k R 7.k R4 3.4k R Q4.k R IN4 3.4k V V OT 4 FREQENCY (khz) TAb C IN3 pf C IN4 pf SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V TAa SES THREE R-C NIVERSAL NOTCHES AT f N = 33kHz, 7kHz, khz. DETAILED DESCRIPTION IN LINEAR TECHNOLOGY DESIGN NOTE. WIDEBAND OTPT NOISE µv RMS th Order khz Elliptic Bandpass Filter R FF 3k V R IN 3.k R IN.3k INVB INVC IN R Q.k C IN R Q 4.k.pF VB VC R.7k 3 R k VB VC V V + V SHDN AGND R3 k VA VD 3 R Q3 7.k R4.3k VA VD R IN3 4k INVA INVD R Q4.k C IN3 pf R IN4.3k R FF4 33k V V OT FREQENCY (khz) TA3b SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V F3a 4 fa

25 TYPICAL APPLICATIONS (Advanced) th Order khz Lowpass Elliptic Filter R IN 4k V + R INA 4k R INB.k C IN 3pF R Q.3k R 34k V R3 k R Q3 3k 3 C IN 33pF INVB INVC VB VC VB VC V + V SHDN AGND VA VD VA VD INVA INVD 3 R Q k R k R4 4k R Q4.k R IN4 3k C IN3 7pF R IN3 3k TO PIN V C IN4 pf V OT F4a SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V FREQENCY (khz) TA4b NOISE + THD (db) Noise + THD vs Frequency =.V RMS = 4.V P-P V S = ±V FREQENCY (khz) TA4c fa

26 TYPICAL APPLICATIONS (Advanced) Dual th Order Lowpass Elliptic Filter R IN C IN R INA R INB INVB INVC V OT C R Q IN VB VC R Q R 3 R VB VC V V + V V SHDN AGND R 3 R VA VD R Q VA VD R Q R INA R INB INVA INVD C IN C IN V OT R IN TAa f C = khz 4 FREQENCY (khz) TAb SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V f C (Hz) R INA R INB C IN R Q R R IN C IN R Q R k.k 7.k pf k 7.k.34k pf.3k.3k 7k.k.4k pf 3.k 3.3k.3k pf.7k k k.k 3.7k 3pF.k 3.k.k pf.7k 44.k Construction and Instrumentation Cautions db rejections at hundreds of kilohertz require electrically clean, compact construction, with good grounding and supply decoupling, and minimal parasitic capacitances in critical paths (such as Operational Filter block INV inputs). In a circuit with k resistances trying for db rejection at khz, a stray coupling of.3pf around the signal path can preclude the db. (By comparison, the stray capacitance between two adjacent pins of an IC can be pf or more.) Also, high quality supply bypass capacitors of near the chip provide good decoupling from a clean, low inductance power source. But several inches of wire (i.e., a few microhenrys of inductance) from the power supplies, unless decoupled by substantial capacitance ( µf) near the chip, can cause a high-q LC resonance in the hundreds of khz in the chip s supplies or ground reference, impairing stopband rejection and other specifications at those frequencies. In demanding filter circuits we have often found that a compact, carefully laid out printed circuit board with good ground plane makes a difference of db in both stopband rejection and distortion performance. Highly selective circuits can even exhibit these issues at frequencies well below khz. Finally, equipment to measure filter performance can itself introduce distortion or noise floors; checking for these limits with a wire replacing the filter is a prudent routine procedure. fa

27 PACKAGE DESCRIPTION G Package -Lead Plastic SSOP (.3mm) (Reference LTC DWG # --4) * (.7.) (.3.3)..3** (..) (..7).3. (..).. (..37) NOTE:. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE * DIMENSIONS DO NOT INCLDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED.mm (.") PER SIDE ** DIMENSIONS DO NOT INCLDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED.4mm (.") PER SIDE. (.) BSC..3 (..).. (..) G SSOP N Package -Lead PDIP (Narrow.3 Inch) (Reference LTC DWG # --).77* (.) MAX 4 3. ±.* (.477 ±.3) (7..).3 ±. (3.3 ±.7).4. (.43.).. (..3) ( ). (.) MIN. (3.7) MIN *THESE DIMENSIONS DO NOT INCLDE MOLD FLASH OR PROTRSIONS. MOLD FLASH OR PROTRSIONS SHALL NOT EXCEED. INCH (.4mm). (.4) BSC Information furnished by Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.. (.) TYP. ±.3 (.47 ±.7) N fa 7

28 TYPICAL APPLICATION Dual 4th Order db Gaussian Lowpass Filter R IN R IN INV B INV C R Q V B V C R Q R 3 R V B V C V V + V SHDN V A AGND V D 3 µf R3 R4 V A V D R IN3 R Q3 R Q4 INV A INV D V OT V OT 4 f C = 4kHz f C = 3kHz f C = khz 3 FREQENCY (khz) TAb 4-Level Eye Diagram f C = khz, Data Clock = 3kHz R IN4 TAa SCHEMATIC INCLDES PIN NMBERS FOR -PIN PACKAGE. PINS 4, 7, 4, 7 (NOT SHOWN) ALSO CONNECT TO V V/DIV µs/div TAc f C (Hz) R IN = R IN3 R = R3 R Q = R Q3 R IN = R IN4 R = R4 R Q = R Q4 k k k 34k 34k 34k 34k 3k.k.k.k 4.k 4.k.k 4k.4k.4k.4k.k k.4k RELATED PARTS PART NMBER DESCRIPTION COMMENTS LTC, LTC-X Quad -Pole Switched Capacitor Building Block Family Clock-Tuned LTC- -Pole Elliptic Lowpass, f C = MHz/.MHz No External Components, SO - Quad -Pole Active RC, khz to 3kHz Same Pinout as the LTC3-/LTC3-3 4th Order Active RC Lowpass Filters f CTOFF(MAX) = khz, Resistor Programmable LTC4 khz to khz Digitally Controlled Filter and PGA Continuous Time Low Noise th Order with PGA LTC-3 khz Continuous Time, Linear Phase Lowpass Filter 7th Order, Differential Inputs and Outputs LTC-.3MHz Continuous Time Lowpass Filter 7th Order, Differential Inputs and Outputs fa LT/TP REV A.K PRINTED IN SA Linear Technology Corporation 3 McCarthy Blvd., Milpitas, CA (4) 43- FAX: (4) LINEAR TECHNOLOGY CORPORATION