Sheng Sun and Lei Zhu. Digital Object Identifier /MMM April 2009

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1 COREL Sheng Sun and Lei Zhu This article describes a class of recently developed multiple-mode-resonator-based bandpass filters for ultra-wide-band (UWB) transmission systems. These filters have many attractive features, including a simple design, compact size, low loss and good linearity in the UWB, enhanced out-of-band rejection, and easy integration with other circuits/antennas. In this article, we present a variety of multiple-mode resonators with stepped-impedance or stub-loaded nonuniform configurations and analyze their properties based on the transmission line theory. Along with the frequency dispersion of parallel-coupled transmission lines, we design and implement various filter structures on planar, uniplanar, and hybrid transmission line geometries. Introduction UWB microwave filters have recently been receiving enormous attention in both academia and industry for applications in wireless transmission systems. Since the Federal Communications Commission (FCC) in the United States approved the unlicensed use of the UWB spectrum in 22 [], researchers in microwave circles have been attracted to the development of UWB bandpass filters covering the FCC-specified emission mask. A general block diagram of a UWB transmitter is depicted in Figure, where the source data are encoded, modulated, and multiplexed at the chip level, and then the multiplexed pulse is transmitted by a UWB antenna after reshaping and amplifying at the package level. As in traditional transmission systems, filter blocks are used to remove unwanted signals and noise from Sheng Sun and Lei Zhu are with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore ( ezhul@ntu.edu.sg). Digital Object Identifier.9/MMM /9/$ IEEE April 29 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

2 UWB transmission systems. However, unlike the traditional narrow-band wireless transmission systems discussed in [2], UWB systems spread the desired signals across a very wide frequency range and broadcast them on separate bands. For indoor and handheld UWB systems, devices must operate with a -db bandwidth in the frequency range of 3..6 GHz. Figure 2 portrays the relevant FCC-specified emission limits of effective isotropic radiated power (EIRP) in dbm/mhz []. These specifications create a tremendous challenge to a filter designer with respect to the design of a fractional bandwidth of about 9.5% at the center frequency of 6.85 GHz. As is well known, existing filter theory was established under the assumption of a narrow passband, and it has been found to be very powerful in the design of microwave filters with various narrow-band filtering performance [3] [5]. A wide-band filter can reasonably be constructed by cascading a few transmission line resonators through enhanced coupling strength. Nevertheless, it has been theoretically difficult to design filters with an ultrawide passband bandwidth. A simple approach was proposed to realize an initial UWB filter by creating two transmission zeros below 3. GHz and above.6 GHz [6]. However, this filter, using multiple ring resonators, has narrow lower and upper stopbands and a large size due to the nature of bandstop behavior. Another solution was to directly cascade a low-pass/bandstop filter with a highpass filter [7] []. Although these filters have a very wide upper-stopband performance, the insertion loss and overall circuit size are inevitably increased. On the other hand, the conventional high-pass prototype with short-circuited shunt stubs has been adopted to explore various quasi-bandpass filters with preferred UWB width [9], [] 3]. This technique has recently been employed to design a UWB filter on a multilayer liquid-crystal-polymer substrate [4]. Nevertheless, via holes are always an issue in the practical implementation of planar filters with short-circuited stubs. In parallel development, the concept of a multiplemode resonator (MMR) with a stepped-impedance configuration was originated in [5] and was extended to the implementation of a class of UWB bandpass filters in [6]. Unlike traditional parallel-coupled resonator filters using a single resonant mode, MMR-based filters were constructed by allocating the first few resonant frequencies into the desired wide passband while setting up the coupling peak of quarter-wave parallelcoupled lines at the center frequency. In this article, we investigate the fundamental properties of MMRs based on simple transmission line theory, and we then present their varied geometries on microstrip lines, coplanar waveguides, and hybrid structures. Next, we review a class of recently developed MMR-based UWB filters with their resultant physical configurations and electrical performance. Finally, we demonstrate UWB By varying the length of the center low-impedance line section or increasing the number of nonuniform sections in the MMR, UWB bandpass filters with more in-band transmission poles have been presented and implemented. filters with good in-band transmission, enhanced out-of-band rejection, and excited band-notch both in theory and experiment. Building Blocks for MMR UWB Filters Multiple-Mode Resonator (MMR) Figure 3 depicts the geometry of an open-circuited MMR using a microstrip topology. It consists of a lowimpedance line section in the center and two identical high-impedance line sections on the two sides. The characteristic impedances of each section are defined as Z and Z 2, and u and u 2 are their electrical lengths. The physical topology of an MMR has the same configuration as the stepped-impedance resonator (SIR) of [7]. However, the objective of the proposed SIR in [7] was to enlarge the frequency spacing between the first- and second-order resonant modes so as to design a narrow-band filter with a widened upper Emission Limits EIRP in dbm Source Data Modulator Pulse Generator UWB Filter On-Chip RF Amplifier Figure. A general block diagram of UWB transmitter systems. UWB Band: 3.~.6 GHz On-Package Transmit Antenna Indoor UWB Systems Hand-Held UWB Systems Figure 2. Emission limits of EIRP for indoor and hand-held UWB systems. April Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

3 In the design of UWB bandpass filters using multiple-mode resonators, the MMR needs to be constructed to allocate the first three resonant frequencies in the UWB passband of interest almost equally. This has never been done before. Based on generalized transmission line theory, the multimode property of this MMR can be characterized in a straightforward way. Figure 3 is its equivalent transmission line model, where the two ends of this MMR are both opencircuited. The input admittance (Y in ) at the left end, looking into the right side, can be derived in the following way: Z 2 Z Z 2 Y in 5 jy 2 2Rtanu tanu 2 2R 2 tanu tanu 2 2 R 2 tan 2 u 2 2 tan 2 u R 2 2tanu tanu 2, () Y in θ 2 Z 2, θ 2 2θ θ 2 Z, 2θ Z 2, θ 2 where R = Z 2 /Z is the impedance ratio of high- and low-impedance lines. At the resonances, we have Y in 5. (2) From (2), a set of resonant frequencies (f, f 2, ) can be determined from u and u 2. In the case of u 2 = 2u = u, we have Figure 3. Geometry of the open-circuited steppedimpedance multiple-mode resonator. Equivalent transmission line model. stopband [8]. This technique was also recently employed to make use of the SIR s first two and three resonant modes in the design of dual- and triple-band filters [9] [2]. In our proposed MMR-based filters, the first several resonant modes are utilized with frequency-dispersive coupled lines, aiming to form a single wide passband. f 2 /f,f 3 /f f 2 /f,f 3 /f,f 4 /f f 2 /f 2 Z 2 <Z Z 2 >Z R = Z 2 /Z f 2 /f f 3 /f f 3 /f θ 2 = 2θ f 4 /f θ 2 = θ Z 2 <Z Z 2 >Z R = Z 2 /Z Figure 4. Normalized resonant frequencies versus impedance ratio (R = Z 2 /Z ) of a multiple-mode resonator. Two-mode resonator with u 2 = 2u. Three-mode resonator with u 2 = u. Therefore, R u f 2 5 tan 2 Å R 2, R 2 u f tan 2 Å R, u f p 2. (3) R 2 f 2 5 u f 2 2 tan 2 f u f 2 5 Å R, R tan 2 Å R 2 f 3 5 u f 3 2 f u f 2 5 p. (4) R 2tan 2 Å R 2 Figure 4 depicts the two normalized frequencies, f 2 /f and f 3 /f, versus R. As R decreases from unity ( = ), both second- and third-order resonances depart from the first resonance, thus widening the upper stopband of the single-mode SIR bandpass filter when R < [7]. On the other hand, as R increases from unity, the first two higher-order resonances both move down in frequency towards the first resonance. This property of an SIR filter with R > was previously used to realize a low-loss diplexer on a very high dielectric substrate [2]. In [5], another novel application of this SIR filter technology with R > was discussed, where the first two resonant modes [the red line in Figure 4] were utilized to design wide-band bandpass filters. Using this initial single resonator, four transmission poles were achieved in the core passband [5]. Later on, the 9 April 29 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

4 first three resonant modes were used together to form a wide passband [22]. In the analysis for the case of u 2 = u = u, we can obtain from () and (2) Thus, u f 2 5 tan 2 "R, u f p 2, u f 3 2 5p2tan 2 "R, u f 4 2 5p. (5) f 2 p 5 f 2tan 2 "R, f 3 p 5 f tan 2 "R 2, f 4 p 5 f tan 2 "R. (6) These three normalized frequencies, f 2 /f, f 3 / f, and f 4 /f, are plotted in Figure 4. Similar to those in the case of u 2 = 2u, the first three resonant frequencies become very close to one another as R increases. Meanwhile, the fourth one is still distinct from them, and it forms the first spurious harmonic outside the constituted wide passband. Compared with the former case in Figure 4, one additional resonant mode is moved down to the formed passband. Figure 5 shows the simulated values of S 2 of an MMR circuit that is driven through capacitive weak coupling at the two ends under a fixed u 2 = p/2 at the center frequency 6.85 GHz. As can be seen, the first few resonant frequencies are equally allocated within the FCC-specified UWB limits. One can easily imagine the formation of the desired ultrawide passband, as long as the coupling strength of the two driven lines is properly increased. Parallel-Coupled Microstrip Line (PCML) The parallel-coupled microstrip line (PCML) has been widely used in the design of multistage bandpass filters [3]. To realize a wideband filter, its strip and slot widths have to be reduced to achieve a tight coupling, and more resonators are usually required to increase the number of transmission poles. This procedure may degrade the filtering performance with a lower Q-factor and a higher insertion loss, however. To circumvent these problems, an aperture is formed on the ground plane underneath the coupled strip conductors, as shown in Figure 6, to offer an alternative PCML with enhanced S 2 (db) coupling strength. Figure 6 shows that a single transmission pole can be split to two poles as the aperture width is raised from 2.2 mm to 3. mm [23]. By combining this aperture-backed PCML and the multiple-mode resonators discussed above, a class of compact and wideband filters can be realized, as reported in [23]. An initial design example of a multiple-mode resonator filter is presented in [5] and [23]. Figure 7 shows the geometry of the wide-band S and S 2 (db) The parallel-coupled microstrip line has been widely used in the design of multistage bandpass filters. 3 Ideal Case (Indoor) 4 Ideal Case (Hand-Held) θ 2 = θ θ 5 2 = 2θ ε r h 2 UWB Band: 3.~.6 GHz Figure 5. Desired FCC limits and simulated limits, S 2, of multiple-mode resonators under weak coupling. S 2 S Coupled-Line Backside Aperture Ground W = 2.2 mm W = 3. mm Unit: mm W Figure 6. Geometry and simulated frequency responses of a parallel-coupled microstrip line (PCML) with a backside aperture. Geometry [23]. S and S 2 of PCML with different aperture widths [23]. April 29 9 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

5 Wf W S +W f ε r =.2 h =.635 mm S and S 2 (db) D L D W =.4 mm D = 5. mm L = 3.36 mm S =.88 mm S 2 (db) Reference erenc Planes ε r =.8 h =.27 mm Group Delay (ns) (c) S (db) Figure 7. Geometry and frequency responses of a multiplemode resonator filter. Geometry [23]. and measured responses ( S and S 2 ) [23]. Figure 9. Photograph and results of the designed microstrip-line UWB bandpass filter [6]. Photograph. S and S 2. (c) Group delay. bandpass filter where the coupled strip conductors in the two sides of a single multiple-mode resonator are backed by apertures. In this case, a two-mode resonator with u 2 = 2u is employed, and a pair of open-circuited stubs is installed in the shunt at the center. By taking advantage of loaded stubs, the harmonic frequency ( f 3 ) that appears above twice the first resonant frequency ( f ) can be suppressed. Figure 7 describes the simulated and measured frequency response of the designed four-pole bandpass filters with a wide fractional bandwidth of about 6%. Both the simulated and measured insertion losses, S 2, are below.6 db within the passband. Meanwhile, the simulated return loss, S, is lower than 2 db, compared with the measured return loss, which is lower than 6 db. The total length of this stub-loaded bandpass filter is about.7l g at 6. GHz, compared to.25l g for a conventional four-pole parallel-coupled microstripline bandpass filter, as discussed in [3]. Lc = Unit: mm Strip Width: W =. Slot Width: S = Ω Line 5 Ω Line Figure 8. Schematic of the MMR-based microstrip-line UWB bandpass filter [6]. Design Examples: MMR-Based UWB Filters Integration of MMR and PCML In this section, the above technique based on the integ ration of MMR and PCML will be extended to make an UWB bandpass filter with a fractional bandwidth of about 9.5%. This initial UWB filter is investigated to demonstrate that the integration of MMRs and PCMLs can result in such a UWB passband. In the design, an MMR with u 2 = u is properly modified so as to reallocate its first three resonant modes close to the lower end, center, and upper end of the specified UWB passband, following the mode graph in Figure 5. By increasing the coupling strength of the two PCMLs, a good UWB passband with five transmission poles can be realized. The first MMR-based UWB bandpass filter was designed and implemented in [6] on Rogers RT/Duroid 6, with a dielectric substrate of e r =.8 and a thickness h =.27 mm. Figure 8 shows its basic schematic. At the center frequency of the UWB passband, 6.85 GHz, this MMR consists of a nonuniform transmission line resonator, as shown in Figure 3. One half-wavelength (l g /2) low-impedance line section is in the center, and two identical l g /4 highimpedance line sections are at the sides. Figure 9 is a photograph of the fabricated UWB bandpass filter. The length between the two reference planes is about 6 mm. The simulated and measured frequency responses are plotted in Figure 9 and (c) 92 April 29 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

6 for quantitative comparison. In the frequency band from. GHz to 3. GHz, the measured results match well with those from the simulation, and they achieve a fractional bandwidth of approximately 3%. Over the UWB passband, the return loss values in the simulation and measurement results are both higher than db with the appearance of five transmission poles. It can be understood from [5] and [23] that three poles are brought out by the first three resonant modes of the MMR and two poles are contributed by the two quarter-wavelength PCMLs. Within the passband, the simulated and measured group delays are both less than.43 ns with a maximum variation of.23 ns, thus implying good linearity in this UWB bandpass filter. In general, this MMR-based UWB bandpass filter can be realized by utilizing other resonant modes, such as the first two, three, or four resonant modes of a constituted MMR. By varying the length of the center low-impedance line section or increasing the number of nonuniform sections in the MMR, UWB bandpass filters with more in-band transmission poles have been presented and implemented in [24] [26]. UWB Bandpass Filters with Varied Geometries Aperture-Backed PCML As discussed above, an aperture-backed PCML is formed by the partial removal of the ground plane underneath the coupled strips. It is used here as a coupling-enhanced PCML in the design of an alternative MMR-based UWB filter with relaxed fabrication tolerance. Following the discussion in the previous section, an aperture-backed PCML is formed with its maximum coupling strength near the UWB emission mask s center frequency of 6.85 GHz. By driving a three-mode MMR with these PCMLs at its two sides, a compact UWB bandpass filter with five transmission poles was developed in [27]. The physical configuration of this filter is shown in Figure. Compared with the parallel-coupled line sections in Figure 8, the strip and slot widths of the coupled-line section are increased from. mm and.5 mm to.2 mm and. mm, respectively, when a backside aperture is installed. Thus, this approach can significantly relax the requirement in fabrication tolerance. Figure shows the simulated and measured S-parameter magnitudes of the designed UWB bandpass filter. The two sets of results compare reasonably well with each other except that the lower cutoff frequency in the measurement is slightly shifted down to 2.9 GHz. Due to the very small interaction between the fields in the aperture and strip conductor, we find that the radiation-related loss is very small in this UWB filter. Similar Unit: mm Figure. Schematic of the designed UWB bandpass filter on an aperture-backed microstrip line [27]. to the filter in Figure 8, the overall length of this aperture-compensated UWB filter is only about one wavelength at 6.85 GHz, which is much smaller than those reported in [6] []. Broadside Microstrip/CPW It is well known that surface-to-surface or broadside-coupled structures can also be utilized to enhance coupling strength by employing both the top and bottom surfaces of the printed circuit board. In [28], an UWB bandpass filter is accordingly constructed using a hybrid microstrip/cpw structure with the back-to-back transition configuration, as shown in Figure 2. Here, an MMR-based CPW resonator is formed on the ground plane, with its first three resonant modes allocated in the desired UWB passband. In this aspect, the broadside-coupled microstrip-to-cpw transition or coupling structure with enhanced coupling strength is utilized to replace the PCML sections in the initial UWB bandpass filter in [6]. This structure can address two problematic issues existing in the initial MMR-based UWB filter in [6]: ) insufficiently tight coupling strength between the two side-to-side coupled lines and 2) parasitic S 2 (db) An aperture-backed PCML is formed by the partial removal of the ground plane underneath the coupled strips S (db) Figure. and measured results of the UWB bandpass filter on aperture-backed microstrip line [27]. April Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

7 Strip Conductor L λ g2 /4 λ g /2 λ g2 /4 ~λ g /4 ~λ g2 /2 ~λ g /4 Substrate Ground Plane Figure 2. Three-dimensional view of the proposed UWB bandpass filter on a hybrid microstrip/cpw structure [28]. radiation loss from a wide strip conductor in the center, especially at high frequencies. Figure 3 shows top and bottom view photographs of the filter fabricated on Rogers RT/Duroid 6 with e r =.8 and h =.635 mm. Similar to the filter in [6], only one wavelength is found in this five-pole filter design; the measured results are shown in Figure 3(c). Over a wide frequency range, flat insertion loss and group delay are observed. Parallel-Coupled CPW with Short-Circuited Ends A parallel-coupled CPW with short-circuited ends is used here as an inductive coupling structure that can avoid the high radiation loss of the opencircuited ends of the conventional capacitively parallel-coupled CPW [29]. Figure 4 is a 3-D view of an alternative UWB bandpass filter based on a short-circuited MMR on CPW and inductively parallel-coupled CPW structure [3]. Thus, a UWB filter can be implemented in the purely CPW topology that is required in direct integration with other uniplanar circuits. Z,φ Z 2,φ 2 Z 3,φ 3 Z 2,φ 2 Z,φ K Figure 4. Proposed short-circuited UWB bandpass filter on CPW [3]. Geometry. Equivalent circuit network. Equivalently, an asymmetrical parallel-coupled CPW with short-circuited ends can be expressed as a K-inverter network with the impedance (K) and two unequal line lengths (f and f 2 ) at two ends, as discussed in [3]. Therefore, the equivalent network topology of this filter can be described as a cascaded network, as shown in Figure 4. Here, three cascaded sections with the phases f 2, f 3, and f 2 between the two K-inverters make up the resultant short-circuited MMR. Figure 5 shows a photograph of the fabricated circuit and results from the calculation through the cascaded network in Figure 4, direct electromagnetic simulation, and microwave measurement. In the realized UWB passband of 3.3 GHz to.4 GHz, the measured maximum results, S 2, achieve.5 db, as compared with.2 db in the simulation, due to unexpected radiation loss in the CPW structure. K 6.9 mm S and S 2 (db) S 2 S Group Delay (c) Group Delay (ns) Figure 3. Photographs and measured results of a fabricated UWB filter using a hybrid microstrip/cpw structure [28]. Top view. Bottom view. (c) S-parameter magnitudes and group delay. S 2 (db) mil Momentum FCC Mask S (db) Figure 5. Photograph [3]. S-parameter magnitudes with FCC-defined UWB mask for indoor devices [3]. 94 April 29 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

8 UWB Filters with Improved Out-of-Band Performance Capacitive-Ended Interdigital Coupled Lines In the design of UWB bandpass filters using multiplemode resonators, the MMR needs to be constructed to allocate the first three resonant frequencies in the UWB passband of interest almost equally. Unfortunately, the fourth or other higher-order resonant frequencies may contribute spurious passbands and thus degrade the upper stopband in the resultant UWB filter. To circumvent this problem, an interdigital coupled line with capacitive-ended loading and/or tapered strip shape is constructed, and its first transmission zero is reallocated toward the full suppression of this fourth resonant frequency in the MMR [32]. Figure 6 shows the schematic and frequency responses of this improved UWB filter. This proposed filter is capable of reducing the insertion loss within the UWB passband and widening the upper-stopband bandwidth as confirmed experimentally. As shown in Figure 6, the measured return and insertion losses are higher than 4 db and lower than.3 db, respectively, over the UWB passband. The measured group delay varies slightly between.2 and.3 ns in the simulation. To further sharpen the roll-off skirts near the lower and upper cutoff frequencies, a two-stage UWB bandpass filter has also been designed with two cascaded MMRs, as reported in [32]. Coupling Between Two Feed Lines Another approach to enhance the filter selectivity is to introduce extra coupling between the two feed lines [33]. In [34], a UWB bandpass filter is presented using an MMR-based slotline resonator and back-to-back microstrip-slotline transition. This slotline resonator is twisted into a W shape, as shown in Figure 7 [34], in order to make the overall size smaller and create an additional coupling path. Without increasing the size, two low-pass filters are installed in the sections of two feed lines so as to improve the out-of-band performance. The filter, shown in Figure 7, was implemented on a substrate with dielectric constant of e r =.8 and h =.635 mm. The measured frequency response and group delay are plotted in Figure 8, which exhibits an ultrawide passband from 2.95 GHz to.6 GHz or % fractional bandwidth at 6.56 GHz. The midband insertion loss is.27 db, and the return loss is higher than 9.47 db within the whole passband. The in-band group delay is less than.98 ns with a maximum variation of.4 ns. The upper stopband is stretched up to 2 GHz as expected. On the other hand, an extra transmission zero is successfully excited around.5 GHz below the UWB passband due to the induced coupling between the two feed lines. This design allows us to implement such a UWB filter on a very thin substrate with relaxed fabrication tolerance. Unfortunately, the fourth or other higher-order resonant frequencies may contribute spurious passbands and thus degrade the upper stopband in the resultant UWB filter. S and S 2 (db) Figure 7. Photograph of the UWB bandpass filter based on stub-loaded MMR [34]. Top view. Bottom view. S and S 2 (db) Ω MMR 5 Ω S S S 2 Group Delay Figure 8. S-parameters and group delay of the UWB bandpass filter based on microstrip/slotline transition [34]. S Group Delay (ns) Figure 6. Schematic of the designed UWB bandpass filter [32]. and measured results of singlestage UWB bandpass filter with capacitive-ended interdigital coupled lines [32]. Group Delay (ns) April Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

9 The symbol l g represents the wavelength of the guided wave propagating in the microstrip. S 2 and S (db) Stub-Loaded MMR To miniaturize the overall size, a UWB filter as shown in Figure 9 was presented in [35] by loading three shunt open-circuited stubs into the middle section of the MMR. In this case, two identical stubs are introduced at two symmetrical positions with respect to the center plane of the low-impedance line section. This provides us with an additional degree of freedom in S and S 2 (db).6 mm 2.7 mm Strip Width:. mm Slot Width:. mm S Top View 2.29 mm 4.3 mm S db Figure 9. Schematic of the UWB bandpass filter based on stub-loaded MMR [35]. and measured results of the optimized filter [35]. Bottom-View S S Figure 2. Photograph of the stopband-enhanced MMR UWB bandpass filter [38]. and measured results [38]. S and S 2 (db) 2 4 S 2 S.7 mm Group Delay (ns) Figure 2. Photograph of the fabricated UWB bandpass filter with a notch-band [44]. and measured results [44]. relocating the first four resonant modes within the UWB band while pushing up the fifth resonant mode. As a result, out-of-band rejection skirts can be sharpened with a widened upper stopband. Figure 9 shows the filtering performance of the UWB bandpass filter described above. The lower and higher cutoff frequencies of the constituted UWB passband are equal to 2.8 and.27 GHz in experiment, compared with their counterparts of 2.98 GHz and.49 GHz in simulation. The fabricated filter achieves a return loss higher than 4.3 db over the realized UWB passband with a fractional bandwidth of 4%. Low-Pass Embedded MMR To suppress harmonics, various low-pass structures with high-frequency rejection can be embedded into the MMR itself, as in [36] [39], in order to make up a class of compact UWB bandpass filters with improved upper-stopband performance. In this respect, the filter topology primarily consists of a stopband-contained resonator and two parallel-coupled lines. Figure 2 is a photograph of the fabricated UWB filter [38]. As a benefit of three controllable transmission zeros excited in the capacitive-ended coupled lines, an improved UWB filtering performance with a stretched upper stopband up to 25 GHz is attained, as shown in Figure 2. This filter is also implemented on Rogers RT/Duroid 6 with a substrate thickness of.635 mm and a dielectric constant of.8. As shown in Figure 2, the measured insertion loss is.6 db at the center of the UWB passband and remains below 2. db from 3. GHz to 9. GHz. The measured return loss exceeds 3.5 db over the passband April 29 Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

10 TABLE. Comparison of measured UWB bandpass filters. Maximum In-Band Insertion Loss (db) Minimum In-Band Return Loss (db) Maximum In-Band Group Delay Variation (ns) Initial one on microstrip [6] Aperture-backed [27] < Broadside microstrip/cpw [28] Inductive coupling on CPW [3] Interdigital-coupled lines [32] Microstrip-slotline [34] Stub-loaded [35] Low-pass embedded [38] < Notch-band [44] <2...6 Experimentally, the 3-dB passband ranges from 3. GHz to.65 GHz, covering the FCC-specified UWB passband. In addition, the overall length of this filter is reduced to only about.52 mm. UWB Bandpass Filter with a Notch-Band All of the MMR-based UWB bandpass filters reviewed above have good in-band performance and are suitable for practical implementation. Nevertheless, the FCC-specified emission mask cannot provide enough protection from interference from generic UWB applications for existing and planned radio systems. Users will have to pay a great deal of attention to the exclusive use of the spectrum without causing interference to other existing services, such as wireless local-area network (WLAN) at 5.6 GHz. To circumvent this issue, single [4] [46] or multiple [46] narrow-band notched UWB filters were developed. By introducing slotline or open-circuited stubs into the filtering topologies, a notch, or narrow rejection band, can be generated in a certain frequency range in the UWB passband. In [43] and [44], asymmetric parallel-coupled or interdigital-coupled lines are designed for the excitation of a notch-band. Figure 2 shows a photograph of a fabricated UWB bandpass filter with a WLAN notch [44]. With the help of the asymmetrical topology of coupled line arms, a narrow notch-band is generated in [44]. and measured frequency responses of this filter are illustrated in Figure 2. Over the plotted frequency range of. GHz to 2. GHz, the realized UWB passband exhibits the emergence of a notch-band at 5.59 GHz with an 8.8-dB rejection and a 3-dB notch bandwidth of 4.6%. In the UWB passband, except for the notch-band, the return loss is higher than. db and the group delay is almost flat except for the sharp variation around the edges of such a notch-band. Conclusions In this article, the multiple resonant modes of an SIR-shaped MMR were extensively investigated and discussed based on transmission line theory. Practically, the MMR-based UWB filters presented exhibit a good UWB passband and compact overall size. To improve out-of-band performance and avoid interference with other served bands, we presented various UWB bandpass filter designs. Detailed comparisons of measured losses and group delays for the discussed UWB bandpass filters are tabulated in Table. Each of these filters exhibits unique features, such as an ultrawide passband, a wide upper stopband, an improved lower stopband, and a WLAN notch-band. As highlighted in this article, MMRs have played an indispensable role in the development of such a class of wide-band bandpass filters for UWB transmission systems. Acknowledgments The authors would like to express their sincere gratitude to Prof. Wolfgang Menzel at the University of Ulm in Germany, Dr. Hang Wang, Dr. Jing Gao, Ms. Rui Li, Mr. Sai Wai Wong, and Mr. Teck Beng Lim at Nanyang Technological University in Singapore for their technical contributions to this article. References [] Federal Communications Commission, Revision of part 5 of the commission s rules regarding ultra-wideband transmission systems, FCC, Washington, DC, Tech. Rep. ET-Docket 98-53, FCC2-48, Apr. 22. [2] J. D. G. Swanson, Narrow-band microwave filter design, IEEE Microwave Mag., vol. 8, no. 5, pp. 5 4, Oct. 27. [3] G. L. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures. Dedham, MA: Artech House, 98. [4] R. Levy and S. B. Cohn, A history of microwave filter research, design, and development, IEEE Trans. Microwave Theory Tech., vol. 32, no. 9, pp , Sept [5] J.-S. Hong and M. J. Lancaster, Microstrip Filters for RF/Microwave Applications. New York: Wiley, 2. April Authorized licensed use limited to: KIZ Abt Literaturverwaltung. Downloaded on May 4, 29 at 5:28 from IEEE Xplore. Restrictions apply.

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