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1 684 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 3, MARCH 2009 Synthesis Design of Ultra-Wideband Bandpass Filters With Composite Series and Shunt Stubs Rui Li, Student Member, IEEE, Sheng Sun, Member, IEEE, and Lei Zhu, Senior Member, IEEE Abstract This paper presents a direct synthesis procedure for efficient design of a class of ultra-wideband bandpass filters with composite series and shunt transmission-line stubs. The proposed single-stage bandpass prototype is formed by cascading two sets of series open-ended and shunt short-ended stubs through a multisection nonuniform transmission line. All the stubs are set with an identical electrical length, i.e.,, at lower cutoff frequency and the middle connecting line is composed of stepped-impedance transmission line with sections, and each section has an electrical length. The proposed filter topology aims to construct an ultrawide bandpass filter with Chebyshev equal-ripple responses and ( +3)in-band transmission poles. Based on the derivation of the transfer function, a synthesis approach is established and systematically described to design this type of filters according to the specifications such as lower/upper cutoff frequencies. Next, a multistage bandpass filter prototype is proposed and the synthesis design procedure is also presented. The implementation is achieved by using hybrid microstrip line and slotline sections. Compared with traditional stub filters, the proposed filters achieve higher order transmission zeros and thus provide higher selectivity. As design examples, two single-stage and one three-stage bandpass filters are designed and fabricated to confirm the theoretical predictions. Index Terms Bandpass filter, equal-ripple response, multistage, series/shunt stubs, single stage, synthesis design, ultra-wideband. I. INTRODUCTION SINCE THE Federal Communications Commission (FCC) in the U.S. authorized the unlicensed use of ultra-wideband devices in the frequency band of GHz in 2002 [1], great interest has been aroused from both academic and industrial areas toward ultra-wideband technology. As a key component in the ultra-wideband wireless communication systems, microwave bandpass filters with high performance, compact size, and low cost are highly demanded. Numerous ultrawideband bandpass filters with various desired features have been reported [2] [15]. Nevertheless, these filters are usually designed relying on the cut-and-try approach where the overall filter layout with electrically large size is simulated again and Manuscript received October 15, 2008; revised December 10, First published February 13, 2009; current version published March 11, R. Li and L. Zhu are with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore ( liru0003@ntu. edu.sg; ezhul@ntu.edu.sg). S. Sun was with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore He is now with the Institute of Microwave Techniques, University of Ulm, Ulm, Germany ( sunsheng@ieee.org). Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TMTT again using the commercial software towards the specified ultrawide passband. As a result, there is a demand for developing a systematic synthesis approach such that these bandpass filters with a fractional bandwidth of 110% can be efficiently designed. To address this issue, the conventional high-pass prototype with short-circuited shunt stubs [16] has been adopted in [2] [6] to explore various bandpass filters with ultra-wide bandwidth. In [2], this type of wideband filter was initially presented and then systematically synthesized in [3]. However, the resultant filter block in [3], composed of high-pass, low-pass, and isolation sections, occupied a large circuitry area. Hence, extensive work has been carried out toward miniaturizing the overall filter size and improving the electrical performance at the same time. In [4], an electromagnetic (EM) bandgap structure was formed on the connecting line between two adjacent stubs to suppress the upper spurious harmonics. An extra cross-coupling route was purposely introduced between the input and output ports in [5] and [6] to enhance the filter selectivity and to improve the group-delay performance. However, in constructing these filters on microstrip-line topology, via-holes are always required to realize the short-ended terminals for all shunt stubs. As a dual of the high-pass prototype with shunt short-circuited stubs, a high-pass filter with series open-circuited stubs was theoretically studied in [17]. Later, this filter network was implemented using a 3/4 wavelength parallel-coupled line resonator to synthesize and design an ultra-wideband bandpass filter [15]. For the readers reference, it needs to be pointed out that the initial generating functions with bandpass filtering behavior were presented in [18] for an ideal transmission-line filter network. However, no work has been reported thus far to physically implement such a microwave filter based on any transmission-line structure and to experimentally demonstrate its real filtering performances. In this paper, an alternative class of ultra-wideband bandpass filters with single and multiple stages implemented using composite series and shunt stubs is proposed. In Section II, a synthesis approach is developed to efficiently design the singlestage bandpass filters in the closed-form format following the studies in [16] [18]. Fig. 1(a) and (b) depicts the general transmission-line models with an odd- and even-integer, respectively. In the synthesis procedure, the transfer functions are at first analytically derived and regulated to exhibit the Chebyshev equal-ripple frequency responses in the ultra-wide passbands. Under equalization of the insertion-loss functions, all the normalized element values in the prototype filters can be accordingly determined. Next, this procedure is applied to synthesis design a multistage bandpass filter. Since there has been no reported work in practical implementation for these filter /$ IEEE

2 LI et al.: SYNTHESIS DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS 685 Fig. 1. Transmission-line model of the proposed single-stage bandpass filter prototype based on composite stubs and an i-section nonuniform connecting line. (a) i is odd-integer (1; 3; 5;...). (b) i is even-integer (0; 2; 4; 6;...). topologies thus far, a multistage prototype ultra-wideband bandpass filter is, for the first time, proposed, designed, and implemented in this work. In the following, two single-stage and one three-stage bandpass filters are designed on hybrid via-free microstrip-line and slotline structures. The predicted frequency responses are evidently verified by direct full-wave simulation of their overall filter layouts and experimental measurement of the fabricated filter circuits. where and are the entries of the overall matrix of the network. Two single-stage prototype filters with and will be illustrated in detail as examples for the odd and even integer. For the filter prototype in Fig. 1(a) with and stage number, the function can be derived as (2) II. SINGLE-STAGE FILTER PROTOTYPE A. Synthesis Procedure The proposed transmission-line model is composed of two sets of series open-ended and shunt short-ended stubs that are connected through an -section nonuniform transmission line in the middle. It is depicted in Fig. 1(a) and (b) for the odd- and even-integer, respectively. All the series and shunt stubs are set with the same electrical length at lower cutoff frequency. The normalized impedances for the series and shunt stubs are labeled as and, where indicates the th set of composite stubs. Meanwhile, each section of the middle connecting line is exactly chosen as, and the normalized impedances of the first section to the one at the center are named as to for the odd integer and to for the even integer. The subscript indicates the connecting line is between the th and th sets of stubs, and the superscript indicates the section number. Plus, the network is horizontally symmetrical while the input and output are terminated by two unit normalized impedances, i.e.,. The overall matrix of the proposed transmission-line model in Fig. 1 can be obtained by multiplying the matrices of individual sections in sequence. Hence, the squared magnitude of transmission coefficient can be expressed in terms of a characteristic function [16] as (1) where (3a) (3b) (3c) As discussed in early works [16] [18], to exhibit Chebyshev responses with equal-ripple in-band behavior and transmission zeros at, the squared -magnitude of the filters can be expressed as and (4) (5a) (5b) where is the specified equal-ripple constant in the passband and ( is the phase at lower cutoff frequency ). Due to the frequency-distribution characteristic of transmission lines, the upper cutoff frequency of this bandpass filter appears

3 686 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 3, MARCH 2009 at. Based on the relationship among those variables in (5a) and (5b), the cosine function in (4) can be expanded as where and are the Chebyshev polynomial functions of the first and second kinds of degree, respectively. Using the identity for and the targeted transfer function in (4) can be analytically expressed in terms of these two kinds of Chebyshev functions. In order to make (1) have the same response as (4), and in (6) must be at first selected as 1 and 3, while the condition in (8) must be enforced as follows: (6) (7) by 1. To achieve the equalization of (1) with (4), and in (6) now have to be chosen as 2 and 3, respectively. As a result, a complicated, but explicit expression can be derived as given in (12), and it represents the necessary condition in this synthesis procedure. By equalizing the corresponding coefficients for different-order variables in (10) and (12), a set of equations can be derived and they are similar to those in (9a) (9c). By solving these equations, normalized characteristic impedances for all the involved stubs and connecting line can be calculated as follows: After equalizing and in (2) with the coefficients of and in (8), the following set of equations can be derived: (8) (9a) (9b) (9c) Therefore, the normalized characteristic impedances and for the series open-ended stub, shunt short-ended stub, and the middle connecting line can be explicitly determined once and or fractional bandwidth are specified. Particularly, (9c) indicates that the normalized characteristic impedance of the connecting line with the length of only relies on the specified ripple constant. Using the same analysis approach, the characteristic function of the filter prototype shown in Fig. 1(b) with and can be derived as where (10) (11a) (11b) (11c) Looking at (10) and (2) comparatively, the highest order in denominator is unchanged, but that in the numerator is increased (12) As the element values are determined according to the specified ripple constant and lower cutoff frequency, all the coefficients in the characteristic functions (2) and (10) can be calculated via (3a) (3c) and (11a) (11c). As such, can be derived from (2) and (10). Fig. 2(a) and (b) plots the frequency responses of two single-stage Chebyshev bandpass prototypes with and, and, and fixed ripple level db, which related to the given ripple constant by (13) As shown in the inserted frames of these figures, the upper cutoff phase is exactly equal to 150, 135, and 120,as its lower counterpart is selected as the above three values. On the other hand, the ripple levels are all restricted to 0.1 db as pre-specified. Moreover, the simulated results show that the passbands repeatedly appear at the periodicity of 180. Thus, the next passband will start at and end at. Looking at Fig. 2(a) and (b) comparatively, the total number of in-band transmission poles for these two filter prototypes are 4 and 5, respectively. As a result, the five-pole bandpass filter has a larger cutoff attenuation skirt than its four-pole counterpart under the same specifications, i.e., and. The extra transmission pole in Fig. 2(b) is introduced by an extra section in the middle line. If the prototype filter has sections of nonuniform transmission lines, as shown in Fig. 1, the highest degree of the numerator for the derived function will equal to, which implies that the number of in-band transmission poles is. On the other hand, the highest degree in the numerator of the expanded (6) is. Therefore, and should equal to and, respectively, in order to equalize the magnitudes of two characteristic functions. Now, let us compare the proposed single-stage filters with the reported ones in [2] [6] and [15] that have either shunt shortended or series open-ended stubs. Observing the polynomial functions in (2) and (10), one can understand that the number of in-band transmission poles exactly equals to the highest order

4 LI et al.: SYNTHESIS DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS 687 Fig. 3. Comparison in js j between the conventional and proposed five-pole bandpass filter prototypes with purely shunt stubs [2] or series stubs [15] and composite stubs. Fig. 2. Predicted frequency responses of two bandpass filter prototypes under varied electrical length. (a) i =1. (b) i =2. of in the numerator. The highest order of in the denominator is and it represents the existence of third-order transmission zeros in the lower and upper ends of the passband, i.e., and. It needs to be highlighted that there exist only first-order transmission zeros in the high-pass prototypes with purely shunt/series stubs in [2] [6] and [15]. Thus, the out-of-band rolloff or attenuation skirts should be significantly enhanced, especially for the multistage prototype with multiplying attenuation that will be introduced later. Fig. 3 plots the simulated graphs of the filter in [2] or [15] and the proposed filter all with five poles in the passband for comparative study. It is quantitatively confirmed that the proposed filter prototype with composite series/shunt stubs has much higher attenuation skirts in the lower stopband of and the upper stopband band of than those with either series or shunt stubs. It is also worth mentioning that the opposite prototype with series short-circuited and shunt open-circuited stubs can be utilized to design low-pass filters with shaper rolloffs [19]. B. Filter Implementations Stemming from the proposed single-stage filter prototypes shown in Fig. 1(a) and (b) for and, two bandpass filters with four and five poles will be designed and implemented on hybrid slotline and microstrip line structures in this part. To account for the distinctive features of these two transmission lines, Fig. 4. Schematic layouts of two proposed ultra-wideband bandpass filters implemented on hybrid microstrip and slotline structures. (a) Four-pole. (b) Fivepole. MSL: microstrip line. the shunt short-ended stubs are formed on the slotline, while the series open-ended stubs are constructed on the microstrip line. In synthesis design based on the filter prototypes in Fig. 1, the microstrip-to-slotline right-angle cross-transition section can be simply expressed as an equivalent transformer with a unity turns ratio [20] since the substrate is selected electrically thin. The substrate used in the study is RT/Duroid 6010 with a dielectric constant of 10.8 and a thickness of mm. Using the closed-form design formula for slotline and microstrip line in [20], the schematic layouts of two filters can be initially determined using the above synthesis procedure and they are depicted in Fig. 4(a) and (b) with all the dimensions indicated. As illustrated in Fig. 4(a) and (b), the microstrip line is formed on one side of the substrate, while the slotline is etched on the opposite side that is perpendicular with the microstrip line.

5 688 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 3, MARCH 2009 TABLE I CIRCUIT PARAMETERS FOR FOUR-POLE FILTER ( 3 GHz) TABLE II CIRCUIT PARAMETERS FOR FIVE-POLE FILTER ( 3 GHz) Fig. 6. Five-pole ultra-wideband bandpass filter in Fig. 4(b). (a) Predicted and measured S-parameters. (b) Predicted and measured group delays. TABLE III COMPARISONS FOR FOUR-POLE BANDPASS PROTOTYPE FILTER Fig. 5. Four-pole ultra-wideband bandpass filter in Fig. 4(a). (a) Predicted and measured S-parameters. (b) Predicted and measured group delays. The electrical lengths from the microstrip-line open-ends and slotline short-ends to the intersection points are both equal to. The connecting slotline is set as or. In this design, and are readily selected with reference to the graphs in Figs. 2 and 3, aiming to work out the four- and fivepole bandpass filters with an ultra-wide passband. From the procedure previously discussed, three normalized impedances and are at first determined such that the characteristic impedances and of all the stubs and connecting line in these two filters can be calculated. Tables I and II tabulate the values of these impedances. C. Simulated and Measured Results Without any change in filter dimensions, as shown in Fig. 4(a) and (b), full-wave EM simulator [21] is employed to simulate their overall filter layouts. Two relevant filter circuits are fabricated and measured. Figs. 5 and 6 illustrate three sets of -magnitudes and groups delays, which are obtained from the transmission-line model, full-wave simulation, and microwave measurement, respectively. All these three sets of frequency responses and group delays are found in good TABLE IV COMPARISONS FOR FIVE-POLE BANDPASS PROTOTYPE FILTER agreement with each other over the plotted frequency range. The detailed comparisons of the electrical parameters are tabulated in Table III and IV for these two bandpass filters. Four and five transmission poles can be clearly observed in the realized wide passband for both filters. The positions of the poles and zeros match each other well, especially for the two sets of graphs obtained from the transmission-line model and full-wave simulation. Meanwhile, the in-band group delays for the four- and five-pole bandpass filters are all flat in the middle passband, and become higher near the lower and upper cutoff

6 LI et al.: SYNTHESIS DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS 689 Fig. 7. Transmission-line model of a three-stage (N =3)bandpass filter prototype with i =1. frequencies as the tradeoff of high rejection skirts. Similar results can also be observed in reported works [2] [15]. On the other hand, some visible discrepancies exist among the three sets of results. To our knowledge, the extra poles appearing in the measured results are primarily brought by multiple reflections along the feed lines due to the unexpected discontinuity effects in the subminiature A (SMA) connectors and input/output terminals of these filters. The frequency shift at high frequencies between the predicted and measured results may be caused by many uncontrollable factors, such as frequency dispersion of the non-tem slotline mode, frequency-dependent permittivity, misalignment of slot/strip positions in two layers, and tolerance in etching fabrication. Radiation loss in the slotline s area can be certainly reduced by packaging these filters in a metallic enclosure, but it increases the total circuitry size and generates unexpected cavity resonances that are strongly dependent of the length/width of this enclosure. The primary purpose of implementing these filters on hybrid microstrip line and slotline structures is to quantitatively verify the synthesis design based on the proposed transmission-line model. Overall, both simulated and measured results have confirmed the feasibility and accuracy of the developed synthesis procedure very well. It is interesting to compare the proposed prototype with those reported in [8] and [9]. In this case, the latter ones can be treated as the specified cases of the former one with. The original idea in [8] and [9] is to consider the overall slotline section as a triple-mode resonator with uniform and nonuniform configurations. In particular, the nonuniform section with different characteristic impedances provides more flexibility and capability in the passband selection. With proper adjustment of impedance ratio, the first three resonances can be relocated uniformly within the desired passband. Together with the tight coupling of microstrip-to-slotline transitions with two additional transmission poles, an ultra-wide passband with varied bandwidth and five in-band poles can be realized. By observing the element values obtained from the above synthesis design, it can be found that as the fractional bandwidth increases, the normalized characteristic impedances become higher for two shunt short-ended stubs and lower for the middle line. This phenomenon exactly matches the multiple-mode theory as reported in [8] and [9]. On the other hand, in the above proposed prototype, the short-/open-circuited stubs have the electrical length at lower cutoff frequency, which is equal to 90 at the center frequency of the passband. This approach well explains and verifies the designs in [8] and [9] from the synthesis design point of view. Therefore, instead of time-consuming cut-and-try method via full-wave simulators, this synthesis method allows us to efficiently design these wideband filters using a set of closed-form design formula. A. Synthesis Procedure III. THREE-STAGE FILTER PROTOTYPE To extend the above synthesis approach to the multistage filter design with improved out-of-band attenuation performance, a three-stage bandpass filter prototype with will be studied herein and its transmission-line model is shown in Fig. 7. This filter is composed of cascading four sets of composite stubs through three sections of transmission line. In the synthesis design, all the series stubs, shunt stubs, and connecting lines have a fixed electrical length as at the specified lower cutoff frequency. The normalized characteristic impedances of the first and second sets of hybrid stubs are indicated as and, respectively, while the ones for the connecting lines between the first set and second set of stubs, the second and third set of stubs are and. Since the entire network is symmetrical with respect to the central plane, the element values of the right half of the network are the same as their corresponding mirror images on the left. Again, the matrix of the entire network can be obtained easily by multiplying the individual matrices of each section in sequence and the characteristic function is derived as follows in (14) and the coefficients of the polynomial are functions of the six element variables are shown as follows in (15): where (14) (15) Function has a similar format as those in (2) and (10), except that the highest order in the numerator is 8 and that in the denominator is 5. Compared with (6) with and, five equations can be derived in terms of six variables. In this aspect, the element values are determined in such a way that the insertion loss function derived from the network is approaching the targeted one. It needs to be noticed that the transmission zeros appeared in multiple integer times of 180 are fifth-order transmission zeros. Thus, if both single- and three-stage bandpass filters are designed under the same specifications, i.e., ripple constant, lower cutoff phase, and same number of transmission poles, the rejection skirts of the three-stage filter definitely becomes sharper at the passband edges.

7 690 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 3, MARCH 2009 Fig. 9. Fabricated eight-pole ultra-wideband bandpass filter in Fig. 8. (a) Top view. (b) Bottom view. Fig. 8. Schematic layout of the proposed three-stage ultra-wideband bandpass filter shown in Fig. 7. TABLE V CIRCUIT PARAMETERS FOR EIGHT-POLE FILTER ( 3 GHz) increases the applicable impedance range in practical design and implementation of these filters. To verify the theoretical predictions, the designed three-stage bandpass filter is finally fabricated and measured. Fig. 9(a) and (b) shows the top- and bottom-view photographs of the fabricated circuits. The measured frequency responses and group delays are plotted in Fig. 10(a) and (b), respectively. Basically, the measured results agree with the simulations well over the frequency range of GHz, achieving a wide passband with the fractional bandwidth of 101.6% at 5.51 GHz. The insertion loss is lower than 2.32 db and the return loss is larger than 7.2 db in the core passband. The measured group delays are basically flat within the passband, while they tremendously vary as the frequency approaches the passband edges, as exhibited in the reported bandpass filters [2] [16]. B. Filter Implementation One design example is given to illustrate the three-stage case with at 3 GHz. The relevant transmission-line model shown in Fig. 7 is implemented using hybrid microstrip line and slotline structures as the single-stage cases. The schematic layout is depicted in Fig. 8 with all dimensions indicated, and design specifications are given in Table V. The series open stubs are all constructed by microstrip lines with open-circuited ends, while the shunt short stubs are all formed by slotlines on the ground. The uniform lines between two sets of stubs can be either microstrip line or slotline. From the multiple-mode resonator point of view [7], this constituted filter is realized by three such resonators, i.e., one microstrip line resonator and two slotline resonators, which are coupled via microstrip-to-slotline transitions. The first two resonant modes of these three resonators contribute to six transmission poles. Together with two poles produced by the input/output coupling structures, this filter can generate eight transmission poles in the passband in total. Due to the physical constraint in achieving specified values of characteristic impedances within the filter layout, the terminal impedances at the two references are chosen as 70. To connect this filter with two 50- coaxial cables in measurement, two quarter-wavelength transformers are additionally designed at the central frequency and they are installed in the two main feed lines, as shown in Fig. 8. Although the transformers degrade the filter performance to a certain degree, it indeed IV. DISCUSSIONS In general, the proposed transmission-line model is more attractive in the design of bandpass filters with ultra-wide bandwidth due to its inherent high-pass property. By using the hybrid microstrip line and slotline to implement the presented filter networks, the range of lower cutoff phase is restricted to due to the limitation of realizable characteristic impedances of the transmission lines. Therefore, the range of applicable fractional bandwidth for the proposed filter structure can be estimated as about 85% 155% under the specified ripple level and pole number in the desired passband. In addition, characteristic impedance and effective permittivity of the slotline involved in this proposed filter are functions of frequency within the passband. It is one primary factor that causes frequency shift in a high frequency range between the synthesis-predicted results and EM-simulated or measured results, as shown in Figs. 5, 6, and 10. To minimize this parasitic effect, a low-impedance slotline with narrow slot width is preferred in implementation if possible. Under the restriction in the synthesis procedure that requires the constant characteristic impedance and effective permittivity, these two parameters for the slotline at center frequency of the passband should be selected. On the other hand, as mentioned earlier, there exist some visible derivations between the synthesis-derived results and EM-simulated or measured results. They are primarily attributed by the common drawback of all the synthesis design procedures in microwave planar filters, i.e., no ability in taking

8 LI et al.: SYNTHESIS DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS 691 provides us with an efficient, effective, and accurate solution in design and exploration of a class of ultra-wideband bandpass filters with composite short/open stubs. REFERENCES Fig. 10. Eight-pole ultra-wideband bandpass filter in Fig. 8. (a) Predicted and measured S-parameters. (b) Predicted and measured group delays. into account all of the unexpected dynamic effects inclusive of frequency dispersion, discontinuity effects, conductor/material/radiation losses and so on. However, these effects have been appearing in both EM-simulated and experimental results. As shown in Figs. 5, 6, and 10, these two sets of results are well matched with each other. Even though it is theoretically approximate, the synthesis procedure is always highly desired in filter design. It is because that the synthesis approach not only gives people a deeply physical insight into the operating principle for the developed microwave circuits, but also remarkably reduces the computation time and design circles especially for the high-order filter structures with many unknown dimensions. In this case, all the dimensions obtained from the synthesis approach can be used as the initial dimensions for more accurate EM-based simulation, tuning, and optimization. V. CONCLUSION A direct synthesis approach for a class of ultra-wideband single-stage and multistage bandpass filters has been systematically presented in this paper. The transfer functions derived from the exact networks were firstly formulated to realize the Chebyshev equal-ripple responses in the wide passband and to explicitly determine the normalized values of transmission-line elements under the required specifications. 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9 692 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 3, MARCH 2009 Rui Li (S 06) received the B. Eng. degree in electrical and electronic engineering from Nanyang Technological University (NTU), Singapore, in 2005, and is currently working toward the Ph.D. degree in electrical and electronic engineering at NTU. Her research interests include synthesis design and practical implementation of microwave passive devices for ultra-wideband wireless communication systems. Ms. Li was the recipient of the NTU Research Scholarship ( ) and Ministry of Education Scholarship ( ), Singapore. Sheng Sun (S 02 M 07) received the B.Eng. degree in information and communication engineering from Xi an Jiaotong University, Xi an, China, in 2001, and the Ph.D. degree in electrical and electronic engineering from Nanyang Technological University (NTU), Singapore, in From 2005 to 2006, he was with the Institute of Microelectronics, Singapore. From 2006 to 2008, he was with the School of Electrical and Electronic Engineering, NTU. Since 2008, he has been a Humboldt Research Fellow with the Institute of Microwave Techniques, University of Ulm, Ulm, Germany. His current research interests include the study of multilayer planar circuits, microwave filters and components, antennas, numerical modeling and deembedding techniques, as well as millimeter-wave and microwave interconnects. Dr. Sun was the recipient of a 2008 Hildegard Maier Research Fellowship of the Alexander von Humboldt Foundation, Germany. He was also the recipient of the Young Scientist Travel Grant presented at the 2004 International Symposium on Antennas and Propagation, Sendai, Japan, and the NTU Research Scholarship, Singapore. Lei Zhu (S 91 M 93 SM 00) received the B. Eng. and M. Eng. degrees in radio engineering from the Nanjing Institute of Technology (now Southeast University), Nanjing, China, in 1985 and 1988, respectively, and the Ph.D. Eng. degree in electronic engineering from the University of Electro-Communications, Tokyo, Japan, in From 1993 to 1996, he was a Research Engineer with Matsushita-Kotobuki Electronics Industries Ltd., Tokyo, Japan. From 1996 to 2000, he was a Research Fellow with the Ecole Polytechnique de Montreal, University of Montreal, Montreal, QC, Canada. Since July 2000, he has been an Associate Professor with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore. His research interests include planar filters, planar periodic structures, planar antennas, numerical EM modeling, and deembedding techniques. He has authored or coauthored over 165 papers in peer-reviewed journals and conferences, including 17 in the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES and 28 in IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS and IEEE MICROWAVE AND GUIDED WAVE LETTERS. He was an Associate Editor for the IEICE Transactions on Electronics ( ). Dr. Zhu has been an associate editor for IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS since October He has been a member of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) Technical Committee 1 on Computer-Aided Design since June He was a general chair of the 2008 IEEE MTT-S International Microwave Workshop Series (IMWS 08) on Art of Miniaturizing RF and Microwave Passive Components, Chengdu, China. He was the recipient of the 1997 Asia Pacific Microwave Prize Award, the 1996 Silver Award of Excellent Invention presented by Matsushita-Kotobuki Electronics Industries Ltd., and the 1993 First-Order Achievement Award in Science and Technology presented by the National Education Committee, China.