A Quarter-Wavelength Shorted Microstrip Antenna with a Slot for Dual-Frequency Operation

Transcription

1 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY PAPER Special Issue on Microwave and Millimeter-Wave Technology A Quarter-Wavelength Shorted Microstrip Antenna with a Slot for Dual-Frequency Operation Takashi AMANO a), Norimichi CHIBA, and Hisao IWASAKI, Members SUMMARY A novel dual-band internal antenna similar in size to the single-band internal antenna for cellular handsets is proposed. Our approach to realize a small and low-profile dual-band internal antenna is to use the dominant mode (TM 10 mode)and the higher-order mode (TM 30 mode). In order to use this approach for recent dual-band cellular systems it is necessary to lower the resonant frequency of the higher-order mode (TM 30 mode). This motivated our development of a new antenna configuration with a slot on the radiation element of a quarter-wavelength shorted microstrip antenna to lower the resonant frequency of the TM 30 mode. In this paper, the experimental and the analytical results for this antenna are presented. In the results, by adjusting the location and the length of the slot, the dual-frequency operation can be achieved with the frequency ratio (TM 30 mode/tm 10 mode)from 2 to 3. In addition, the enhancement of bandwidth is presented. key words: dual-frequency, dual-band, microstrip antenna, slot 1. Introduction Cellular communication systems have shown a rapid growth and many types of cellular handsets have been developed around the world. Recently, dual-band cellular systems, such as GSM (900 MHz) and DCS (1800 MHz) or AMPS (800 MHz) and PCS (1900 MHz), have been introduced. In such dual-band cellular handsets, dual-band antennas for handsets have been required now. Here, a small and low-profile antenna is required to keep the handset compact. An internal antenna is more suitable than a wired antenna for compact handsets. Up to now, many types of dual-band internal antennas based on two separate antenna elements has been developed (Fig. 1). In these antennas, the antenna elements are generally arranged either stacked [1], [2] (Fig. 1) or side-by-side [3] [6] (Fig. 1). A compact planar inverted F antenna, which is suitable for the dual-frequency operation, has been also developed by using a capacitive feed and a capacitive load [7]. In another significant development, dual-frequency operation with the frequency ratio (TM 30 mode/tm 10 mode) from about 1.6 to 2 has been achieved using slots close to the edge of a microstrip antenna [8]. The Manuscript received December 26, Manuscript revised March 15, The authors are with Wireless Communication Technology Center, Toshiba Corp., Hino-shi, Japan. a) takashi.amano@toshiba.co.jp This paper was presented at 1998 Asia-Pacific Microwave Conference. development of a small and low-profile antenna using these antenna configurations is, however, expected to be difficult. In this paper, a novel dual-band internal antenna similar in size to the single-band internal antenna is proposed. Our approach to realize a small and low profile is to use the dominant mode (TM 10 mode) and higher-order mode (TM 30 mode). In order to use this approach for recent dual-band cellular systems, it is necessary to lower the resonant frequency of the TM 30 mode. Because the required frequency ratio in the above-mentioned systems is about 2 to 3, the optimum location of a slot in the radiation element of a quarterwavelength shorted microstrip antenna is proposed to lower the resonant frequency of the TM 30 mode. The performance of our proposed antenna has been measured, simulated and is discussed in this paper. 2. Antenna Configuration Figure 2 shows the configuration of our proposed antenna. There is a slot, with dimensions Ls and Ws, in the radiation element of a quarter-wavelength shorted microstrip antenna, with dimensions L and W, to lower the resonant frequency of TM 30 mode. The height of the radiation element is T from the finite ground plane. The length of the shorting plate is Lg. The dual-frequency operation can be achieved by using the dominant mode (TM 10 mode) and higher-order mode (TM 30 mode). To use this dual-band approach in the above-mentioned systems, it is necessary to adjust the frequency ratio (TM 30 mode/tm 10 mode) to between 2 and 3. Figure 3 shows models of current distributions on the radiation element of a quarter-wavelength shorted microstrip antenna. It is expected that by locating a Fig. 1 Dual-band internal antennas based on two separate elements. Stacked type, Side-by-side type.

2 1212 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY 1999 Table 1 Experimental dimensions. Fig. 2 Antenna configuration. Fig. 3 Models of current distributions. Fig. 4 Resonant frequency versus slot length (Ls). slot on the radiation element, the current distribution will be modified and the route of the current distribution will be longer, so the resonant frequency will be lowered. If the slot is located at the point of maximum current of the TM 30 mode, the current distribution of the TM 30 mode will be strongly modified along the slot, without causing any significant difference in the current distribution of the TM 10 mode [8]. Then, it is expected that the resonant frequency of the TM 30 mode will be lowered significantly by locating the slot at the point of maximum current of the TM 30 mode compared with that of TM 10 mode. 3. Experimental and Simulation Results The antenna was designed, fabricated and tested to verify the dual-frequency operation of the proposed antenna configuration (Fig. 2). The experimental dimensions are shown in Table 1. The slot length (Ls) and the location of the slot (S) were varied experimentally to confirm the relationship between Ls, S and the resonant frequency of each resonant mode. The antenna was mounted over the finite ground plane and the dielectric constant of the material between the antenna element and the finite ground plane is 1 (air). Figure 4 shows the relationship between the slot length (Ls) and the resonant frequency of each resonant mode with the location of the slot (S) as a parameter. The length of the slot (Ls) is varied from 0mm (no slot) to 50mm and the location of the slot (S) is varied from 30mm to 60mm. Ls = 0 indicates the conventional antenna which has no slot. Generally, the resonant frequency of the TM 30 mode is about three times larger than that of the TM 10 mode. In the results, the resonant frequency of each mode decreases gradually with the slot length (Ls). In the case of S = 50mm, the resonant frequency of the TM 30 mode is lowered significantly, but the resonant frequency of the TM 10 mode shows no major difference. On the other hand, in the case of S = 30mm, the resonant frequency of the TM 10 mode is lowered, but the TM 30 mode shows no major difference. This means that if the slot is located near the point of maximum current in the radiation element, the resonant frequency is lowered significantly. With these antenna dimensions, good dual-frequency operations were obtained experimentally with a frequency ratio (TM 30 mode/tm 10 mode) from 2 to 3. Figure 5 shows the example of measured returnloss characteristics of this antenna. We have confirmed experimentally that a dual-frequency operation with a frequency ratio of about 2 can be obtained by adjusting the slot length (Ls) and the slot location (S). Figures 6 through 9 show measured radiation patterns of our proposed antenna compared with those of a conventional antenna. Solid lines show the copolarization (Etheta) pattern, and dashed lines show the cross-polarization (Ephi) pattern. The coordination is shown in Fig. 2. Figure 6 shows TM 10 mode H-

3 AMANO et al: A QUARTER-WAVELENGTH SHORTED MICROSTRIP ANTENNA 1213 Fig. 5 Return loss characteristics. Fig. 8 Radiation pattern of TM 30 mode (H-plane (X-Y plane)). Conventional, Proposed. Fig. 6 Radiation pattern of TM 10 mode (H-plane (X-Y plane)). Conventional, Proposed. Fig. 9 Radiation pattern of TM 30 mode (E-plane (Z-X plane)). Conventional, Proposed. Fig. 7 Radiation pattern of TM 10 mode (E-plane (Z-X plane)). Conventional, Proposed. plane (X-Y plane) radiation patterns and Fig. 7 shows TM 10 mode E-plane (Z-X plane) radiation patterns. In the dominant mode (TM 10 mode), broad radiation patterns can be obtained and these are similar to the conventional antenna. This indicates that the current distribution of the TM 10 mode is not significantly affected by locating a slot at the point of maximum current of the TM 30 mode. On the other hand, Fig. 8 shows the TM 30 mode H-plane radiation patterns and Fig. 9 shows the TM 30 mode E-plane radiation patterns. In the higher-order mode, the conventional antenna has broadside nulls in the H-plane, but these are eliminated when a slot is located in the radiation element, and the cross-polarization level increases in the H-plane when the slot is used. This indicates that the slot generates the current distribution toward the y direction, so the radiation patterns are different from the conventional antenna and are more suitable for handset antennas. Figure 10shows the simulation results of electric fields between the radiation element and the finite ground plane. The finite element method has been used for this simulation. Figures 10 and 10(d) show the simulation models of the conventional antenna and the proposed antenna. Figures 10 and (e) show the TM 10 mode electric field, and Figs. 10(c) and (f) show the TM 30 mode electric field. In these simulation results, it is confirmed that the TM 10 mode electric fields are similar to each other, but the TM 30 mode electric fields are strongly modified by locating the slot at the point of the TM 30 mode maximum current, and these are distributed near this slot. By the way, the TM 30 mode cross-polarization level increased when the slot was used, as shown in Figs. 8 and 9. From these measured and simulated results, it appears that the TM 30 modes radiation fields have been mainly generated by

4 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY Fig. 10 Elactric fields (simulated). Simulation model for conventional antenna, Conventional TM10 mode, (c) Conventional TM30 mode, (d) Simulation model for proposed antenna, (e) Proposed TM10 mode, (f) Proposed TM30 mode. Fig. 12 Fig. 11 Antenna configuration for enhanced bandwidth. the slot. It is necessary to verify this theory. 4. Enhanced return loss characteristics. Enhancement of Bandwidth Generally, enhancement of the bandwidth is expected when such an antenna is applied to a practical dualband cellular system. This is because this type of antenna, which is based on a microstrip antenna, has a narrow bandwidth compared with the wired antenna, which is generally used for handsets. Figure 11 shows the antenna configuration for enhanced bandwidth. Generally in the type of this antenna, such as an inverted F antenna, the size reduction of the antenna is realized by decreasing the length (Lg) of the shorting plate. Here, it is expected that by decreasing the length (Lg) of the shorting plate a pair of current distributions (#1 and #2) will be generated on both sides of the slot in the radiation element, and the bandwidth will be enhanced by the difference between the pair of current distributions (#1 and #2). Figure 12 shows the example of measured return loss characteristics when decreasing the length of the shorting plate. At the higher frequency band, a couple of resonant modes appeared near the conventional

5 AMANO et al: A QUARTER-WAVELENGTH SHORTED MICROSTRIP ANTENNA 1215 Fig. 13 Eletric fields when decreasing the length (Lg)of the shorting plate (simulated). Simulation model, First resonant mode (1.75 GHz), (c)second resonant mode (1.87 GHz). TM 30 mode resonant frequency. The first resonant mode is 1.75 GHz and the second resonant mode is 1.87 GHz. Then, a remarkable enhancement of bandwidth can be realized at the higher frequency band. The enhanced bandwidth, less than VSWR = 2, realized by the pair of resonant modes is about three times larger than the conventional bandwidth realized by the TM 30 mode. It was confirmed experimentally that this enhancement of bandwidth could be applied with the frequency ratio from about 2 to 3 by adjusting the length (Lg) of the shorting plate. On the other hand, at the lower frequency band, the bandwidth is only slightly increased compared with the conventional bandwidth. Figure 13 shows the simulation results of the electric field between the radiation element and the finite ground plane. Figure 13 shows the simulation model of the proposed antenna. Figure 13 shows the electric field of the first resonant mode (1.75 GHz) and Fig. 13(c) shows the electric field of the second resonant mode (1.87 GHz) at the higher frequency band. In these simulation results, it is confirmed that different electric fields have been generated between the first resonant mode and second resonant mode at the higher frequency band. As a result of this difference, the enhancement of bandwidth can be realized at the higher frequency band. 5. Conclusion also discussed how the enhancement of bandwidth is easily realized by decreasing the length of the shorting plate. This proposed antenna is therefore suitable for a small and low-profile dual-band internal antenna for cellular handsets. References [1] S.A. Long and M.D. Walton, A dual-frequency stacked circular-disk antenna, IEEE Trans. AP., vol.ap-27, pp , March [2] J.S. Dahele, Kai-Fong Lee, and D.P. Wong, Dual-frequency stacked annular-ring microstrip antenna, IEEE Trans. AP., vol.ap-35, no.11, pp , Nov [3] J. Fuhl, P. Nowak, and E. Bonek, Improved internal antenna for hand held terminals, Electron. Letts., vol.30, no.22, pp , Oct [4] K. Virga and Y. Rahmat-Samii, An enhanced bandwidth integral dual L antenna for mobile communication systems, IEEE AP-S Int. Symp. Digest, pp , June [5] Z.D. Liu and P.S. Hall, Dual band antenna for hand held portable telephones, Electron. Letts., vol.32, no.7, pp , March [6] K. Virga and Y. Rahmat-Samii, Low-profile enhancedbandwidth PIFA antennas for wireless communications, IEEE Trans. MTT., vol.45, no.10, pp , Oct [7] C.R. Rowell and R.D. Murch, A compact PIFA suitable for dual-frequency 900/1800-MHz operation, IEEE Trans. AP., vol.46, no.4, pp , April [8] S. Maci, G.B. Gentili, P. Piazzesi, and C. Salvador, Dualband slot-loaded patch antenna, IEE Proc. Microw. AP., vol.142, no.3, pp , June A novel dual-band internal antenna, which has a slot in the radiation element of a quarter-wavelength shorted microstrip antenna, has been proposed. Through experiments and simulations, the relationship between the length of the slot, location of the slot and the resonant frequency was clarified. It was confirmed that the resonant frequency is lowered significantly when the slot is located near the point of maximum current in the radiation element. Good dual-band operations were obtained experimentally with the frequency ratio (TM 30 mode/tm 10 mode) from 2 to 3. In addition, suitable radiation patterns for handsets were obtained. We have

6 1216 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY 1999 Takashi Amano was born in Tokyo, Japan, on February 6, He received B.E. and M.E. degrees in 1989 and 1991, respectively, from Sophia University, Tokyo, Japan. In 1991, he joined Toshiba Research and Development Center, Kawasaki, Japan, where he did research on microwave circuits and microwave antennas for personal communication systems. He is currently a Specialist at Wireless Communication Technology Center, Information and Communication Systems Laboratories, Hino, Japan, where he is engaged in research on microwave circuits, microwave antennas and RF packaging technologies for wireless communication terminals. terminals. Norimichi Chiba was born in Iwate, Japan, on January 23, He received B.S. in 1992 from Yamagata University, Yamagata, Japan. Since 1992, he has been with Wireless Communication Technology Center, Information and Communication Systems Laboratories, Toshiba Corporation, Hino, Japan, where he is engaged in research on microwave circuits, microwave antennas and RF packaging technologies for wireless communication Hisao Iwasaki received the B.E. and D.E. degrees in electrical engineering from Tohoku University, Sendai, Japan, in 1975 and 1994, respectively. In 1975, he jointed the Toshiba Research and Development Center, Kawasaki, Japan, where he did research on antennas for communications. In 1986, he joined ATR Optical and Radio Communications Research Laboratories, Kyoto, Japan, where he was engaged in research on active array antennas. From 1989 to 1997, he worked at the Communication and Information Systems Research Labs. Toshiba Research and Development Center, Kawasaki, Japan, where he has been engaged in the development of the micro-strip antennas for communications and was a Senior Research Scientist there. He is currently a Senior Research Scientist at Wireless Communication Technology Center Toshiba Information and Communication Technology Systems Laboratories, Hino, Japan, where he has been engaged in the development of the microstrip antennas for communications and the antennas for handsets. Dr. Iwasaki received the Young Engineers Award from IEICE Japan in Dr. Iwasaki was Secretary and Treasurer of the IEEE AP-S Tokyo Chapter, from 1996 to 1997.