AAT1185 PRODUCT DATASHEET. High Voltage Step-Down Controller. General Description. Features. Applications. Typical Application AAT1185

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1 SwitchReg TM General Description The is a single output step-down (Buck) regulator controller with an input range of 6V to 24V. The output range is adjustable from.8v to 5.5V. The device provides high and low-side pins to drive external n-channel MOSFETs; allowing fully synchronous operation for maximum efficiency and performance. Alternately, the low-side MOSFET may be replaced with a Schottky rectifier. Both high and low-side drive pins are compatible with a wide range of external MOSFETs making the device the ideal control solution for low power and high power configurations. Voltage mode control allows for optimum performance across the entire output voltage and load range. The 49kHz fixed switching frequency allows wide range of L/C filtering components, achieving smallest size and maximum efficiency. External compensation allows the designer to optimize the transient response. The controller includes programmable over-current, integrated soft-start and over-temperature protection. The is available in the Pb-free, 4-pin TSOPJW package. The rated operating temperature range is -4 C to 85 C. Features V IN = 6.V to 24.V V OUT Adjustable from.8v to 5.5V I OUT from <A up to A Small Solution Size Ultra-small External L/C Synchronous or Non-Synchronous Shutdown Current <3μA High Switching Frequency Voltage Mode Control PWM Fixed Frequency for Lowest Noise Programmable Over-Current Protection Over-Temperature Protection Internal Soft Start 2.85x3mm TSOPJW-4 Package -4 C to 85 C Temperature Range Applications DSL and Cable Modems Notebook Computers Satellite Set Top Boxes Wireless LAN Systems Typical Application D BAS6 U BST C8.μF VIN 6V - 24V C7 2.2μF VCC DH HV LX R 3.32 Q L 3.9μH VOUT 3.3V/A C3 47μF 25V C5 μf 25V EN PGND DL RS OS R Q2 R3.74K C9.47μF R5 K C2 68pF R6 27.4K C3, C4 2 47μF COMP FB TSOPJW-4 R4 2K C 33pF C 68pF R7 6.4k

2 Pin Descriptions Pin # Symbol Function Description RS I Output sense voltage pin. Connect to the output capacitor to enable over-current sense for step-down converter. 2 OS I Output current sense pin. Connect a small signal resistor from this pin to small signal resistor which is tied to switching node (LX) to enable over-current sense for stepdown converter. The current limit threshold varies with inductor parasitic winding resistance (R DC(L) ); see the Applications Information section of this datasheet for details. 3 EN I Step-down regulator enable input pin. Active high or tied to high voltage input (IN) enables internal linear regulator and output. 4 BST I Step-down regulator boost drive input pin. Connect the cathode of fast rectifier from this pin and connect a nf capacitor from this pin to the switching node (LX) to provide drive to external hi-side MOSFET gate. 5 DH O High side driver for external high side n-channel MOSFET. Connect this pin to gate of external high side n-channel MOSFET device. 6 LX O Step-down converter switching pin. Connect output inductor to this pin. 7 PGND GND Power ground pin for step-down regulator. When using synchronous option, tie to PCB ground plane near source pins of external low-side MOSFET(s). 8 DL O Low side driver for external low side n-channel MOSFET. When using synchronous option, connect this pin to gate of external low side n-channel MOSFET device. Otherwise, leave pin open. 9, VL I/O Internal linear regulator for step-down converter. Connect a 2.2μF/6.3V capacitor from this pin to GND. IN I High voltage input pin. 2 GND GND Ground pin for step-down regulator. Tie to PCB ground plane. 3 FB I Feedback input pin for step-down converter. Connect an external resistor divider to this pin to program the output voltage to the desired value. 4 COMP I Compensation pin for step-down converter. Connect a resistor, capacitor network to compensate the voltage mode control loop. Pin Configuration TSOPJW-4 (Top View) RS OS EN BST DH LX PGND COMP FB GND VL IN VL DL

3 Absolute Maximum Ratings T A = 25 O C unless otherwise noted. Symbol Description Value Units V IN(HI), V EN IN, LX, EN to GND -.3 to 3. V V IN(LO) VL to GND -.3 to 6. V V BST-LX BST to LX -.3 to 6. V V CONTROL DH, DL, FB, COMP, RS, OS to PGND, GND -.3 to V IN(LO) +.3 V T J Operating Junction Temperature Range -4 to 5 C T LEAD Maximum Soldering Temperature (at leads, sec) 3 C Thermal Information 2 Symbol Description Value Units Θ JA Thermal Resistance 3 4 O C/W P D Maximum Power Dissipation.7 W. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at conditions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time. 2. Mounted on an FR4 board. 3. Derate 7mW/ C above 25 C

4 Electrical Characteristics V IN = 2.V; T A = -4 C to 85 C unless otherwise noted. Typical values are at T A = 25 C. Symbol Description Conditions Min Typ Max Units V IN Input Voltage V V IN Rising 5. V V UVLO UVLO Threshold V IN Hysteresis 3 mv V IN Falling 3. V V OUT Output Voltage Range V V FB Feedback Pin Voltage V I Q Quiescent Current V EN = High, No load. ma I SHDN Shutdown Current V EN = Low, V L = V 3 μa V OCP Over-Current Offset Voltage V EN = High, V IN = 6.V to 24.V, T A = 25 C 7 3 mv I LX LX Pin Leakage Current V IN = 24.V, V EN = Low -.. μa D MAX Maximum Duty Cycle 85 % T ON(MIN) Minimum On-Time V IN = 6.V to 24.V ns R DH High Side Drive Source Resistance Pull-Up 5. Pull-Down.7 Ω R DL Low Side Drive Source Resistance Pull-Up 5. Pull-Down.7 Ω F OSC Oscillator Frequency khz F FOLDBACK Short Circuit Foldback Frequency Current Limit Triggered khz T S Start-Up Time From Enable to Output Regulation 2.5 ms Over-Temperature Shutdown Threshold 35 C T SD Over-Temperature Shutdown Hysteresis 5 C V EN(L) Enable Threshold Low.6 V V EN(H) Enable Threshold High 2.5 V I EN Input Low Current -.. μa. The is guaranteed to meet performance specifications over the 4 C to +85 C operating temperature range and is assured by design, characterization and correlation with statistical process controls

5 Typical Characteristics Circuit of Figure 4, unless otherwise specified. Efficiency (%) Step-Down Controller Efficiency vs. Load (V OUT = 3.3V; L = 3.9µH) V IN = 6V 3 VIN = 8V 2 V IN = 2V V IN = 8V VIN = 24V. Output Current (ma) Output Error (%) Step-Down Controller DC Regulation (V OUT = 3.3V; L = 3.9µH) -.5 V IN = 6V V IN = 8V -. V IN = 2V -.5 V IN = 8V V IN = 24V -2.. Output Current (ma) Accuracy (%) Step-Down Controller Line Regulation (V OUT = 3.3V; L = 3.9µH) Input Voltage (V) I OUT =.ma I OUT = ma I OUT = A I OUT = 5A I OUT = 8A I OUT = A Output Voltage Error (%) Step-Down Controller Output Voltage Error vs. Temperature (V IN = 2V; V OUT = 3.3V) Temperature ( C) I OUT =.ma IOUT = ma IOUT = A IOUT = 5A I OUT = 8A IOUT = A Step-Down Controller Output Ripple (V IN = 2V; V OUT = 3.3V; I OUT = ma) Step-Down Controller Output Ripple (V IN = 2V; V OUT = 3.3V; I OUT = A) Output Voltage (middle) (V) V 2V - LX Voltage (top) (V) Inductor Current (bottom) (A) Output Voltage (middle) (V) V 2V 2 8 LX Voltage (top) (V) Inductor Current (bottom) (A) Time (µs/div) Time (µs/div)

6 Typical Characteristics Circuit of Figure 4, unless otherwise specified. Step-Down Controller Load Transient Response (V IN = 2V; I OUT = A to A; C OUT = 2x47µF) Output Voltage (bottom) (V) A A 5 5 Output Current (top) (A) Step-Down Controller Load Transient Response (V IN = 2V; I OUT = 5A to A; C OUT = 2x47µF) Output Voltage (bottom) (V) A A 5 5 Output Current (top) (A) Time (µs/div) Time (µs/div) Step-Down Controller Load Transient Response (V IN = 2V; I OUT = 7.5A to A; C OUT = 2x47µF) Step-Down Controller Soft Start (V IN = 2V; V OUT = 3.3V; I OUT = A) Output Voltage (bottom) (V) A A Output Current (top) (A) Enable Voltage (top) (V) Output Voltage (middle) (V) Inductor Current (bottom) (A) Time (µs/div) Time (5µs/div) Step-Down Controller Line Transient Response (V IN = 8V to 2V; V OUT = 3.3V; I OUT = 5A) Step-Down Controller Line Transient Response (V IN = 8V to 2V; V OUT = 3.3V; I OUT = A) Input Voltage (top) (V) Output Voltage (bottom) (V) Input Voltage (top) (V) Output Voltage (bottom) (V) Time (2µs/div) Time (2µs/div)

7 Typical Characteristics Circuit of Figure 4, unless otherwise specified. Frequency Variation (%) Step-Down Controller Switching Frequency vs. Input Voltage (V OUT = 3.3V; I OUT = A) Input Voltage (V) Switching Frequency (khz) Step-Down Controller Switching Frequency vs. Temperature (V IN = 2V; V OUT = 3.3V; I OUT = A) Temperature ( C) Input Current (ma) No Load Step-Down Controller Input Current vs. Input Voltage (V EN = V IN ).4 85 C C -4 C Output Voltage (top) (V) 4 2 Step-Down Controller Current Limit (V IN = 2V; V OUT = 3.3V; L = 3.9µH) 2V 2 LX Voltage (middle) (V) Inductor Current (bottom) (A) Input Voltage (V) Time (4µs/div)

8 Functional Block Diagram VL VINT Reg. IN OT OSC FB Error Amp Comp. BST COMP Logic DH EN Voltage Ref Control Logic LX DL PGND EN Comp RS OS V OCP =.V Applications Information The is a single output step-down (Buck) regulator controller with an input range of 6V to 24V. The output range is adjustable from.8v to 5.5V. The device provides high and low-side pins to drive external n-channel MOSFETs; allowing fully synchronous operation for maximum efficiency and performance. Alternatively, the low-side MOSFET may be replaced with a Schottky rectifier and the DL pin left open. Both high and low-side drive pins are compatible with a wide range of external MOSFETs making the device the ideal control solution for low power and high power configurations. Voltage mode control allows for optimum performance across the entire output voltage and load range. 49kHz fixed switching frequency allows wide range of L/C filtering components, achieving smallest size and maximum efficiency. External compensation allows the designer to optimize the transient response components. The controller includes programmable over-current, integrated soft-start and over-temperature protection. The is available in the Pb-free, 4-pin TSOPJW package. The rated operating temperature range is -4 C to 85 C. Regulator Output Capacitor Selection Two 47μF ceramic output capacitors are required to filter the inductor current ripple and supply the load transient current for I OUT = A. The 2 package with V minimum voltage rating is recommended for the output capacitors to maintain a minimum capacitance drop with DC bias. Output Inductor Selection The step-down converter utilizes constant frequency (PWM-mode) voltage mode control. A 3.9μH to 4.7μH inductor value with appropriate DCR is selected to maintain the desired output current ripple and minimize the

9 converter s response time to load transients. The peak switch current should not exceed the inductor saturation current of the MOSFETs. The DCR of the inductor sets the designed current limit in the following formula: I LIM = mv DCR For A output load, the selected DCR should be less than mω to avoid the peak inductor current triggers the current limit. MOSFET Selection The step-down (buck) converter utilizes synchronous rectification (Q) for constant frequency (PWM mode) voltage mode control. The synchronous rectifier is selected based on the desired R DS(ON) value and Q G (total gate charge), these two critical parameters are weighed against each other. To get a low R DS(ON) value, the MOSFET must be very large; a larger MOSFET will have a large Q G. Conversely, to get a low Q G, the MOSFET must be small and thus have a large R DS(ON) value. In addition to the trade off between R DS(ON) and Q G, the maximum voltage rating for the external synchronous MOSFET must exceed the maximum application input voltage value (V DS [max] > V IN [max]). The Q G affects the turn-on/turn-off time of the synchronous MOSFET; the longer the turn-on/turn-off time, the more likely the step-down converter will have shootthrough current issues. Shoot-through current occurs when the high-side MOSFET and the low-side MOSFET are conducting current at the same time. This will result in a low impedance path to ground from the input voltage through the two MOSFETs, and the current may exceed the maximum current rating of the MOSFETs. Exceeding the maximum current ratings will lead to the destructive derating of the MOSFETs. The critical parameter recommendations for the external minimum 25V MOSFET are as follows: Q G (Total Gate Charge): 5nC to 5nC (max) (V GS : 4.5V to 5V) R DS(ON) : mω to 3mΩ (max) (V GS : 4.5V to 5V) Input Capacitor Selection For low-cost applications, a 47μF/25V electrolytic capacitor is selected to control the voltage overshoot across the high side MOSFET. A μf/25v ceramic capacitor with a voltage rating at least.5 times greater than the maximum input voltage is connected as close as possible to the input pins (Pins 9 and ) for high frequency decoupling. Feedback and Compensation Networks COMP C C R4 REF Figure : Feedback and Compensation Networks for Type III Voltage-Mode Control Loop. The transfer function of the error amplifier is dominated by DC gain and the L C OUT output filter of the regulator. This output filter and its equivalent series resistance (ESR) create a double pole at F LC and a zero at F ESR in the following equations: Eq. : F LC = 2 π L C OUT FB C2 R7 R6 R5 Eq. 2: F ESR = 2 π ESR C OUT The feedback and compensation networks provide a closed loop transfer function with the highest db crossing frequency and adequate phase margin for system stability. Equations 3, 4, 5 and 6 relate the compensation network s poles and zeros to the components R 4, R 5, R 6, C, C, and C 2 : Eq. 3: F Z = 2 π R 4 C Eq. 4: F Z2 = 2 π (R 5 + R 6 ) C 2 Eq. 5: F P = 2 π R 4 Eq. 6: F P2 = C C C + C 2 π R 5 C 2 Components of the feedback, feed-forward, and compensation networks need to be adjusted to maintain the system's stability for different input and output voltages applications as shown in Table. VOUT

10 Network Feedback Feed-forward Compensation Components V OUT =3.3V V IN = 6V-24V V OUT = 5.V V IN = 6V-24V R kΩ.96kΩ R 7 6.4kΩ 4.3kΩ C 2 68pF 2.2nF R 5 kω 453Ω C 33pF 2.2nF C 68pF 5pF R 4 2kΩ 3.92kΩ Table : Feedback and Compensation Components for V OUT =3.3V and V OUT = 5.V. Over-Current Protection The controller provides true-load DC output current sensing which protects the load and limits component stresses. The output current is sensed through the DC resistance in the output inductor (DCR). The controller reduces the operating frequency when an over-current condition is detected; limiting stresses and preventing inductor saturation. This allows the smallest possible inductor for a given output load. A small resistor divider may be necessary to adjust the over-current threshold and compensate for variation in inductor DCR. The preset current limit threshold is triggered when the differential voltage from RS to OS exceeds mv (nominal). LX RS OS R3.74k L 3.9μH C9.47μF R8 V OUT 5.V/A R9 LX L 3.9μH V OUT 5V/A Figure 3: Resistor Network to Adjust the Current Limit Greater than the Pre-Set Over-Current Threshold (Add R8, R9). RS OS R3.74k C9.47μF R9 R L (μh) R3 (kω) C9 (μf) Part Number B82559A392A3, 3.9μH, Epcos, I SAT = 2A, DCR = 4.8mΩ RLF256T-4R2N, 4.2μH, TDK, I SAT =.2A, DCR = 7.4mΩ SER23-472ML, 4.7μH, Coilcraft, I SAT = 8A, DCR =.7mΩ Table 2: Current Limit Network vs. Inductor DCR. Figure 2: Resistor Network to Adjust the Current Limit Less than the Pre-Set Over-Current Threshold (Add R9, R). Thermal Protection The has an internal thermal protection circuit which will turn on when the device die temperature exceeds 35 C. The internal thermal protection circuit will actively turn off the high side regulator output device to prevent the possibility of over temperature damage

11 The Buck regulator output will remain in a shutdown state until the internal die temperature falls back below the 35 C trip point. The combination and interaction between the short circuit and thermal protection systems allows the Buck regulator to withstand indefinite short-circuit conditions without sustaining permanent damage. Thermal Calculations There are three types of losses associated with the step-down converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the R DS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified form of the synchronous step-down converter and LDO losses is given by: I OUT2 (R DS(ON)H V OUT + R DS(ON)L [V IN - V OUT ]) P TOTAL = + (t SW F S I OUT + I Q ) V IN I Q is the step-down converter quiescent currents. The term t SW is used to estimate the full load step-down converter switching losses. The power dissipation that relates to the R DS(ON) occurs in the external high side and low side MOSFETs. Therefore, the total package losses for reduce to the following equation: V IN P TOTAL = (t SW F S I OUT + I Q ) V IN Since quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. Given the total losses, the maximum junction temperature can be derived from the θ JA for the TSOPJW-4 package, which is 4 C/W. Layout Considerations The suggested PCB layout for the is shown in Figures 5, 6, 7, and 8. The following guidelines should be used to help ensure a proper layout.. The power input capacitors (C3 and C5) should be connected as closely as possible to the high voltage input pin (IN) and power ground. 2. C5, L, Q, C3, and C4 should be placed as closely as possible to each other to minimize any parasitic inductance in the switched current path, which generates a large voltage spike during the switching interval. The connection of inductor to switching node should be as short as possible. 3. The feedback trace or FB pin should be separated from any power trace and connected as closely as possible to the load point. Sensing along a highcurrent load trace will degrade DC load regulation. 4. The resistance of the trace from the load returns to PGND should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal signal ground and the power ground. 5. Connect unused signal pins to ground to avoid unwanted noise coupling. 6. The critical small signal components, include feedback components and compensation components, should be placed close to the FB and COMP pins. The feedback resistors should be located as close as possible to the FB pin with its ground tied straight to the signal ground plane, which is separated from the power ground plane. 7. C9 and R3 should be connected as closely as possible to the RS and OS pins and placed on the bottom side of the layout to avoid noise coupling from the inductor. 8. For good thermal coupling, a 4-layer PCB layout is recommended and PCB vias are required from the exposed pad (EP) for the MOSFETs paddle to the middle plane and bottom plane. T J(MAX) = P TOTAL θ JA + T AMB

12 VIN 6V - 24V C, C2, C4, C6 open C3 47μF 25V C7 2.2μF C5 μf 25V D BAS6 EN U VCC BST VCC HV EN DL 4 DH 5 LX 6 8 GND RS PGND OS 2 R 3.32 R C8.μF Q Si7326DN Q2 Si7326DN L 3.9μH R3.74K C9.47μF R5 k C2 68pF R6 27.4K V OUT 3.3V/A C3, C4 2x47μF C5, C6 open 4 COMP FB 3 TSOPJW-4 R4 2K C 33pF C 68pF R7 6.4k U Analogic Technologies, Hi-Voltage Buck Controller, TSOPJW-4 C3 Cap, MLC, 47μF/25V, Electrolytic C5 Cap, MLC, μf/25v, 2 C7 Cap, MLC, 2.2μF/6.3V, 63 C8 Cap, MLC,.μF/6.3V, 63 C9 Cap, MLC,.47μF/6.3V, 63 C, C, C2 Cap, MLC, misc, 42 C3, C4 Cap, MLC, 47μF/V, 2 R-R7 Carbon film resistor, 42 D BAS6, Generic, Rectifier,.2A/85V, Ultrafast, SOT23 Q, Q2 Si7326DN, Vishay, N-Channel, 3V, A, PAK 22-8 L B82559A392A3, 3.9μH, Epcos, I SAT = 2A, DCR = 4.8mΩ L RLF256T-4R2N, 4.2μH, TDK, I SAT =.2A, DCR = 7.4mΩ L SER23-472ML, 4.7μH, Coicraft, ISAT = 8A, DCR =.7mΩ Figure 4: ITO Evaluation Board Schematic for V IN = 6V-24V and V OUT = 3.3V

13 Figure 5: ITO Evaluation Board Top Layer. Figure 6: ITO Evaluation Board MID Layer. Figure 7: ITO Evaluation Board MID2 Layer. Figure 8: ITO Evaluation Board Bottom Layer

14 Design Example Specifications V O = A, Pulsed Load ΔI LOAD = A V IN = 2V F S = 49kHz T AMB = 85 C in TSOPJW-4 Package Output Inductor For Epcos inductor B82559A392A3, 3.9μH, DCR = 4.8mΩ max. V OUT V OUT 3.3V 3.3V ΔI = - = - =.25A L F S V IN 3.9μH 49kHz 2V ΔI I PK = I OUT + = A +.6A =.6A 2 P L = I OUT 2 DCR =.6A 2 4.8mΩ = 539mW Output Capacitor V DROOP =.6V 3 ΔI LOAD 3 A C OUT = = = 2μF; use 2x47μF V DROOP F S.6V 49kHz I RMS(MAX) = 2 3 V OUT (V IN(MAX) - V OUT ) 3.3V (24V - 3.3V) = = 43mA RMS L F S V IN(MAX) μH 49kHz 24V P RMS = ESR I RMS 2 = 5mΩ (43mA) 2 =.9mW Input Capacitor Input Ripple V PP = 6mV C IN = = = 5μF V PP 6mV - ESR 4 F I S - 5mΩ 4 49kHz OUT A For low cost applications, a 47μF/25V electrolytic capacitor in parallel with a μf/25v ceramic capacitor is used to reduce the ESR. I RMS I OUT = = 5A 2 P = ESR (I RMS ) 2 = 5mΩ (5A) 2 = 25mW

15 Current Limit Over-current offset voltage V OCP = mv Total trace parasitic resistor and inductor DCR is 6mΩ V mv I LIMIT = S = = 7A DCR 6mΩ In order to sense the inductor current correctly during dynamic operation the R-C network time constant R3 *C9 should match the inductor time constant L/DCR: L DCR = R 3 C 9 Choose C3 =.47μF R 3 = L DCR C 9 3.9μH = =.74kΩ 4.8mΩ.47μF Losses All values assume 25 C ambient temperature and thermal resistance of 4 C/W in the TSOPJW-2 package. P TOTAL = (t SW F S I OUT + I Q ) V IN P TOTAL = (5ns 49kHz A + 7μA) 2V P TOTAL = 295mW T J(MAX) = T AMB + Θ JA P LOSS = 85 C + (4 C/W).295mW = 26.3 C

16 Ordering Information Package Voltage Marking Part Number (Tape and Reel) 2 TSOPJW-4 Adj (.6V) 4UXYY ITO-.6-T All AnalogicTech products are offered in Pb-free packaging. The term Pb-free means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed.% by weight in homogeneous materials. For more information, please visit our website at Package Information TSOPJW ± ±.2.4 BSC Top View REF ± ± ± ±.5 Side View End View All dimensions in millimeters.. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD

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