Dual FET-Input, Low Distortion OPERATIONAL AMPLIFIER

Dual FET-Input, Low Distortion OPERATIONAL AMPLIFIER FEATURES LOW DISTORTION:.3% at khz LOW NOISE: nv/ Hz HIGH SLEW RATE: 2V/µs WIDE GAIN-BANDWIDTH: 2MHz UNITY-GAIN STABLE WIDE SUPPLY RANGE: V S = ±4. to ±24V DRIVES 6Ω LOADS APPLICATIONS PROFESSIONAL AUDIO EQUIPMENT PCM DAC I/V CONVERTER SPECTRAL ANALYSIS EQUIPMENT ACTIVE FILTERS TRANSDUCER AMPLIFIER DATA ACQUISITION DESCRIPTION (8) V+ The is a dual, FET-input operational amplifier designed for enhanced AC performance. Very low distortion, low noise and wide bandwidth provide superior performance in high quality audio and other applications requiring excellent dynamic performance. New circuit techniques and special laser trimming of dynamic circuit performance yield very low harmonic distortion. The result is an op amp with exceptional sound quality. The low-noise FET input of the provides wide dynamic range, even with high source impedance. Offset voltage is laser-trimmed to minimize the need for interstage coupling capacitors. The is available in 8-pin plastic mini-dip and SO-8 surface-mount packages, specified for the 2 C to +8 C temperature range. (+) (3, ) ( ) (2, 6) Distortion Rejection Circuitry* * Patents Granted: #378, 9789 Output Stage* (4) V (, 7) International Airport Industrial Park Mailing Address: PO Box 4 Tucson, AZ 8734 Street Address: 673 S. Tucson Blvd. Tucson, AZ 876 Tel: (2) 746- Twx: 9-92- Cable: BBRCORP Telex: 66-649 FAX: (2) 889- Immediate Product Info: (8) 48-632 99 Burr-Brown Corporation PDS-69D Printed in U.S.A. May, 99

SPECIFICATIONS ELECTRICAL T A = +2 C, V S = ±V unless otherwise noted. AP, AU PARAMETEONDITION MIN TYP MAX UNITS OFFSET VOLTAGE Input Offset Voltage ± ± mv Average Drift ±8 µv/ C Power Supply Rejection V S = ± to ±24V 8 db INPUT BIAS CURRENT () Input Bias Current V CM = V pa Input Offset Current V CM = V ±4 pa NOISE Input Voltage Noise Noise Density: f = Hz 2 nv/ Hz f = Hz nv/ Hz f = khz nv/ Hz f = khz nv/ Hz Voltage Noise, BW = 2Hz to 2kHz. µvp-p Input Bias Current Noise Current Noise Density, f =.Hz to 2kHz 6 fa/ Hz INPUT VOLTAGE RANGE Common-Mode Input Range ±2 ±3 V Common-Mode Rejection V CM = ±2V 8 db INPUT IMPEDANCE Differential 2 8 Ω pf Common-Mode 2 Ω pf OPEN-LOOP GAIN Open-Loop Voltage Gain = ±V, R L = kω 8 db FREQUENCY RESPONSE Gain-Bandwidth Product G = 2 MHz Slew Rate 2Vp-p, R L = kω 2 V/µs Settling Time:.% G =, V Step. µs.% µs Total Harmonic Distortion + Noise (THD+N) G =, f = khz.3 % = 3.Vrms, R L = kω Channel Separation f = khz, R L = kω 42 db OUTPUT Voltage Output R L = 6Ω ± ±2 V Current Output = ±2V ±3 ma Short Circuit Current ±4 ma Output Resistance, Open-Loop 2 W POWER SUPPLY Specified Operating Voltage ± V Operating Voltage Range ±4. ±24 V Current, Total Both Amplifiers ±. ±2 ma TEMPERATURE RANGE Specification 2 +8 C Storage 4 +2 C Thermal Resistance (2), θ JA 9 C/W NOTES: () Typical performance, measured fully warmed-up. (2) Soldered to circuit board see text. ABSOLUTE MAXIMUM RATINGS Power Supply Voltage... ±2V Input Voltage... (V ) V to (V+)+V Output Short Circuit to Ground... Continuous Operating Temperature... 4 C to + C Storage Temperature... 4 C to +2 C Junction Temperature... + C Lead Temperature (soldering, s) AP... +3 C Lead Temperature (soldering, 3s) AU... +26 C PACKAGING INFORMATION PACKAGE DRAWING MODEL PACKAGE NUMBER () AP 8-Pin Plastic DIP 6 AU SO-8 Surface-Mount 82 NOTE: () For detailed drawing and dimension table, please see end of data sheet, or Appendix D of Burr-Brown IC Data Book. ORDERING INFORMATION MODEL PACKAGE TEMP. RANGE AP 8-Pin Plastic DIP 2 C to +8 C AU SO-8 Surface-Mount 2 C to +8 C 2

PIN CONFIGURATION Top View DIP/SOIC ELECTROSTATIC DISCHARGE SENSITIVITY Output A In A +In A V 2 3 4 8 7 6 V+ Output B In B +In B Any integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. TYPICAL PERFORMANCE CURVES T A = +2 C, V S = ±V unless otherwise noted. THD + N (%)... TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY = 3.Vrms kω G = V/V G = V/V Measurement BW = 8kHz See Distortion Measure- ments for description of test method. THD + N (%)... TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT VOLTAGE See Distortion Measurements for description of test method. kω f = khz Measurement BW = 8kHz G = V/V. 2 k k 2k Frequency (Hz).. Output Voltage (Vp-p) Voltage Gain (db) 2 8 6 4 2 OPEN-LOOP GAIN/PHASE vs FREQUENCY G φ 4 9 3 8 Phase Shift (Degrees) Voltage Noise (nv/ Hz) k INPUT VOLTAGE AND CURRENT NOISE SPECTRAL DENSITY vs FREQUENCY Voltage Noise Current Noise k Current Noise (fa/ Hz) 2 k k k M M Frequency (Hz) k k k M Frequency (Hz) 3

TYPICAL PERFORMANCE CURVES (CONT) T A = +2 C, V S = ±V unless otherwise noted. na INPUT BIAS AND INPUT OFFSET CURRENT vs TEMPERATURE na na INPUT BIAS AND INPUT OFFSET CURRENT vs INPUT COMMON-MODE VOLTAGE na Input Bias Current (pa) na na 7 Input Bias Current Input Offset Current na. 2 2 7 2 Input Offset Current (pa) Input Bias Current (pa) na Input Bias Current Input Offset Current Input Offset Current (pa) Ambient Temperature ( C) Common-Mode Voltage (V) na INPUT BIAS CURRENT vs TIME FROM POWER TURN-ON 2 COMMON-MODE REJECTION vs COMMON-MODE VOLTAGE Input Bias Current (pa) V S = ±24VDC V S = ±VDC V S = ±VDC Common-Mode Rejection (db) 9 3 4 Time After Power Turn-On (min) 8 Common-Mode Voltage (V) 2 POWER SUPPLY AND COMMON-MODE REJECTION vs FREQUENCY +PSR 2 2 A OL, PSR, AND CMR vs SUPPLY VOLTAGE Power Supply Rejection (db) 8 6 4 2 PSR CMR 8 6 4 2 Common-Mode Rejection (db) A OL, PSR, CMR (db) 9 8 CMR PSR A OL k k k M M 7 2 2 Frequency (Hz) Supply Voltage (±V S ) 4

TYPICAL PERFORMANCE CURVES (CONT) T A = +2 C, V S = ±V unless otherwise noted. 28 GAIN-BANDWIDTH AND SLEW RATE vs SUPPLY VOLTAGE 33 28 GAIN-BANDWIDTH AND SLEW RATE vs TEMPERATURE 3 Slew Rate Gain-Bandwidth (MHz) 24 2 6 Gain-Bandwidth G = + Slew Rate 29 2 2 Slew Rate (V/µs) Gain-Bandwidth (MHz) 24 2 6 Gain-Bandwidth G = + 2 2 Slew Rate (V/µs) 2 7 2 2 Supply Voltage (±V S ) 2 7 2 2 7 2 Temperature ( C) Settling Time (µs) 4 3 2 SETTLING TIME vs CLOSED-LOOP GAIN = V Step R L = kω = pf.%.% Closed-Loop Gain (V/V) Channel Separation (db) 6 4 2 8 CHANNEL SEPARATION vs FREQUENCY A = 2Vp-p R L R L = kω B k k k Frequency (Hz) Measured Output R L = 3 MAXIMUM OUTPUT VOLTAGE SWING vs FREQUENCY 4 SUPPLY CURRENT vs TEMPERATURE V = ±V S Total for Both Op Amps Output Voltage (Vp-p) 2 Supply Current (ma) 2 8 V S = ±VDC V S = ±24VDC V S = ±VDC k k M M 6 7 2 2 7 2 Frequency (Hz) Ambient Temperature ( C)

TYPICAL PERFORMANCE CURVES (CONT) T A = +2 C, V S = ±V unless otherwise noted. LARGE-SIGNAL TRANSIENT RESPONSE SMALL-SIGNAL TRANSIENT RESPONSE Output Voltage (V) + FPO Bleed to edge Output Voltage (mv) + Time (µs) µs 2µs Time (µs) Short-Circuit Current (ma) 6 4 3 SHORT-CIRCUIT CURRENT vs TEMPERATURE I SC+ and I SC Power Dissipation (W).9.8.7.6..4.3.2 POWER DISSIPATION vs SUPPLY VOLTAGE Worst case sine wave R L = 6Ω (both channels) Typical high-level music R L = 6Ω (both channels) No signal or no load 2 7 2 2 7 2 Ambient Temperature ( C). 6 8 2 4 6 8 2 22 24 Supply Voltage, ±V S (V) Total Power Dissipation (W).4.2..8.6.4.2 MAXIMUM POWER DISSIPATION vs TEMPERATURE Maximum Specified Operating Temperature 8 C θj-a = 9 C/W Soldered to Circuit Board (see text) 2 7 2 Ambient Temperature ( C) 6

APPLICATIONS INFORMATION The is unity-gain stable, making it easy to use in a wide range of circuitry. Applications with noisy or high impedance power supply lines may require decoupling capacitors close to the device pins. In most cases µf tantalum capacitors are adequate. DISTORTION MEASUREMENTS The distortion produced by the is below the measurement limit of virtually all commercially available equipment. A special test circuit, however, can be used to extend the measurement capabilities. Op amp distortion can be considered an internal error source which can be referred to the input. Figure shows a circuit which causes the op amp distortion to be times greater than normally produced by the op amp. The addition of R 3 to the otherwise standard non-inverting amplifier configuration alters the feedback factor or noise gain of the circuit. The closed-loop gain is unchanged, but the feedback available for error correction is reduced by a factor of. This extends the measurement limit, including the effects of the signal-source purity, by a factor of. Note that the input signal and load applied to the op amp are the same as with conventional feedback without R 3. Validity of this technique can be verified by duplicating measurements at high gain and/or high frequency where the distortion is within the measurement capability of the test equipment. Measurements for this data sheet were made with the Audio Precision System One which greatly simplifies such repetitive measurements. The measurement technique can, however, be performed with manual distortion measurement instruments. CAPACITIVE LOADS The dynamic characteristics of the have been optimized for commonly encountered gains, loads and operating conditions. The combination of low closed-loop gain and capacitive load will decrease the phase margin and may lead to gain peaking or oscillations. Load capacitance reacts with the op amp s open-loop output resistance to form an additional pole in the feedback loop. Figure 2 shows various circuits which preserve phase margin with capacitive load. Request Application Bulletin AB-28 for details of analysis techniques and applications circuits. For the unity-gain buffer, Figure 2a, stability is preserved by adding a phase-lead network, and C C. Voltage drop across will reduce output voltage swing with heavy loads. An alternate circuit, Figure 2b, does not limit the output with low load impedance. It provides a small amount of positive feedback to reduce the net feedback factor. Input impedance of this circuit falls at high frequency as op amp gain rolloff reduces the bootstrap action on the compensation network. Figures 2c and 2d show compensation techniques for noninverting amplifiers. Like the follower circuits, the circuit in Figure 2d eliminates voltage drop due to load current, but at the penalty of somewhat reduced input impedance at high frequency. Figures 2e and 2f show input lead compensation networks for inverting and difference amplifier configurations. NOISE PERFORMANCE Op amp noise is described by two parameters noise voltage and noise current. The voltage noise determines the noise performance with low source impedance. Low noise bipolarinput op amps such as the OPA27 and OPA37 provide very low voltage noise. But if source impedance is greater than a few thousand ohms, the current noise of bipolar-input op amps react with the source impedance and will dominate. At a few thousand ohms source impedance and above, the will generally provide lower noise. R SIG. GAIN DIST. GAIN R R 3 R 3 = Vp-p (3.Vrms) Ω Ω kω kω kω Ω Ω Generator Output Analyzer Input Audio Precision System One Analyzer* R L kω IBM PC or Compatible * Measurement BW = 8kHz FIGURE. Distortion Test Circuit. 7

(a) (b) C C e i C C = 2 X 2 82pF 7Ω e o pf e i C C.47µF Ω e o pf = 4 X C C = X 3 (c) (d) R kω kω R C C 24pF 2Ω e i 2Ω e o e i C C.22µF e o C C = pf = 2 X ( + /R ) pf C C = X 3 (e) (f) e R e i R 2Ω C C.22µF e o pf = 2 X ( + /R ) C C = X 3 e 2 2Ω C C.22µF R 3 R 4 = 2 X ( + /R ) C C = X 3 e o pf NOTE: Design equations and component values are approximate. User adjustment is required for optimum performance. FIGURE 2. Driving Large Capacitive Loads. 8

POWER DISSIPATION The is capable of driving 6Ω loads with power supply voltages up to ±24V. Internal power dissipation is increased when operating at high power supply voltage. The typical performance curve, Power Dissipation vs Power Supply Voltage, shows quiescent dissipation (no signal or no load) as well as dissipation with a worst case continuous sine wave. Continuous high-level music signals typically produce dissipation significantly less than worst case sine waves. Copper leadframe construction used in the improves heat dissipation compared to conventional plastic packages. To achieve best heat dissipation, solder the device directly to the circuit board and use wide circuit board traces. OUTPUT CURRENT LIMIT Output current is limited by internal circuitry to approximately ±4mA at 2 C. The limit current decreases with increasing temperature as shown in the typical curves. R 4 V IN R 2.7kΩ 2 C 3pF R 3 kω C 2 2pF 2 C 3 pf f p = 2kHz FIGURE 3. Three-Pole Low-Pass Filter. V IN R 6.4kΩ 4. R C 3 pf 4. Low-pass 3-pole Butterworth f 3dB = 4kHz C pf R 4.36kΩ C 2 pf See Application Bulletin AB-26 for information on GIC filters. FIGURE 4. Three-Pole Generalized Immittance Converter (GIC) Low-Pass Filter. 9

C * I-Out DAC C OUT R 2.94kΩ C 2 22pF R 3 * C = ~ C OUT 2π R f c C 3 47pF Low-pass 2-pole Butterworth f 3dB = 2kHz R = Feedback resistance = f c = Crossover frequency = 8MHz FIGURE. DAC I/V Amplifier and Low-Pass Filter. 7.87kΩ kω kω V IN + pf G = 7.87kΩ khz Input Filter kω kω FIGURE 6. Differential Amplifier with Low-Pass Filter.

Ω kω * C C OUT 2π R f f c Piezoelectric Transducer MΩ* G = (4dB) * Provides input bias current return path. PCM63 2-bit D/A Converter 6 9 R f = Internal feedback resistance =.kω f c = Crossover frequency = 8MHz C * = ±3Vp To low-pass filter. FIGURE 7. High Impedance Amplifier. FIGURE 8. Digital Audio DAC I-V Amplifier. /2 A 2 I 2 /2 R 3 Ω R 4 Ω A I L = I + I 2 V IN i UT Load R UT = V IN ( + /R ) FIGURE 9. Using the Dual Op Amp to Double the Output Current to a Load. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems.

SOUND QUALITY The following discussion is provided, recognizing that not all measured performance behavior explains or correlates with listening tests by audio experts. The design of the included consideration of both objective performance measurements, as well as an awareness of widely held theory on the success and failure of previous op amp designs. SOUND QUALITY The sound quality of an op amp is often the crucial selection criteria even when a data sheet claims exceptional distortion performance. By its nature, sound quality is subjective. Furthermore, results of listening tests can vary depending on application and circuit configuration. Even experienced listeners in controlled tests often reach different conclusions. Many audio experts believe that the sound quality of a high performance FET op amp is superior to that of bipolar op amps. A possible reason for this is that bipolar designs generate greater odd-order harmonics than FETs. To the human ear, odd-order harmonics have long been identified as sounding more unpleasant than even-order harmonics. FETs, like vacuum tubes, have a square-law I-V transfer function which is more linear than the exponential transfer function of a bipolar transistor. As a direct result of this square-law characteristic, FETs produce predominantly even-order harmonics. Figure shows the transfer function of a bipolar transistor and FET. Fourier transformation of both transfer functions reveals the lower odd-order harmonics of the FET amplifier stage. I C (ma) I D (ma) V BE V GS I C.6 V BE (V) I D V GS (V) FFT FFT log ( ) log ( ) V BE = khz + DC Bias f O 2f O 3f O 4f O f O 3 4 Frequency (khz) 3 4 Frequency (khz) V GS = khz + DC Bias f O 2f O 3f O 4f O f O FIGURE. I-V and Spectral Response of NPN and JFET. I 8µA R R R 7 7Ω 7Ω Ω 4kΩ (+) J J 2 J 3 J 4 ( ) R 3 kω R 4 kω Q Distortion Rejection Circuitry R 8 3kΩ Q3 R 6 Ω R 9 3kΩ Output Stage THE DESIGN The uses FETs throughout the signal path, including the input stage, input-stage load, and the important phase-splitting section of the output stage. Bipolar transistors are used where their attributes, such as current capability are important and where their transfer characteristics have minimal impact. The topology consists of a single folded-cascode gain stage followed by a unity-gain output stage. Differential input transistors J and J 2 are special large-geometry, P- channel JFETs. Input stage current is a relatively high 8µA, providing high transconductance and reducing voltage noise. Laser trimming of stage currents and careful attention to symmetry yields a nearly symmetrical slew rate of ±2V/µs. The JFET input stage holds input bias current to approximately pa, or roughly 3 times lower than common bipolar-input audio op amps. This dramatically reduces noise with high-impedance circuitry. The drains of J and J 2 are cascoded by Q and Q 2, driving the input stage loads, FETs J 3 and J 4. Distortion reduction circuitry (patent pending) linearizes the openloop response and increases voltage gain. The 2MHz bandwidth of the further reduces distortion through the user-connected feedback loop. The output stage consists of a JFET phase-splitter loaded into high speed all-npn output drivers. Output transistors are biased by a special circuit to prevent cutoff, even with full output swing into 6Ω loads. The two channels of the are completely independent, including all bias circuitry. This eliminates any possibility of crosstalk through shared circuits even when one channel is overdriven. I 2 Q 2 2µA Q 4 J 2