MGA-63P8 1.9 GHz low noise amplifier Application Note 595 Introduction The MGA-63P8 is a GaAs EPHEMT LNA with integrated active bias. The target applications are Tower Mounted Amplifiers and LNAs in cellular infrastructure, including Personal Cellular System and UMTS bands. This paper provides application information for 1.9 GHz operation. The MGA-63P8 is packaged in a miniature by mm leadless package. The package s bottom side has a ground contact in the centre with a large surface area for efficient heat dissipation and low inductance RF grounding. This application note describes the use of the MGA-63P8 in an extremely high dynamic range low noise amplifier (LNA). The demonstration board s nominal performance at 1.9 GHz is: G = 17.5 and output P1 = 18.9 m. Without any deliberate attempt to optimize the output match for best linearity, an output intercept point (OIP 3 ) of 35 m can be easily achieved. The input and output return losses are better than. Biasing Requirement The enhancement mode technology provides superior performance while allowing a DC grounded source amplifier with a single polarity power supply to be easily designed and built. The built-in active bias circuit takes care of the V GS spread required to cope with the batchto-batch variation in forward transconductance, g m. Circuit Description The input network consisting of C1 & L1 provides a match to Q1 s input and also impart a high pass characteristic to roll off an undesirable gain increase below the operating frequency. To reduce circuit loss at microwave frequencies, L1 should have the following characteristics: - high unloaded Q, (Q ul ) and, operated below its Self Resonant Frequency (SRF). High-Q wire-wound chip inductors are preferred to multilayer chip inductors to minimize potential degradation of noise figure and gain. Assuming that the matching network s loaded Q (Q l ) is known, the loss in the inductor can be estimated from: - loss = log C7 Q ul Q Q On the output side, L and C form the matching network. L also serves at the bias decoupling network in conjunction with C5 and C6. Resistor R1 provides a simple mean to tailor the gain to suit the application requirement. R1 nominal value is 16R; however, increasing R1 from a minimum value of 6R to a maximum of k will reduce the gain correspondingly from 18.5 to 16.. ul l Vd C5 C6 In C1 R 1 8 L C Out L1 Q1 4 5 C3 C4 R1 Figure 1. Demonstration board circuit
Demonstration Board A generic demonstration board is available for quick prototyping and evaluation of the MGA-63P8 in the VHF through 3 GHz range. The demonstration board is made from a 1 mil Rogers RO435 substrate. An FR4 substrate is used as backing material to provide mechanical rigidity and to increase the overall thickness to.8 mm to suit standard edge launched coax transitions. RF connections to the demonstration board are made via edge-mounted microstrip to SMA coax transitions, J1 and J. The demonstration board requires a single 4. V power supply. Measured Performance The demonstration board performance was measured under the following test conditions: - V d = 4. V, I d = 6 ma and f c = 1.9 GHz. The MGA-63P8 is intended for either first or second stage LNAs in cellular infrastructures. Both low noise figure (NF) and good return loss over a broad bandwidth are the critical requirements. Although a ceramic chip wire-wound inductor is suggested for the input matching network, it can be replaced with a higher Q air-core spring wound inductor for even lower NF. Figure. Component Placement Diagram 18 Gain Vs Rev. Isolation Vs Frequency S1 S1 17-36 16-38 15-4 14-4 13-44 1-46 11-48 1-5 9-5 Figure 3. Gain and Reverse Isolation vs. Frequency -34 8-54 Start: 9. MHz Stop:.9 GHz 4/7/6 17:1:3 8753ES dm Gain & Noise figure Vs. Frequency G F 19.5.95 19.9 18.5.85 18.8 17.5.75 17.7 16.5.65 16.6 15.5.55 15 Centre: 1.9 GHz Span: 5. MHz.5 6/7/6 1:6: HP897 dm1 Figure 4. Gain & Noise Figure vs. Frequency 1
The demonstration board amplifier exhibits good input and output return losses over a wide bandwidth. This minimizes detuning effects when the LNA is cascaded with other stages in the RF chain. For example, filters and aerials are especially susceptible to the adverse effects of reflective terminations. Designing the amplifier s input and output for a close match to 5 Ω over the operating bandwidth, prevents unpredictable shift in the cascaded frequency response. -5-1 -15 - -5-3 -35-4 -45 Input and Output Return Loss Vs Frequency S11 S -5 Centre: 1.9 GHz Span: 1. GHz 4/7/6 17:1:3 8753ES dm 18 16 14 1 1 8 6 4 k Start: 3. khz Stop: 6. GHz 4/7/6 17:1:3 8753ES dm Figure 6. Stability Factor vs. Frequency (calculated from measured s-par) Inadvertent coupling between the amplifier s input and output sides and other component parasitics can lead to instability in the upper microwave region. If there are pronounced gain peaks above its operating frequency, the amplifier may oscillate under certain operating conditions. In a wideband sweep test of the MGA-63P8 demonstration board up to GHz, no abnormal peak was observed in the frequency response. Figure 5. Return Loss vs. Frequency Like all microwave transistors, the MGA-63P8 s gain increase with decreasing frequency. If the low frequency gain increase is not rolled-off with the appropriate high-pass matching networks, the amplifier can break into self-oscillation below its operating frequency; in the tens of MHz range. To assess the effectiveness of the low frequency circuit stabilization described previously, the Rollett stability criterion was calculated from the measurement of the demonstration board s s-parameters. The MGA-63P8 demonstration board is designed to be unconditionally stable (k >1) over the range of frequencies that an 8753 network analyzer is capable of operating. This reduces the design effort required to adapt the MGA-63P8 into the final product. 15 1 5-5 -1-15 - -5-3 Wideband Gain Sweep G Start: Hz dm1 Stop:. GHz Figure 7. Wideband Gain Sweep 3
In third generation (3G) cellular systems such as WCDMA, the combination of high peak-to-average power ratio (PAR) of the hybrid-qpsk modulation and the simultaneous transmission and reception (frequency domain duplexing) impose a stringent linearity demand on the RF front-end. The duplex filters separating the transmit path from receive path have finite isolation. As a result, transmitter leakage may desensitize the LNA stage. The 1 gain compression point, P 1, indicates the upper limit of either the input or the output power level at which saturation has started to occur in the LNA. The GaAs PHEMT s well-recognized linearity advantage over other semiconductor technologies results in the MGA-63P8 having a relatively high P 1 value of 18.9 m. 17.8 G () Idd 17.6 78 17.4 76 17. 74 17 7 16.8 1 7 16.6 68 16.4 66 16. 1 G () 64 18.9 m 16 16.6 6 15.8 6 Start: m Pout Stop:. m Figure 8. Gain & Idd Current vs. Output Power The intercept point is another measure of amplifier linearity. The theoretical point when the fundamental signal and the third order intermodulation distortion are of equal amplitude is the third order intercept point, IP 3. The distortion level at other power levels can be conveniently calculated from the amplifier s IP 3 specification. Two test signals spaced 5 MHz apart were used for evaluating the MGA-63P8 demoboard. The large dynamic range between the fundamental tones and the intermodulation products meant that the latter is barely above the spectrum analyzer s noise floor. To measure the 3 rd order product amplitude accurately, a very narrow sweep span can be used to improve the signal to noise ratio. As a tradeoff from the narrow sweep span, only one fundamental and one 3 rd order intermodulation output signal can be practically displayed on the graph. Both the fundamental and intermodulation tones are overlaid over the same frequency axis for amplitude comparison ma 8 purpose. The IP 3, referenced to the output, can be calculated from: - IP = P + fund 3 IM where P fund +is the amplitude of either one of the fundamental outputs, and IM is the amplitude difference between the fundamental tones and the intermodulation products. The output intercept point, OIP 3, is approximately 35 m. m -1 - -3-4 -5-6 -7-8 -9 fund imd -1 Start: 1.89 GHz Vid BW: 3 Hz Stop: 1.89 GHz Res BW: 3 Hz Sweep: 67 ms 13/7/6 17::54 HP8563E dm1, f1 & f=1897.5 & 19.5 MHz, Pi = - m Figure 9. Fundamental Output Tone Overlaid Over the 3rd. Order Inter- Modulation Product m Pout Imd Oip3 Figure 1. Fundamental Output Power, 3rd Order IM & OIP3 vs. Input Power 1 1 fund 1.89 GHz -. m imd 1.89 GHz -8.7 m 1 35 3-1 5 - -3 15-4 1-5 5-6 -7-5 m 4-8 -1 Start: -. m Pin Stop: -6. m 4
The nominal performance of the MGA-63P8 demonstration board is summarized in Table 1. Table 1. Demonstration board nominal performance values Vsupply (V) 4. Isupply (ma) 6 Fc (MHz) 19 G () 17.5 RL in () < - RL in () < - k > 1 P1 (m) 18.9 OIP3 (m) 35 Demonstration board part List The demonstration board s table of components is listed in Table. below. Part Size Value Description L1 4 3.3 nh Coilcraft 4CS3N3XB L 4.7 nh Coilcraft 4CSN7XB C1 4 3. pf Rohm MCH155A3JK C 4. pf Rohm MCH155AJK C3 4 1. pf Rohm MCH155A1RJK C4 4 1 pf Rohm MCH155A11JK C5 4.1 µf Kyocera CM5X5R14K1AH C6 4 9 pf Rohm MCH155A9DK C7 4.1 µf Kyocera CM5X5R14K1AH Q1 Avago MGA-63P8 R1 4 16 W Rohm MCR1MZSJ161 R 4 6 W Rohm MCR1MZSJ511 5
Appendix 1: Designing for stability The absence of Miller capacitance in the cascode topology results in considerably higher gain than a common emitter amplifier [1]. The increased gain at microwave frequencies may cause undesired gain peaks way above the design frequency []. Depending on variation in the layout and component parasitic, the MGA-63P8 design in AN595 may exhibit a pronounced gain peak ( 1 ) at 9 GHz. The gain peak may reduce the stability factor (k) to less than 1 at 9 GHz. mga63p8, f=1g9, c6=9p S1 k 1 15 9 1 8 5 7 6-5 5-1 4-15 3 - -5 1-3 Start: 5. MHz Stop:. GHz Feb 8 c6=9p Figure 1. Gain and Stability Factor (k) of the MGA-63P8 Amplifier in AN595 Showing Potential Instability at 9 GHz A 63 sized capacitor is suitable for C9 as it can bridge the gap between the MGA-63P8 output terminal and a nearby pair of grounded via-holes. One end of C9 is soldered to C and the other end to the via-holes. In C1 L1 C7 C4 R 1 Q1 8 4 5 R1 Vd L C3 C5 C6 C Out C9 SOLDER TO C Figure 3. Assembly Details for Adding Shunt Capacitor C9 (yellow high-lighted) to the Demonstration Board S1 k 1 15 9 1 8 5 7 6-5 5-1 4-15 3 - -5 1-3 Start: 5. MHz e Stop:. GHz 5 Feb 8 Figure 4. Gain and Stability Factor (k) of the MGA-63P8 after Adding C9 The modification to improve stability has little effect on other critical parameters. However, the stability is greatly improved. C9 C SOLDER TO VIA-HOLES Figure. MGA-63P8 amplifier Circuit Showing the Addition of a Shunt Capacitor C9 at the Output 6
Table 1. Performance Comparison Before and After the Addition of the C8 Shunt Capacitor Test parameter AN595 at.1 GHz Shunt C (.5 pf) at.1 GHz Shunt C (.5 pf) at 1.95GHz IRL () 15 15 15.6 ORL () 19 1 14.3 Gain () 16.6 16. 16.8 Isolation () 4 4 4 NF ().78.78.74 OIP3 (m) +35.5 +34. +34.5 Stability factor, k <1 (around 9 GHz) >1 >1 References 1. C. Baringer, and C. Hull, Amplifiers for Wireless Communications in RF And Microwave Circuit Design for Wireless Communications, L. E. Larson, Ed., Artech House, Norwood, MA., 1997.. A. J. Ward, Appendix I Determining the optimum amount of source inductance in High Intercept Low Noise Amplifier for the 193-199 MHz PCS Band using the Avago ATF-34143 Low Noise PHEMT, Avago Technologies Application Note 1191, Nov. 1999. For product information and a complete list of distributors, please go to our web site: www.avagotech.com Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies in the United States and other countries. Data subject to change. Copyright 5-1 Avago Technologies. All rights reserved. Obsoletes AV1-379EN AV-119EN - July 16, 1