Waveform Measurements on a HEMT Resistive Mixer

Jan Verspecht bvba Gertrudeveld 15 1840 Steenhuffel Belgium email: contact@janverspecht.com web: http://www.janverspecht.com Waveform Measurements on a HEMT Resistive Mixer D. Schreurs, J. Verspecht, B. Nauwelaers, A. Barel, M. Van Rossum Presented at the 47th ARFTG Conference 1996 Agilent Technologies - Used with Permission

WAVEFORM MEASUREMENTS ON A HEMT RESISTIVE MIXER D. Schreurs, J. Verspecht', B. Nauwelaers, A. Barel" and M. Van Rossum*** K.U.Leuven, div. ESAT-TELEMIC, Kard. Mercierlaan 94, B-3001 Heverlee, Belgium phone: +32-16-281485, fax: +32-16-281214, e-mail: schreurs@imec.be * Hewlett-Packard NMDG, VUB-ELEC, Pleinlaan 2, B-1050 Brussel, Belgium ** Vrije Universiteit Brussel, Dpt. ELEC, Pleinlaan 2, B-1050 Brussel, Belgium *** IMEC, div. ASP/CSP, Kapeldreef 75, B-3001 Heverlee, Belgium ABSTRACT Calibrated on-wafer waveform measurements under two-tone stimuli are demonstrated on a HEMT, configured as a resistive mixer. These large-signal measurements allow us not only to determine the conventional performance parameters, but also to analyse the influence of the phase relationship between the two excitation signals on the characteristics. For the considered HEMT resistive mixer, the dependency of the intermodulation products on the phase relationship between the LO-signal and the RF-signal becomes significant at high RF-powers. I. INTRODUCTION Calibrated on-wafer waveform measurements under two-tone stimuli are described. The ex- amined non-linear circuit is a HEMT, configured as a resistive mixer. As first been proposed by Maas in 1987 [l], the practical interest of this mixer type is its low DC-power consumption and its low intermodulation products, e.g. [2-51. The measurement system consists of the non-linear vector network analyzer [6], which measures the phase and amplitude of the spectral components of the incident and scattered voltage waves at the signal ports of a non-linear microwave device under test. This instrument provides an enhanced phase calibration accuracy by the nose-to-nose calibration part, as compared to non-linear measurement systems based on a Microwave Transition Analyzer [7-91. Moreover these MTA-based measurement systems currently serve only its verification or extraction medium for active devices under a one-tone excitation [lo-121. Section I1 describes the measurement set-up and the measurement conditions. The measurement results and the analysis will be presented in section 111. Section IV shows that a behavioural model for the resistive mixer can be deduced, since these non-linear measurements contain both the amplitude and the phase information of all the voltage waves. 11. MEASUREMENT CONFIGURATION For this experiment we configured a 0.2 pm gatelength, 100 pm gatewidth pseudomorphic GaAs HEMT a,s a resistive mixer, but the proposed procedure is valid for any MESFET or HEMT device. The on-wafer measurement set-lip is schematically depicted in Fig. 1. 129

- I, - f, *- -- - - The LO-signal (3 GHz) is a voltage wave arriving at the gate of the transistor, while the RF-signal (4 GHz) is a voltage wave sent towards the drain of the transistor. Both the LO- and RF-peak amplitude are swept and at each setting of the two powers, the phase relationship between the two components is randomized. This randomization is necessary since the phase of the LO and RF sources can not be set to a predefined value. The gate and drain load impedance is 50 0. The incident and scattered voltage waves at gate and drain, together with the DC gate current and the DC drain current are measured. Although the DC gate bias V,, can be swept to find the maximum conversion gain at a gate bias near pinch-off, we have fixed it at 0 V. The DC drain bias is zero, which means that the HEMT is biased in the cold HEMT condition. The small-signal equivalent scheme of the cold HEMT [13] is presented in Fig. 2. - - drain Ld =3 Ls RS source Fig. 2: Small-signal equivalent scheme of the cold HEMT. The dominant non-linearity in the cold HEMT is the channelresistor Rch between drain and source. This means that the drain terminal will behave as a resistor, with a value depending on the instantaneous gate voltage. Fig. 3 presents the V,, dependence of %( ZZ2), transformed from S-pa.rameter measurements at miilt,iple r;, values. %(&2) is equal to the slim of Rch a.nd the extrinsic, bias-indepenclent drain resistor Rd and source resistor R, [13]. 130

-600.OE-03 4OO.OE-03 Fig. 3: S(Z22) versus V,, of a 0.2 pm, 100 pm cold pseudomorphic GaAs HEMT. 111. MEASUREMENT RESULTS AND ANALYSIS The behaviour of the drain terminal as a resistor is illustrated in Fig. 4. It presents the waveform of the incident LO voltage wave at the gate, the incident RF voltage wave at the drain and the scattered voltage wave at the drain.... 0 oom --- >>> YYY 4 m4.4... 0 oom I l l I O.OEtO0 TIME [SI 1.OE-09 Fig. 4: HEMT resistive mixer waveforms: incident LO voltage wave A1 at the gate (x), incident RF voltage wave A2 at the drain (0) and the scattered waveform B2 a.t the drain (+) (LO-power= 5.4 dbm, RF-power=-S.4 dbm). When the instantaneous incident LO voltage wave is minimum, the resistor at the drain terminal behaves as an open (Fig. 3). This implies that the instantaneous scattered voltage wave at the drain is in phase to the instantaneous incident RF voltage wave at the drain. This can be seen on Fig. 4, due to the accurately measured phase relationship between 131

the voltage waves. The opposite holds when the instantaneous incident LO voltage wave is maximum and consequently the resistor at the drain terminal behaves as a short. The corresponding instantaneous scattered voltage wave at the drain is indeed in opposite phase to the instantaneous incident voltage wave at the drain. The measured conversion gain is -4.9 db, while the theoretically maximum conversion gain for an ideal mixer is or -3.9 db. The difference is caused by the not perfect short condition at maximum instantaneous incident LO voltage wave. This is due to the ohmic access resistances R, and Rd, which are not negligible compared to the 50 R load. Namely, a typical value for R, + Rd of a 100 pm gatewidth HEMT is 7 R. Fig. 5 shows the waveform of the incident LO voltage wave at the gate, the incident RF voltage wave at the drain and the scattered voltage wave of the intermodulation product at 1 GHz at the drain, By normalizing the latter to e-j'('lo)ej@('rf), with (1~0) the phase of the incident LO voltage wave and (JRF) the phase of the incident RF voltage wave, the measured phase of this normalized intermodulation product at 1 GHz is 182". This is nearly lso", as we expect from theory. I 0. OEtOO TIME [SI 1.OE-09 Fig. 5: :HEMT resistive mixer waveforms: incident LO voltage wave at the gate (x), incident R,F voltage wave at the drain (0) and the scattered waveform IF1 of the 1 GHz intermodulation product at the drain (+) (LO-power= 5.4 dbm, RF-power=-8.4 dbm). IV. NON-LINEAR MEASUREMENTS BASED MODELLING These measurements can be represented by a behavioural black-box model such as describing functions [14] or can immediately be implemented in tabular format in a commercial circuit simulator, e.g. HP-MDS. The dependent variables in these tables are the complex spectra of all the intermodulation products, the DC gate current and the DC drain current. The independent variables are the LO-power, the RF-power and the phase-relationship PH between the incident LO voltage wave and the incident RF voltage wave. This phase relationship arises from the time iiivariance requirement: applying a certain time delay to the input has 132

to result in the same delay at the output. For the intermodulation product at 1 GHz, PH is equal to @ (ej~~('~0)e-j3~('rf)) [14]. Fig. 6 shows the magnitude of the normalized intermodulation product at 1 GHz as a function of PH. Since this phase relationship has been randomized during the measurement, we haved fitted the measurements to a sine-series [15] in order to facilitate their listing in tabular format. -180. OEtOO PH [degrees] 180.OEt00 Fig. 6: Measured (+) and fitted (-) magnitude of the normalized intermodulation product at 1 GHz versus PH (LO-power=6.5 drm, RF-power=6.7 dbm). Fig. 7 presents 600 a.utomatically performed measurements of the conversion gain at 1 GHz versus RF-power and PH at LO-power equals 5.4 dbm. CG [db] Fig. 7: Conversion gain [db] vs. RF-power [dbm] and PH [degrees] at LO-power=5.4 dbm. Since maximum conversion gain is achieved at high LO-power and low RF-power, the validity range of most existing cold MESFET, e.g. [16], and HEhfT, e.g. [13], models is limited

to these conditions. From Fig. 7 we see that for the above measurement conditions, the conversion gain is constant up to a RF-power of about -5 dbm. We also notice that the influence of the phase-relationship PH on the conversion gain can be neglected for the normal resistive mixer working condition, but that this influence becomes significant at high RF-powers. This implies that the phase-relationship PH has to be included as independent variable in the non-linear behavioural black-box description of the resistive mixer. V. CONCLUSIONS Calibrated waveform measurements on a device, configured in resistive mixer mode, allow us not only to determine the resistive mixer performance, to verify the validity range of non-linear cold FET models, but also to analyse the influence of the input signal phases on the conversion gain at high input powers. VI. ACKNOWLEDGEMENTS The authors wish to express their gratitude to the complete staff of the ESAT-TELEMIC, ASP/CSP and HP-NMDG groups. We especially wish to thank P. Richardson for the pseudomorphic GaAs HEMT processing. D. Schreurs is supported by the National Fund for Scientific Research as a Research Assistant. REFERENCES [l] A. Maas, "A GaAs MESFET mixer with very low intermodulation", IEEE Trans. Microwave Theory and Techn., Vol. 35, No. 4, pp. 425-429, April 1987 [2] F. De Flaviis and S.A. Maas, "X-Band Doubly Balanced Resistive FET Mixer with Very Low Intermodulation", IEEE Trans. Microwave Theory and Techn., Vol. 43, No. 2, pp. 457-460, February 1995 [3] T. Chen, K. Wang, S. Bui, L. Liu, G. Dow and S. Pak, "Broadband Single- and Double- Balanced Resistive HEMT Monolithic Mixers", IEEE Trans. on Microwave Theory and Techn., Vol. 43, No. 3, March 1995, pp. 477-484, [4] I<. Yhland, N. Rorsman and H. Zirath, "A Novel Single Device Balanced Resistive HEMT Mixer", IEEE MTT-S, pp. 1411-1414, 1995 [5] C. I<arlsson, N. Rorsman and H. Zirath, "A Monolithically Integrated F-Band Resistive InAlAs/InGaAs/InP HFET Mixer", IEEE Microwave and Guided Wave Letters, Vol. 5, No. 11, pp. 394-395, November 1995 [6] J. Verspecht, P. Debie, A. Bare1 and L. Martens, "Accurate On Wafer Measurement Of Phase And Amplitude Of The Spectral Components Of Incident And Scattered Voltage Waves At the Signal Ports Of A Nonlinear Microwave Device", IEEE MTT-S, pp. 1029-1032, 1995 [7] F. van Raay and G. Kompa, "A New On-Wafer Large-Signal Waveform Measurement System with 40 GHz Harmonic Bandwidth", IEEE MTT-S, pp. 1435-1438, 1992 [SI M. Demmler, P.J. Tasker and M. Schlechtweg, "A Vector Corrected High Power On-wafer Measurement System with a frequency Range for the higher Harmonics up to 40 GHz", in Proc. of the 24th European Rlicrowave Conference, pp. 1367-1372, 1994 [9] J.G. Leckey, A.D. Patterson and J.A.C. Stewart, "A Vector Nonlinear Measurement System for Microwave Transistor Characterisation", CAE Modelling and Measurement Verification Workshop, London, U.K., pp. 190-193, 1994 134

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