Spectral shaping of Galileo signals in the presence of frequency offsets and multipath channels

Transcription

1 Spectral shaping of Galileo signals in the presence of frequency offsets and multipath channels Elena Simona Lohan, Abdelmonaem Lakhzouri, and Markku Renfors Institute of Communications Engineering, Tampere University of Technology P.O. Box 553, FIN-33101, Finland, Tel: , Fax: {elena-simona.lohan, abdelmonaem.lakhzouri, markku.renfors}@tut.fi Abstract Recently, the authors have proposed a new class of Binary- Offset-Carrier BOC modulation, namely the Double-BOC DBOC modulation which generalizes the main modulation types used in GPS and Galileo under a gle, unified formula, and allows extra-flexibility in designing the suitable spectral shaping of future GNSS signals. The goal of this paper is to extend the previous analysis to multipath channels and to study the interference between various DBOC-modulated signals and the standard GPS signals in the presence of frequency offsets. Keywords: Double-Binary-Offset-Carrier DBOC modulation, Galileo, GPS, power spectral densities PSD, spectral separation coefficients SSC. I. INTRODUCTION The modulation type used for standard GPS signals, such as C/A and PY codes, is the Binary Phase Shift Keying BPSK modulation [1]. The more recently proposed GPS and Galileo signals, such as GPS M-code, Open-Services OS and Publicly-Regulated-Services PRS signals, use a e or coe Binary-Offset-Carrier BOC modulation, described in [2], [3], [4], [5]. The three modulation types, namely BPSK, e BOC SinBOC and coe BOC CosBOC, can be described by the help of a new, generalized class, introduced in [6] and denoted there as Double-BOC DBOC modulation class. DBOC modulation acts, indeed, as a double-stage-boc modulation, and allows not only a general formulation of the current GPS and Galileo modulation types, but also more flexibility in shaping the desired signal spectrum and in achieving the target interference between signals sharing the same frequency bands. The goal of this paper is to analyze the DBOC-modulated signals in the presence of multipath channels and frequency offsets and to show the additional benefits of ug the DBOC class in the context of Galileo OS and PRS signals. By difference with previously defined modulation types i.e., BPSK, Sin- BOC, and CosBOC, the DBOC modulation is fully described via three parameters: the BOC-modulation order of the first stage N BOC1, the BOC-modulation order of the second stage N BOC2, and the chip rate f c. In the next section, we will briefly review the concept and main properties of DBOC modulation class. In section III, we introduce the theoretical derivations for multipath fading channels and frequency offsets. Section IV presents the simulation results and Section V summarizes the results of this paper. This work was carried out in the project Advanced Techniques for Mobile Positioning MOT funded by the National Technology Agency of Finland Tekes. This work is also partly supported by the Academy of Finland. II. DOUBLE-BOC DBOC MODULATION A DBOC-modulated wideband signal zt can be seen as the convolution between a DBOC waveform s DBOC t and a data waveform, as follows: zt=s DBOC t + S F n= k=1 b n c k,n δt nt sym k,1 where is the convolution operator, b n is the n-th complex data symbol for the pilot channels, they may be assumed to be all equal to 1, T sym is the symbol period, c k,n is the k-th chip corresponding to the n-th symbol, is the chip period, S F is the spreading factor S F = T sym /, δt is the Dirac pulse, and s DBOC t is the DBOC waveform, defined as [6]: s DBOC t =p TB t k=0 i=0 1 i+k δ t it B1 kt B. 2 Above, N BOC1 is the BOC-modulation order of the first stage, N BOC2 is the BOC-modulation order of the second stage, T B1 is the sub-chip interval after the first modulation stage: T B1 = /N BOC1, and p TB is a rectangular pulse of amplitude 1 T and support T B = c N BOC1 N BOC2, where T B is the sub-chip interval after the second modulation stage. The definition from eq. 2 of the DBOC waveform was inferred in [6] from the observation that CosBOC modulation is equivalent with a two-stage SinBOC modulation, in which the signal is first e-bocmodulated with a modulation order N BOC1 or, equivalently, a sub-carrier frequency f c N BOC1 /2, where f c =1/ is the chip rate, and then, the sub-chip is further split into two parts i.e., a second e-boc-modulation stage with N BOC2 =2is applied. By extending the concept to N BOC2 > 2, we got the expression of eq. 2. From the point of view of implementation, we notice that eq. 2 is also equivalent with: N sign BOC1 N BOC2 πt, if N BOC2 is odd, N s DBOC t= BOC1 N BOC2 πt 3 sign, if N BOC2 is even. N BOC1 πt

2 Therefore, for N BOC2 odd, the DBOC signal is generated exactly as a e-boc signal, with subcarrier frequency N BOC1 N BOC2 /, while for N BOC2 even, we need two subcarrier frequencies: one equals to N BOC1 N BOC2 /, and the other one equals to N BOC1 /. Hence, the generation of DBOC waveforms may be done via a quite straightforward implementation, by ug some voltage controlled oscillators, similarly with what was explained in [7] for e-boc signals. The DBOC modulation is completely defined via eq. 2 according to three parameters N BOC1,N BOC2,f c. For various factors N BOC2, we also cover the BPSK, the SinBOC and the CosBOC cases i.e., the main modulation types existing nowadays in GPS and Galileo systems: N BOC1 =1, N BOC2 =1 DBOC BPSK N BOC1 > 1, N BOC2 =1 DBOC SinBOC 4 N BOC1 > 1, N BOC2 =2 DBOC CosBOC N BOC1 > 1, N BOC2 > 2 Higher-order DBOC PSD [dbw Hz] PSD of various DBOC waveforms, f c =1.023 MHz BPSK: NBOC1=1, NBOC2=1 e BOC1,1: NBOC1=2, NBOC2=1 coe BOC1,1: NBOC1=2, NBOC2=2 Higher order DBOC: NBOC1=2, NBOC2= Frequency [MHz] The expression of eq. 2 allowed us to compute the normalized Power Spectral Density PSD P DBOC f of DBOCmodulation class in a straightforward and generic way, as shown in [6] BPSK, SinBOC, and CosBOC are particular cases of the following expressions, as seen from eq. 4: 1. If N BOC1 = odd and N BOC2 = odd : P DBOC f= 1 2 πftb cos πf πfcos 5 πft B 2. If N BOC1 = even and N BOC2 = odd : P DBOC f= 1 2 πftb πf πfcos 6 πft B 3. If N BOC1 = odd and N BOC2 = even : P DBOC f= 1 2 πftb πftb1 cos πf πfcos 7 πft B cos πftb1 4. If N BOC1 = even and N BOC2 = even : P DBOC f= 1 πftb πftb1 πf πfcos πft B cos πftb1 N BOC1 N BOC2 where Λ TB t is the triangular pulse of support 2T B, i.e., the ACF of a rectangular pulse of support T B. Eq. 9 allows us to compute, in a generic way, the ACF of a DBOC-modulated 2 signal in the presence of multipath fading channels, as it will be shown in Section III. 8 The ACFs of several DBOC-modulated signals are shown in Fig. 2. We notice that, typically, the higher the N BOC2 is, the are the sub- where T B1 = N BOC1 and T B = chip intervals after the first and, respectively, the second BOC-modulation stages. In eqs. 5 to 8, the normalization was done with respect to the chip period, or, equivalently, to the signal power over infinite bandwidth, similar to [2]. Fig. 1 shows the normalized PSD of several DBOCmodulated signals, for a chip frequency f c =1.023 MHz. The SinBOC in this figure corresponds to the classical case BOC1,1 [4]. The PSDs are given in dbw-hz P DBOC [dbw- Hz] 10log 10 P DBOC 60. We remark that, when N BOC2 modulation order increases, the signal main lobes move further towards the outer sides of the spectrum. This is also in accordance with the observations in [4], where it was noticed that Fig. 1. Examples of PSDs for DBOC-modulated waveforms. the CosBOC modulation had the advantage that it concentrated more power on outer sides of spectrum compared to the SinBOC modulation of the same order. Hence, the interference with GPS signals could be reduced via increag N BOC2, at the expense of a higher bandwidth as seen in Fig. 1. The autocorrelation function ACF of a DBOC waveform can also be easily derived based on eq. 2: R DBOC t s DBOC t s DBOC t=λt j=0 i=0 l=0 k=0 1 k+j+i+l δt it B1 + lt B1 kt B + jt B, 9 smaller width of the main lobe we have and hence, better resolution during delay tracking. On the other hand, due to an increased N BOC1 N BOC2 product, the number of sidelobes within two-chips interval increases, thus increag the ambiguities in the signal ACF, which might make the acquisition process more difficult [8]. III. BEHAVIOUR IN THE PRESENCE OF MULTIPATH FADING CHANNELS AND FREQUENCY OFFSETS The baseband equivalent model of a DBOC-modulated signal xt received over a fading multipath channel with additive Gaussian noise ηt is yt =xt s DBOC te j2πfot ε ch t+ηt, 10

3 Normalized ACF ACF for DBOC modulated signals, OS BPSK: NBOC1=1, NBOC2=1 SinBOCn,n: NBOC1=2, NBOC2=1 CosBOCn,n: NBOC1=2, NBOC2=2 Higher order DBOC: NBOC1=2, NBOC2= Chips Fig. 2. Examples of the normalized ACFs of DBOC-modulated waveforms where xt= S F n= k=1 b n c k,n δt nt sym k is the spread data sequence, f o is the frequency offset with respect to the carrier frequency, and ε ch t is the channel impulse response: ε ch t = L α l δt τ l, l=1 with α l being the complex channel coefficients of l-th path, τ l the corresponding multipath delays, and L the number of channel paths. If we assume that the signal and the noise are uncorrelated processes, the PSD of signal yt, conditional to the channel coefficients α l and delays τ l, can be derived from eq. 10 as: 2 L P y f =X PSD fp DBOC f f o α l e j2πfτ l + N 0, l=1 11 where N 0 is the noise PSD in the considered bandwidth and X PSD f is the PSD of the spread data sequence, and P DBOC f f o is the PSD of the DBOC-modulated waveform given in Section II, eqs. 5 to 8 and centered at the frequency offset f o. The unconditional PDF of yt signal was computed in the simulations by averaging over several random channel realizations. A theoretical derivation of the PSD of a spread sequence xt can be found in [9]. However, in what follows, we assume that the code chips and the data symbols are independent and that the code sequence has ideal autocorrelation properties, i.e., Ec k,n c i,m =δn mδk i. Inthis case, X PSD f=1. The spectral separation coefficient SSC between two DBOC-modulated signals in the presence of multipath fading channels and frequency offsets, is defined similar with [2], [6]: = BT /2 B T /2 P y1 f P y2 fdf, 12 where B T is the complex receiver bandwidth over which the SSC is computed, Pyυ f is the PSD of the υ-th signal υ =1, 2, normalized over bandwidth B T : Pyυ f BT /2 P yυ f/ B T /2 P y υ fdf, and P yυ f is given by eq. 11. The SSC coefficients are useful measures of the interference between future Galileo signals and existing GPS signals i.e., C/A code, M-code and PY-code, as well as for the characterization of the self-interference [2], [6]. In terms of the autocorrelation function, the ACF of the received signal, after the removal of data modulation and for zerofrequency offset is: R y t y y t= L l=1 l 1=1 L α l α l 1 R DBOC t τ l + τ l1 + N 0 δt, 13 where the upper-script stands for the conjugate and R DBOC t is given in eq. 9. IV. SIMULATION RESULTS In the simulations, we considered Rayleigh and Rician fading channels, with decaying power delay profiles, meaning that the relationship between the average path powers of two consecutive paths P l E α l 2 and P l+1 E α l+1 2 was given by P l+1 = P l e µτ l+1 τ l, where µ is the exponential-decaying coefficient here, µ =10 6 f c. The number of channel paths was assumed to be random and uniformly distributed between 1 and L max L max is specified in the figures captions or labels. The maximum separation between successive paths was assumed to be 2 chips however, the simulation results showed that this parameter has no impact on the results from the point of view of spectral properties of the DBOC signals. Fig. 3 shows the SSC mean and maximum or worst-case values between various DBOC signals and the standard GPS signals. The average and the maximum values of SSC are computed over 5000 random channel realizations. The average values are plotted with continuous lines, and the worst-case values are plotted with dashed lines this holds also for Fig. 4. Both desired and interfering signals are assumed to have a Rician distribution of the first path and Rayleigh distribution for successive paths if any, and the channels of desired and interfering signals are modelled as explained above. Here, we assume that the maximum number of channel paths is the same for the desired and interfering signal, and this number is shown in the horizontal axis. We see here that the channel profiles the number of paths have little impact on the SSC factors, fact which was also verified for Rayleigh-fading-first-path channels and for various other levels of Carrier-to-Noise Ratio CNR. We also see that, when N BOC1 =2as in Fig. 3, by increag the modulation order of the second stage, we may decrease the spectral separation with C/A code. The spectral separation factors with the other GPS codes remain almost unchanged in this case. The difference between the worst-case and the average SSC values can be up to 3.5 dbw/hz and it is slightly worst for C/A code and self-ssc factors than for SSC with PY and M codes.

4 =1 =2 mean and max [dbw/hz] Mean SSC with PY code Max SSC with PY code mean and max [dbw/hz] Mean SSC with PY code Max SSC with PY code mean and max [dbw/hz] =4 Mean SSC with PY code Max SSC with PY code mean and max [dbw/hz] =6 Mean SSC with PY code Max SSC with PY code Fig. 3. Average spectral-separation coefficients for 4 DBOC signals in the presence of multipath fading channels: N BOC1 =2as for OS and N BOC2 =1,2,4, and 6, respectively. Zero frequency offset, CNR=45 db-hz. The maximum Value of the Spectrum MVS and the Root Mean Square bandwidth RMS BW for various DBOC signals are shown in Fig. 4. For MVS and RMS BW, the definition of [2] was used. An increased number of paths means a higher level of interference, and therefore MVS value increases, as seen in the upper plot of Fig. 4. On the other hand, the RMS BW and hence, the inverse of the variance of delay tracking process [10] decreases very slowly with the increase in the number of paths. The best values for MVS and RMS BW both the average and worst-case values are achieved here with a higher-order DBOC modulation i.e., N BOC2 =4. Therefore, increag the second-stage modulation order may be beneficial in achieving better spectral properties. Fig. 5 shows the average SSC coefficients for the interference between the current Galileo candidates and the GPS signals in the presence of multipath channels and frequency offsets. The multipath channels were having up to 4 paths, with power decay profiles and maximum separation between consecutive paths of 2 chips. Similar results have been obtained for all the studied channel profiles, as illustrated before e.g., in Fig. 3, therefore the channel profile does not seem to affect the results in any significant way. On the other hand, the impact of the frequency offsets is much more significant, as seen in Fig. 5. The candidate considered for OS services is SinBOC1,1 upper plot of Fig. 5 and the candidate for PRS services is CosBOC15,2.5 lower plot of Fig. 5. The channels were assumed to be Rician fading channels, with random number of paths and decaying power delay profile. Similar curves were obtained for other channel profiles as well. The receiver bandwidth was taken equal to B T =8MHz for OS and B T =40 MHz for PRS [11]. We remark from Fig. 5 upper plot that the lowest,c/a code for OS is obtained for about 4 MHz frequency offset between the SinBOC1, 1 signal and the C/A code. Similarly, the lowest,m code for OS is obtained for about 5.11 MHz frequency offset between the SinBOC1, 1 signal and the M code. For PRS signals shown in the lower plot of Fig. 5, the smallest,c/a code and,m code coefficients occur at zero frequency offset between the CosBOC15, 2.5 signal and the M-code. V. CONCLUSIONS This paper presents a new class of modulation for Galileo and modernized GPS signals, the Double-BOC modulation, in the presence of multipath fading channels and frequency offsets. The main properties of the DBOC family, namely the autocorrelation function and the power spectral densities, are derived analytically, in a generic framework, which allows a

5 Mean and worst case MVS values, B T =8 MHz, f c =1.023 MHz Mean MVS, SinBOC1,1: NBOC1=2, NBOC2=1 Mean MVS, CosBOC1,1: NBOC1=2, NBOC2=2 Mean MVS, higher order DBOC: NBOC1=2, NBOC2= 4 Max MVS, SinBOC1,1: NBOC1=2, NBOC2=1 Max MVS, CosBOC1,1: NBOC1=2, NBOC2=2 Max MVS, higher order DBOC: NBOC1=2, NBOC2= 4 SinBOC, L des =4, L int =4, CNR=60 db Hz, B T =8 MHz MVS [dbw/hz] [dbw/hz] SSC with C/A code SSC with M code SSC with PY code SSC with itself RMS bandwidth mean and worst case [dbw/hz] mean and worst case Mean and worst case RMS BW values, B =8 MHz, f =1.023 MHz 5 Mean RMS BW, SinBOC1,1: NBOC1=2, NBOC2=1 4.5 Mean RMS BW, CosBOC1,1: NBOC1=2, NBOC2=2 Mean RMS BW, higher order DBOC: NBOC1=2, NBOC2=4 Min RMS BW, SinBOC1,1: NBOC1=2, NBOC2=1 4 Min RMS BW, CosBOC1,1: NBOC1=2, NBOC2=2 Min RMS BW, higher order DBOC: NBOC1=2, NBOC2= [dbw/hz] Frequency offset [MHz] CosBOC, L des =4, L int =4, CNR=60 db Hz, B T =40 MHz SSC with C/A code SSC with M code SSC with PY code SSC with itself Frequency offset [MHz] Fig. 4. Maximum value of the spectrum upper plot and RMS bandwidth lower plot for 3 DBOC signals in the presence of multipath fading channels. B T =8MHz, f c =1.023 MHz, zero frequency offset, CNR=45 db-hz. unified analysis of the existing BPSK, SinBOC and CosBOC modulation types. Simulation results are also shown in order to exemplify the potential use of DBOC modulation in the context of future satellite navigation systems. We also showed that, via the DBOC modulation, the spectral shaping can be made more flexible, due to the two modulation orders, N BOC1 and N BOC2 which control the power spectral density of the signal and its bandwidth consumption. REFERENCES [1] J. Betz and D. Goldstein, Candidate Designs for an Additional Civil Signal in GPS Spectral Bands. MITRE Technical Papers, Jan [2] J. Betz, The Offset Carrier Modulation for GPS modernization, in Proc. of ION Technical meeting, pp , [3] B. Barker, J. Betz, J. Clark, J. Correia, J. Gillis, S. Lazar, K. Rehborn, and J. Straton, Overview of the GPS M Code Signal, in CDROM Proc. of NMT, [4] G. Hein, M. Irsigler, J. A. Rodriguez, and T. Pany, Performance of Galileo L1 signal candidates, in CDROM Proc. of European Navigation Conference GNSS, May [5] V. Heiries, D. Oviras, L. Ries, and V. Calmettes, Analysis of non ambiguous BOC signal acquisition performance, in CDROM Proc. of ION GNSS, Sep [6] E. Lohan, A. Lakhzouri, and M. Renfors, A novel family of Binary- Offset-Carrier modulation techniques with applications in satellite navigation systems. Technical Report, Tampere University of Technology, ISBN , ISSN , Apr 2005 also submitted to Wiley International Journal of Wireless Comm. and Mobile Computing. Fig. 5. SSC coefficients for the current OS upper plot and PRS lower plot Galileo candidates in the presence of multipath fading channels and frequency offsets. L des is the maximum number of paths of the desired DBOC signal, L int is the maximum number of paths of the interfering signal i.e., GPS signals or another DBOC signal. [7] L. Ries, F. Legrand, L. Lestarquit, W. Vigneau, and J. Issler, Tracking and multipath performance assessments of BOC signals ug a bit-level signal procesg simulator, in Proc. of ION-GPS2003, Portland, OR, US, pp , Sep [8] N. Martin, V. Leblond, G. Guillotel, and V. Heiries, BOCx,y signal acquisition techniques and performances, in Proc. of ION-GPS2003, Portland, OR, US, pp , Sep [9] N. Pronios and A. Polydoros, On the power spectral density of certain digitally modulated signals with applications to code despreading, IEEE Journal on Selected Areas in Comm., vol. 8, pp , Jun [10] J. Betz, Design and Performance of Code Tracking for the GPS M Code Signal, in CDROM Proc. of ION Meeting, Sep [11] J. A. Rodriguez, M. Irsigler, G. Hein, and T. Pany, Combined Galileo/GPS frequency and signal performance analysis, in CDROM Proc. of ION GNSS, Sep 2004.