Broadband 1.2- and 2.4-mm Gallium Nitride (GaN) Power Amplifier Designs

Transcription

1 ARL-TR-8180 OCT 2017 US Army Research Laboratory Broadband 1.2- and 2.4-mm Gallium Nitride (GaN) Power Amplifier Designs by John E Penn

2 NOTICES Disclaimers The findings in this report are not to be construed as an official Department of the Army position unless so designated by other authorized documents. Citation of manufacturer s or trade names does not constitute an official endorsement or approval of the use thereof. Destroy this report when it is no longer needed. Do not return it to the originator.

3 ARL-TR-8180 OCT 2017 US Army Research Laboratory Broadband 1.2- and 2.4-mm Gallium Nitride (GaN) Power Amplifier Designs by John E Penn Sensors and Electron Devices Directorate, ARL

4 REPORT DOCUMENTATION PAGE Form Approved OMB No Public reporting burden for this collection of information is estimated to average 1 hour per response, including the time for reviewing instructions, searching existing data sources, gathering and maintaining the data needed, and completing and reviewing the collection information. Send comments regarding this burden estimate or any other aspect of this collection of information, including suggestions for reducing the burden, to Department of Defense, Washington Headquarters Services, Directorate for Information Operations and Reports ( ), 1215 Jefferson Davis Highway, Suite 1204, Arlington, VA Respondents should be aware that notwithstanding any other provision of law, no person shall be subject to any penalty for failing to comply with a collection of information if it does not display a currently valid OMB control number. PLEASE DO NOT RETURN YOUR FORM TO THE ABOVE ADDRESS. 1. REPORT DATE (DD-MM-YYYY) October REPORT TYPE Technical Report 4. TITLE AND SUBTITLE Broadband 1.2- and 2.4-mm Gallium Nitride (GaN) Power Amplifier Designs 3. DATES COVERED (From - To) FY17 5a. CONTRACT NUMBER 5b. GRANT NUMBER 5c. PROGRAM ELEMENT NUMBER 6. AUTHOR(S) John E Penn 5d. PROJECT NUMBER 5e. TASK NUMBER 5f. WORK UNIT NUMBER 7. PERFORMING ORGANIZATION NAME(S) AND ADDRESS(ES) US Army Research Laboratory ATTN: RDRL-SER-E 2800 Powder Mill Road Adelphi, MD PERFORMING ORGANIZATION REPORT NUMBER ARL-TR SPONSORING/MONITORING AGENCY NAME(S) AND ADDRESS(ES) 10. SPONSOR/MONITOR'S ACRONYM(S) 11. SPONSOR/MONITOR'S REPORT NUMBER(S) 12. DISTRIBUTION/AVAILABILITY STATEMENT 13. SUPPLEMENTARY NOTES 14. ABSTRACT The US Army Research Laboratory is exploring devices and circuits for radio frequency communications, networking, and sensor systems of interest to Department of Defense applications, particularly for next-generation radar systems. Broadband, efficient, high-power monolithic microwave integrated circuit amplifiers are extremely important in any system that must operate reliably and efficiently in continually crowded spectrums, with multiple purposes for communications, networking, and radar. This report describes the design of a broadband Class A/B power amplifier using Raytheon s high-frequency, efficient, gallium nitride on 4-mil silicon carbide process. While this design was not part of the initial wafer fabrication for the original effort, it could be finalized and fabricated at a future date. 15. SUBJECT TERMS radio frequency, RF, gallium nitride, GaN, Raytheon, broadband power amplifier, fabrication 16. SECURITY CLASSIFICATION OF: a. REPORT Unclassified b. ABSTRACT Unclassified c. THIS PAGE Unclassified 17. LIMITATION OF ABSTRACT UU ii 18. NUMBER OF PAGES 26 19a. NAME OF RESPONSIBLE PERSON John E Penn 19b. TELEPHONE NUMBER (Include area code) Standard Form 298 (Rev. 8/98) Prescribed by ANSI Std. Z39.18

5 Contents List of Figures List of Tables Acknowledgments iv v vi 1. Introduction 1 2. Broadband Power Amplifier 1 3. Summary and Conclusion 14 List of Symbols, Abbreviations, and Acronyms 16 Distribution List 17 iii

6 List of Figures Fig. 1 Fig. 2 Microwave Office (MWO) schematic for the ideal power load and match ( μm HEMT nominal DC bias)... 1 MWO simulation of the ideal power load and match ( μm HEMT 2 to 10 GHz)... 2 Fig. 3 Schematic and MMIC layout of the broadband matching circuit ( µm HEMT)... 3 Fig. 4 MWO simulation of the ideal (magenta) and MMIC output match (purple)... 3 Fig. 5 Broadband impedance of output match ideal (solid) vs. MMIC (dotted) (RLoad -CDS)... 4 Fig. 6 ADS S-parameter simulation of the µm (1.2-mm) GaN HEMT... 5 Fig. 7 Stabilizing resistors added to the μm GaN HEMT plus broadband output match... 5 Fig. 8 Ideal, coupled line input match for the 2- to 7-GHz, 1.2-mm GaN HEMT power amplifier... 6 Fig. 9 Broadband, compact, folded, coupled line MMIC input match for the 2- to 8-GHz, 1.2-mm GaN HEMT power amplifier... 6 Fig. 10 Unfinished layout of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier... 7 Fig. 11 Small-signal simulation of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier... 7 Fig. 12 ADS simplified schematic of the 4- to 5-W, broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier... 8 Fig. 13 ADS dynamic load line simulation of the broadband (4.5-GHz), 1.2- mm HEMT power amplifier... 9 Fig. 14 ADS performance simulation of the broadband (4.5-GHz), 1.2-mm HEMT power amplifier Fig. 15 ADS small-signal simulation of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier Fig. 16 ADS performance simulation of the ideal broadband (4.5-GHz), 1.2- mm HEMT power amplifier Fig. 17 MWO partial schematic for the ideal, parallel, 2-way combined circuit ( µm HEMT) Fig. 18 ADS schematic for the ideal, parallel, 2-way combined circuit (2- to 1.2-mm HEMTs) iv

7 Fig. 19 Double, tuned, ideal load match for the parallel, 2-way combined circuit vs. the single HEMT Fig. 20 ADS performance simulation of the ideal broadband (4.5-GHz), 2.4- mm HEMT power amplifier List of Tables Table 1 ADS relative performance simulations of the 1.2- and 2.4-mm broadband HEMT power amplifiers v

8 Acknowledgments Research was sponsored by the US Army Research Laboratory (ARL) and was accomplished under Cooperative Agreement Number W911NF The views and conclusions contained in this document are those of the authors and should not be interpreted as representing the official policies, either expressed or implied, of ARL or the US Government. The US Government is authorized to reproduce and distribute reprints for Government purposes notwithstanding any copyright notation herein. I would like to acknowledge and thank the Raytheon team, especially Steve Bernstein, Steve Lardizabal, Rob Leoni, and Ken Wilson for the design support, computer-aided design support, technical expertise, and fabrication of these gallium nitride (GaN) circuits for emerging Department of Defense systems and applications. The ARL team of Edward Viveiros, Ken McKnight, Ali Darwish, Abigail Hedden, and John Clark provided their expertise and support in the design of these GaN designs in the Raytheon process. vi

9 1. Introduction The US Army Research Laboratory (ARL) has been working with Raytheon to design efficient, broadband, linear, high-power amplifiers and robust, broadband, low-noise amplifiers for future adaptive, multimodal radar systems. Raytheon has a high-performance, W-band, gallium nitride (GaN) fabrication process and a process design kit (PDK) that researchers at ARL used to design low-noise amplifiers, power amplifiers, and other circuits for future radar, communications, and sensor systems. After the first set of ARL and Raytheon designs were submitted to fabrication, I performed a couple test designs of broadband Class A/B power amplifiers. While these designs did not get fabricated in the initial effort, they serve to demonstrate the performance, bandwidth, and capability of this GaN process and could potentially be fabricated in the future. 2. Broadband Power Amplifier This report documents the preliminary design of a single high-electron mobility transistor (HEMT) and 2-way parallel combined HEMT power amplifier. These initial broadband power amplifiers are based on a µm HEMT at a nominal recommended DC bias. This size HEMT had an optimal match provided by Raytheon as RLoad ohms in parallel with a negative CDS pf in capacitance. Since a negative reactance can only be matched over a limited band, an initial design was performed of an ideal, double, tuned, Q bandpass match for broadband operation centered around 4.5 GHz, with a goal of achieving 2 to 10 GHz. A schematic of the ideal load as a resistor in parallel with a capacitor and the ideal, double, tuned output matching circuit is shown in Fig. 1. The simulation from 2 to 10 GHz of the ideal load (blue S11 trace) and ideal bandpass match (magenta S11 trace) are shown in the Smith chart plot (Fig. 2). PORT P=1 Z=50 Ohm PORT P=1 Z=50 Ohm CAP ID=C1 C=CP pf RES ID=R1 R=RP Ohm CAP ID=C3 C=CP1 pf IND ID=L3 L=LP1 nh IND ID=L2 L=Lser2 nh CAP ID=C2 C=Cser2 pf IND ID=L1 L=LP1 nh CAP ID=C1 C=CP1 pf RES ID=R1 R=RP Ohm PORT P=2 Z=50 Ohm Fig. 1 Microwave Office (MWO) schematic for the ideal power load and match ( μm HEMT nominal DC bias) 1

10 S(1,1) RP_CP S(1,1) OMN_1_2mm_Idl GHz g 1 b OMN GHz g 1 b GHz g b GHz g 1 b Swp Max 10GHz GHz g b Swp Min 2GHz Fig GHz) MWO simulation of the ideal power load and match ( μm HEMT 2 to After designing an ideal, lumped-element output match, the capacitors and inductors were replaced with monolithic microwave integrated circuit (MMIC) elements from the Raytheon GaN design library and retuned to achieve a broadband match. Then microstrip bends, tees, and decoupling elements for the DC bias were added to complete a layout of the MMIC output match (Fig. 3). A simulation of the output match (Fig. 4) shows a better than 20-dB return loss from 2.3 GHz to above 8.7 GHz (purple trace) versus the ideal, lumped-element, double, tuned match with slightly less bandwidth (magenta trace) but an excellent match midband. 2

11 SUBCKT ID=S4 NET="Svia1" 1 ID=TL6 TFC ID=V2 ID=TL4 MTEE ID=TL7 MRINDSB3 ID=MSP2 ID=TL3 3 MTEE ID=TL1 ID=TL PORT P=1 Z=50 Ohm MRINDSB3 ID=MSP4 3 TFC ID=V1 MTEE ID=TL2 3 MRINDSB3 ID=MSP3 Zout = j * 18 PORT P=2 Z=50 Ohm CAP ID=C3 C=CP1 pf IND ID=L3 L=LP1 nh IND ID=L2 L=Lser2 nh CAP ID=C2 C=Cser2 pf IND CAP ID=L1 ID=C1 L=LP1 nh C=CP1 pf ID=TL9 SUBCKT ID=S1 NET="BondPad1" ID=TL8 1 1 SUBCKT ID=S3 TFC ID=V4 NET="Svia1" 1 SUBCKT ID=S2 NET="Svia1" Fig. 3 HEMT) Schematic and MMIC layout of the broadband matching circuit ( µm db Fig. 4 MWO simulation of the ideal (magenta) and MMIC output match (purple) Comparing the impedance match of the ideal, lumped-element output match (solid lines) to the lossy MMIC output match (dotted lines) over frequency to the ideal RLoad normalized impedance (left axis) and negative CDS normalized capacitance (right axis) shows good broadband performance (Fig. 5). The ideal output match is close to the ideal RLoad load line of a 1.2-mm HEMT from 3.5 3

12 to 6 GHz, while the MMIC output match undershoots the real part of the impedance but stays very close to 95% of RLoad over a broader range of 3 to 7 GHz. Since an ideal reactance equivalent to a negative CDS capacitance can only be maintained over a finite bandwidth, the output matching circuits can be seen as matching well over the band, diverging at the low end of the frequency range (2 to 3 GHz). Resistances (left axis) in the plot are represented by shades of red and magenta, while capacitances (right axis) are represented by shades of blue. Fig. 5 Broadband impedance of output match ideal (solid) vs. MMIC (dotted) (RLoad -CDS) After designing the output match for the broadband power amplifier, the S- parameters of the μm (1.2-mm) HEMT are generated at the nominal DC bias. Initially, these S-parameters were exported from Advanced Design System (ADS) (Fig. 6) and imported into MWO to perform an initial amplifier design. Small-signal stability was analyzed and establish with a shunt resistor and a parallel series resistor and capacitor on the gate of the HEMT. Figure 7 shows that the source stability circles are all outside the Smith chart, indicating unconditional stability. After stabilizing the 1.2-mm HEMT, the input impedance at midband (4.5 GHz) was simulated resulting in a higher Q matching impedance (Q = 2.4) than the output, making it more difficult to broadband match the power amplifier input. An initial ideal input match provided better than 10-dB return loss from 3.5 to 6 GHz, but was limiting the amplifier bandwidth compared to the output matching circuit. An ideal, coupled line provided a broader frequency range for the input match, while sacrificing additional loss. A compact, spiral, coupled line was 4

13 p1: Freq = 0.5 GHz p5: Freq = 0.9 GHz p9: Freq = 1.3 GHz p13: Freq = 1.7 GHz p17: Freq = 2.1 GHz p21: Freq = 2.5 GHz p25: Freq = 2.9 GHz p29: Freq = 3.3 GHz p33: Freq = 3.7 GHz p37: Freq = 4.1 GHz p41: Freq = 4.5 GHz p45: Freq = 4.9 GHz p49: Freq = 5.3 GHz p53: Freq = 5.7 GHz p57: Freq = 6.1 GHz p61: Freq = 6.5 GHz p65: Freq = 6.9 GHz p69: Freq = 7.3 GHz p73: Freq = 7.7 GHz p77: Freq = 8.1 GHz p81: Freq = 8.5 GHz p85: Freq = 8.9 GHz p89: Freq = 9.3 GHz p93: Freq = 9.7 GHz p97: Freq = 10.1 GHz Disp Temp p2: Freq = 0.6 GHz p6: Freq = 1 GHz p10: Freq = 1.4 GHz p14: Freq = 1.8 GHz p18: Freq = 2.2 GHz p22: Freq = 2.6 GHz p26: Freq = 3 GHz p30: Freq = 3.4 GHz p34: Freq = 3.8 GHz p38: Freq = 4.2 GHz p42: Freq = 4.6 GHz p46: Freq = 5 GHz p50: Freq = 5.4 GHz p54: Freq = 5.8 GHz p58: Freq = 6.2 GHz p62: Freq = 6.6 GHz p66: Freq = 7 GHz p70: Freq = 7.4 GHz p74: Freq = 7.8 GHz p78: Freq = 8.2 GHz p82: Freq = 8.6 GHz p86: Freq = 9 GHz p90: Freq = 9.4 GHz p94: Freq = 9.8 GHz p98: Freq = 10.2 GHz p3: Freq = 0.7 GHz p7: Freq = 1.1 GHz p11: Freq = 1.5 GHz p15: Freq = 1.9 GHz p19: Freq = 2.3 GHz p23: Freq = 2.7 GHz p27: Freq = 3.1 GHz p31: Freq = 3.5 GHz p35: Freq = 3.9 GHz p39: Freq = 4.3 GHz p43: Freq = 4.7 GHz p47: Freq = 5.1 GHz p51: Freq = 5.5 GHz p55: Freq = 5.9 GHz p59: Freq = 6.3 GHz p63: Freq = 6.7 GHz p67: Freq = 7.1 GHz p71: Freq = 7.5 GHz p75: Freq = 7.9 GHz p79: Freq = 8.3 GHz p83: Freq = 8.7 GHz p87: Freq = 9.1 GHz p91: Freq = 9.5 GHz p95: Freq = 9.9 GHz p99: Freq = 10.3 GHz Raytheon p4: Freq = 0.8 GHz p8: Freq = 1.2 GHz p12: Freq = 1.6 GHz p16: Freq = 2 GHz p20: Freq = 2.4 GHz p24: Freq = 2.8 GHz p28: Freq = 3.2 GHz p32: Freq = 3.6 GHz p36: Freq = 4 GHz p40: Freq = 4.4 GHz p44: Freq = 4.8 GHz p48: Freq = 5.2 GHz p52: Freq = 5.6 GHz p56: Freq = 6 GHz p60: Freq = 6.4 GHz p64: Freq = 6.8 GHz p68: Freq = 7.2 GHz p72: Freq = 7.6 GHz p76: Freq = 8 GHz p80: Freq = 8.4 GHz p84: Freq = 8.8 GHz p88: Freq = 9.2 GHz p92: Freq = 9.6 GHz p96: Freq = 10 GHz p100: Freq = 10.4 GHz a technique suggested by the Raytheon team to provide broadband match, but at these frequencies requires significant area in the MMIC layout. A preliminary ideal transmission line input matching circuit provided good performance from 2 to 7 GHz (Fig. 8). The ideal input matching elements were replaced with MMIC components resulting in 2 relatively large inductors. Next, the folded, spiral, coupled line requires electromagnetic (EM) simulation to verify its performance. NsCircle GaCircle GaCircle GaCircle1 GaCircle1=ga_circle(S,2,51) GpCircle Term Term Term1 Term2 Num=1 Num=2 Z=50 Ohm Z=50 Ohm GpCircle L GpCircle1 P82_0p15_CSFET_4x35_SS404_B_2_0_0 FET1 L1 GpCircle1=gp_circle(S,2,51) Nf=12 L= nh R= Wg=100 um {t} S-PARAMETERS dls=0 ph SS_Bias=105:17.5Vds148mA/mm (N) LStabCircle S_Param fcgs=1 SP1 fgm=1 L_StabCircle RayGlobals Start=0.1 GHz fcds=1 L_StabCircle1 L_StabCircle1=l_stab_circle(S,51) Stop=150 GHz fcgd=1 MSub RayGlobals Step=0.1 GHz fgds=1 RayGlobals NDF_pos=1 Process=P82_A MSUB DisplayTemplate Stats=0: Nominal MSub1 disptemp1 StatsDate= SStabCircle H=50 um "S_Params_Quad_dB_Smith" Er=10.15 StdDev=1 S_StabCircle Mur=1 GridSnap=500 nm S_StabCircle1 Cond=4e7 RayTestV=1 uv S_StabCircle1=s_stab_circle(S,51) Hu=1.0e+036 um T=3.2 um S-PARAMETERS TanD= S_Param SP2 Start=6 GHz Stop=12 GHz Step=2 GHz NsCircle NsCircle1 NsCircle1=ns_circle(nf2,NFmin,Sopt,Rn/50,51) MaxGain MaxGain MaxGain1 MaxGain1=max_gain(S) Mu Mu Mu1 Mu1=mu(S) MuPrime MuPrime MuPrime1 MuPrime1=mu_prime(S) MaxGain1 db(s(2,1)) db(s(2,2)) db(s(1,1)) m3 m5 m4-10 db db freq, GHz db Fig. 6 ADS S-parameter simulation of the µm (1.2-mm) GaN HEMT 0.2 p100 p99 p98 p97 p96 p95 p94 p93 p92 p91 p90 p89 p88 p87 p86 p85 p84 p83 p82 p Stability p10 Swp Max 20GHz p9 p SCIR1() db Swp Min 0.5GHz GHz 15.1 db 2 GHz 1.06 amp_pa_1_2mm 8 GHz 8.78 db DB( S(1,1) ) (L) DB( S(2,1) ) (L) DB( S(2,2) ) (L) MU1() (R) MU2() (R) DB(GA()) (L) Frequency (GHz) Fig. 7 Stabilizing resistors added to the μm GaN HEMT plus broadband output match 5

14 PORT P=1 Z=50 Ohm CLIN ID=TL4 ZE=71 Ohm ZO=43 Ohm EL=45 Deg F0=4.5 GHz 1 W 2 W 3 4 TLIN ID=TL3 Z0=40 Ohm EL=47 Deg F0=4.5 GHz TLIN ID=TL2 Z0=80 Ohm EL=23 Deg F0=4.5 GHz amp_pa_1_2mm 2 GHz 17.1 db DB( S(1,1) ) (L) 7 GHz 12.5 db DB( S(2,1) ) (L) 2 GHz 1.11 DB( S(2,2) ) (L) MU1() (R) MU2() (R) DB(GA()) (L) TLIN ID=TL1 Z0=62 Ohm EL=5 Deg F0=4.5 GHz db GHz db Frequency (GHz) Fig. 8 Ideal, coupled line input match for the 2- to 7-GHz, 1.2-mm GaN HEMT power amplifier A preliminary input match including DC bias input for the gate is shown in Fig. 9. A pseudo layout of the full one-stage, 1.2-mm, 2- to 8-GHz power amplifier is shown in Fig. 10; note the large area required for the broadband input match. The resulting single-stage amplifier performance is shown in Fig. 11, with good gain at 2 GHz, dropping gradually to 10 db at 8.5 GHz. MRINDSB3 ID=MSP4 ID=TL19 ID=TL17 MRINDSB3 ID=MSP2 NS=5 L1=240 um L2=240 um L3=200 um LN=120 um AB=0 W=20 um S=10 um WB=25 um HB=2 um LB=0 um EPSB=1 TDB=0 TB=1 um RhoB=1 MCURVE2 ID=TL18 ID=TL8 W=20 um L=20 um PORT P=2 Z=50 Ohm ID=TL2 W=26 um L=1068 um SUBCKT ID=S1 NET="EM_MCLIN" MCURVE2 ID=TL PORT P=1 Z=50 Ohm 1 4 MCLIN ID=TL4 W=70 um S=60 um L=1800 um 3 W 2 1 MTEE ID=TL15 MRINDSB3 ID=MSP3 MCURVE2 ID=TL10 MRINDSB3 ID=MSP1 NS=10 2 ID=TL13 L1=885 um L2=850 um L3=850 um LN=80 um AB=90 W=70 um S=60 um WB=70 um HB=2 um LB=0 um EPSB=1 TDB=0 TB=1 um RhoB=1 ID=TL9 W 4 ID=TL1 W=56 um L=510 um ID=TL3 W=140 um L=3350 um ID=TL7 TLIN W=90 ID=TL5 um L=3700 Z0=40 um Ohm EL=48 Deg F0=4.5 GHz TLIN ID=TL6 ID=TL14 Z0=50 Ohm EL=52 Deg F0=4.5 GHz MTEE ID=TL11 2 TFRM ID=TL SUBCKT ID=S6 NET="BondPad1" 1 TFC ID=V3 TAND=0 1 SUBCKT ID=S7 NET="Svia1" Fig. 9 Broadband, compact, folded, coupled line MMIC input match for the 2- to 8-GHz, 1.2-mm GaN HEMT power amplifier 6

15 Fig. 10 Unfinished layout of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier GHz 14.8 db amp_pa_1_2mm 7 GHz 10.5 db DB( S(1,1) ) (L) DB( S(2,1) ) (L) DB( S(2,2) ) (L) MU1() (R) MU2() (R) DB(GA()) (L) GHz GHz 10.1 db db Frequency (GHz) 0 Fig. 11 Small-signal simulation of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier 7

16 With a preliminary layout and MWO simulations for a stable, broadband power amplifier from 2 to 8 GHz based on a 1.2-mm GaN HEMT, the next step was to perform nonlinear simulations using the design kit and ADS. The nonlinear HEMT model within the ADS Raytheon design kit is needed to do performance simulations. MWO schematics for the MMIC input and output circuits were translated into ADS schematics. Ideal bias tees were added to provide the DC bias as a convenience to simulating the ADS schematics (Fig. 12), though the matching circuits have the appropriate components for DC and RF decoupling. This power amplifier design would still need design rule checks (DRCs), layout versus schematic (LVS), and final EM simulations. If another fabrication opportunity appears, this is a good starting point and could be easily completed using the circuits documented in this technical report. Fig. 12 ADS simplified schematic of the 4- to 5-W, broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier A dynamic load line simulation at the center frequency of 4.5 GHz for the one-stage power amplifier at nominal DC bias is shown in Fig. 13. Performance simulations for power-added efficiency (PAE) and output power at the center frequency of 4.5 GHz are shown in Fig. 14, with output power within 0.6 db of ideal for a 1.2-mm HEMT. As an additional verification, ADS was used to repeat the smallsignal S-parameter simulations, but with the nonlinear HEMT model at the nominal DC bias. The gain seems a little higher than the previous simulations in MWO, but the return loss and gain with frequency has a similar shape, as expected. To get a measure of the losses due to the physical MMIC output, input, and matching circuits, an ADS schematic of the power amplifier using the original lossless 8

17 element input and output matching circuits was simulated. Output power and efficiency is slightly higher in comparison for the broadband, 1.2-mm HEMT power amplifier with lossless matching elements. Figure 15 shows the performance simulation at the center frequency of 4.5 GHz, for the lossless, matched, 1.2-mm HEMT single-stage power amplifier. Performance simulations for PAE and output power with an ideal lossless matching circuit at the center frequency of 4.5 GHz are shown in Fig. 16, with output power equal to that expected for a 1.2-mm HEMT and PAE of 56% 57%. A summary table (see Table 1, shown later in this section) compares the relative performance for the ideal lossless and lossy MMIC one-stage, 1.2-mm HEMT power amplifier at various frequencies, as well as a lossless 2-way parallel, combined, 2.4-mm HEMT power amplifier. Fig. 13 ADS dynamic load line simulation of the broadband (4.5-GHz), 1.2-mm HEMT power amplifier 9

18 Fig. 14 ADS performance simulation of the broadband (4.5-GHz), 1.2-mm HEMT power amplifier Forward Transmission, db m1 m1 freq= 4.000GHz db(s(2,1))= db(s(2,2)) db(s(1,1)) db(s(2,1)) Fig. 15 ADS small-signal simulation of the broadband (2- to 8-GHz), 1.2-mm GaN HEMT power amplifier freq, GHz

19 Fig. 16 ADS performance simulation of the ideal broadband (4.5-GHz), 1.2-mm HEMT power amplifier In addition to the 1.2-mm broadband power amplifier, a 2.4-mm power amplifier was implemented using 2 parallel combined 1.2-mm HEMTs. First, the ideal output match for a single 1.2-mm HEMT was transformed from a 50-Ω output match to 100 Ω so that 2 devices could be easily paralleled into a 50-Ω load. Figure 17 shows the ideal broadband output match from a single 1.2-mm HEMT transformed to a 100-Ω output match, as well as the composite schematic of the 2-way combined output match (Fig. 18). This simple lossless combiner circuit would need to be modified to supply DC bias, and there are a several easy ways to modify it. The 2- way combiner output matching circuit has the same broadband return loss, with a better than 20-dB return loss match to the ideal load from 2.7 to 7.6 GHz (Fig. 19). PORT P=1 Z=50 Ohm CAP ID=C3 C=CP1 pf IND ID=L3 L=LP1 nh CAP ID=C2 C=Cser2 pf IND ID=L4 L=5.123 nh IND ID=L2 L=2.12 nh IND ID=L1 L=-52.7 nh CAP ID=C1 C=CP2x pf RES ID=R1 R=100 Ohm Fig. 17 HEMT) MWO partial schematic for the ideal, parallel, 2-way combined circuit ( µm 11

20 Fig. 18 ADS schematic for the ideal, parallel, 2-way combined circuit (2- to 1.2-mm HEMTs) db Fig. 19 HEMT Double, tuned, ideal load match for the parallel, 2-way combined circuit vs. the single 12

21 ADS was used to simulate the performance of the broadband power amplifier as a 2-way combined (2.4 mm) HEMT power amplifier using the ideal output matching circuit from Fig. 18. The input of the 2-way combined amplifier was simulated as 2 of the coupled line, ideal, input matching circuits into a 25-Ω source. An input matching circuit into a 50-Ω source would require a redesign but should not change the gain or bandwidth of the 2.4-mm power amplifier. Output power would be expected to double (+3 db), with similar efficiency and bandwidth in comparison to the single 1.2-mm HEMT power amplifier. Figure 20 shows the performance simulation at the center frequency of 4.5 GHz, with output power equal to that expected and PAE of 55% for a lossless matched broadband, 2.4-mm HEMT single-stage power amplifier. A summary showing relative performance for the ideal lossless and lossy MMIC one-stage, 1.2-mm HEMT power amplifier at various frequencies, as well as an ideal, lossless, 2 parallel combined (2.4-mm) HEMT power amplifier are shown in Table 1. Fig. 20 ADS performance simulation of the ideal broadband (4.5-GHz), 2.4-mm HEMT power amplifier 13

22 Table 1 ADS relative performance simulations of the 1.2- and 2.4-mm broadband HEMT power amplifiers Frequency 2.5 GHz 3.0 GHz 4.5 GHz 6.5 GHz 7.0 GHz MMIC P out PAE Ideal 1 P out PAE Ideal 2 P out PAE 1.1 db, 78% 0.7 db, 85% 0.7 db, 85% 0.9 db, 81% 1.1 db, 78% 48.8% 49.4% 49.6% 42.5% 41.0% 0.9 db, 81% 0.6 db, 87% 0 db, 100% 0.3 db, 93% 0.4 db, 91% 57.0% 54.8% 56.5% 59.1% 59.3% 0.8 db, 83% 0.4 db, 91% 0 db, 100% 0.3 db, 93% 0.4 db, 91% 55.5% 53.2% 54.8% 57.6% 57.7% Losses for the MMIC output match were calculated to be a reasonable 0.3 db over most of the band, with up to a 0.5-dB loss at the low end of the band, 2.5 to 3 GHz. Additional losses on the MMIC input match would similarly affect small signal gain and PAE. The performance data were typically 3 to 4 db compressed for the Class A/B, biased power amplifier. For the ideal, 2.4-mm power amplifier, the input power level is 3 db higher, corresponding to a 3-dB higher output power, with the same large signal gain as the ideal, 1.2-mm power amplifier. Nominal performance for the MMIC 1.2-mm HEMT amplifier was within 85% (0.6 db) of expected output power with 50% PAE at 4.5 GHz. In comparison, the ideal version of the 1.2-mm power amplifier was 100% (0 db) of expected output power with 57% PAE. As expected the 2-way combined, ideal amplifier has double the output power with similar bandwidth and efficiency, showing double the power of a single 1.2-mm HEMT with 55% PAE at a comparable gain compression level. 3. Summary and Conclusion A preliminary design of a broadband, 1.2-mm HEMT power amplifier and a 2.4-mm HEMT power amplifier using Raytheon s GaN process was performed. The intent was to explore the bandwidth and performance of a Class A/B, biased, 1.2-mm HEMT power amplifier designed to maximize bandwidth, output power, and PAE over the 2- to 8-GHz band. Trying to increase the band to 2 to 10 GHz would certainly require more matching losses to extend the bandwidth. A similar 2-way combined, 2.4-mm HEMT power amplifier should achieve comparable performance based on a preliminary design using ideal, lossless matching elements. For the one-stage, 1.2-mm HEMT design, a preliminary layout was implemented, including EM simulations of critical elements such as the folded coupled line for the broadband input match. 14

23 These designs illustrate broadband, Class A/B power amplifiers using a 1.2-mm HEMT cell, which should provide good efficiency with matching network losses within 0.6 db of ideal at these frequencies at the recommended DC bias. To get these designs ready for fabrication would require additional steps to pass DRC and LVS checks, perform full EM simulations, simulate process variation effects, and perform normalized determinant function stability analyses. The Raytheon process is very capable for high-power RF amplifiers and robust lownoise amplifiers for receivers. 15

24 List of Symbols, Abbreviations, and Acronyms ADS ARL CAD DC DRC EM GaN HEMT LVS MMIC MWO PAE PDK RF Advanced Design System (CAD tool) US Army Research Laboratory computer-aided design direct current design rule checks electromagnetic gallium nitride high-electron mobility transistor layout versus schematic monolithic microwave integrated circuit Microwave Office (CAD tool) power-added efficiency process design kit radio frequency 16

25 1 DEFENSE TECHNICAL (PDF) INFORMATION CTR DTIC OCA 2 DIR ARL (PDF) RDRL DCM IMAL HRA RECORDS MGMT RDRL DCL TECH LIB 1 GOVT PRINTING OFC (PDF) A MALHOTRA 9 DIR USARL (PDF) RDRL SER 2 P AMIRTHARAJ (HC) RDRL SER E R DEL ROSARIO (1 PDF, 1 HC) A DARWISH A HUNG T IVANOV P GADFORT J PENN (1 PDF, 1 HC) E VIVEIROS J WILSON 2800 POWDER MILL RD ADELPHI MD

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