WIDE-BAND tunable filters are extensively used for

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1 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 3, MARCH A Differential 4-bit GHz RF MEMS Tunable Filter Kamran Entesari, Student Member, IEEE, and Gabriel M. Rebeiz, Fellow, IEEE Abstract This paper presents a state-of-the-art RF microelectromechanical systems wide-band miniature tunable filter designed for GHz frequency range. The differential filter, fabricated on a glass substrate using digital capacitor banks and microstrip lines, results in a tuning range of 44% with very fine resolution, and return loss better than 16 db for the whole tuning range. The relative bandwidth of the filter is % over the tuning range and the size of the filter is 5 mm 4 mm. The insertion loss is 4.1 and 5.6 db at 9.8 and 6.5 GHz, respectively, for a 1-k sq fabricated bias line. The simulations show that, for a bias line with 10- k sq resistance or more, the insertion loss improves to 3 db at 9.8 GHz and 4 db at 6.5 GHz. The measured IIP 3 level is 45 dbm for khz, and the filter can handle 250 mw of RF power for hot and cold switching. Index Terms Differential filter, digital capacitor bank, microelectromechanical systems (MEMS), RF MEMS, wide-band tunable filter. I. INTRODUCTION WIDE-BAND tunable filters are extensively used for multiband communication systems and wide-band tracking receivers. The most practical implementation is based on yttrium iron garnet (YIG) resonators. They have multioctave bandwidths and are high quality-factor resonators [1], [2]. However, they are bulky and cannot be easily miniaturized for wireless communications. They also consume considerable amount of dc power ( W), and their linearity is not high dbm. 1 An alternative to YIG filters is based on miniaturized planar filters with solid-state or microelectromechanical systems (MEMS) devices. Solid-state varactors can provide a wide tuning range, but they have loss and linearity problems at microwave frequencies [3] [5]. MEMS switches and varactors have very low loss, they do not consume any dc power, and their linearity is excellent dbm [6], [7]. There are two different types of frequency-tuning methods for MEMS-based filters, analog and digital. Analog tuning provides continuous frequency variation of the passband, but the tuning range is limited. For example, the tuning range in [8] [10] is Manuscript received April 12, 2004; revised September 9, This work was supported by the National Science Foundation under Contract ECS K. Entesari is with the Radiation laboratory, Department of Electrical Engineering and Computer Science, The University of Michigan at Ann Arbor, Ann Arbor, MI USA ( kentesar@umich.edu). G. M. Rebeiz was with the Radiation Laboratory, The University of Michigan at Ann Arbor, Ann Arbor, MI USA. He is now with the Department of Electrical and Computer Engineering, University of California at San Diego, La Jolla, CA USA ( rebeiz@ece.ucsd.edu). Digital Object Identifier /TMTT YIG tuned filters, Micro Lambda Inc., Fremont, CA. Fig. 1. Lumped model for a two-pole differential tunable filter. 4.2%, 10%, and 14%, respectively. In digital MEMS filters, discrete center frequencies and wide tuning ranges are possible. Young et al. [16], and Brank et al. [17] presented excellent tunable filters with high-frequency resolution at GHz. However, existing digital MEMS filters at microwave frequencies GHz do not have enough resolution to result in near continuous coverage of the frequency band. The MEMS filter in [11] shows four states (2-bit filter) and a 44% tuning range with poor frequency resolution. The filter in [12] has two states (1-bit filter) and can switch from 15 to 30 GHz. Other designs have two states and lower tuning range: 28.5% in [13] and 12.8% in [14]. In this paper, we will present a 4-bit digital differential tunable filter with 44% tuning range from 6.5 to 10 GHz. The frequency band is covered by 16 filter responses with very fine frequency resolution. Practically, this filter behaves like a continuous-type tunable filter. To achieve such a high tuning resolution, capacitive MEMS switches are connected in series with high- metal air metal (MAM) capacitors to make a capacitor bank. As a result, the capacitor variation can be controlled accurately by choosing the correct values for MAM capacitors. The MEMS capacitor bank is inserted in a lumped differential filter to result in a miniature GHz tunable filter. A nonlinear study of the differential tunable filter is presented in Section IV. II. DESIGN A. Design Equations Fig. 1 presents a two-pole filter suitable for a differential implementation. This filter is a practical realization of a standard Chebyshev bandpass filter with parallel resonators and -inverters [see Fig. 2(a)]. The element values for the normalized bandpass filter are calculated from (1) (2) (3) /$ IEEE

2 1104 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 3, MARCH 2005 Fig. 3. J-inverter implementation using a practical transformer. Fig. 2. (a) Standard Chebyshev bandpass filter. (b) Admittance scaling. (c) Practical transformer realization using capacitive dividers. where is the bandpass filter center frequency, is the fractional bandwidth, and and are the parallel capacitor and the admittance inverter for the normalized low-pass prototype filter [15]. The shunt inductance is inversely proportional to and for narrow bandwidths; it can be too small to be physically realizable. Therefore, the entire admittance of the filter (including source and load) can be scaled by a factor, where is an arbitrary scaling factor to result in realizable element values, and an impedance transformer must be inserted between the filter and its terminations [see Fig. 2(b)]. To realize this ideal transformer at high frequencies, a narrow-band impedance transformer is implemented using and capacitors [see Fig. 2(c)]. The values of and can be found using the two conditions at the source and load and can be calculated based on the self-inductance of the transformer and the transformer coupling factor Using (1) (9), the values of,,, and can be calculated based on,,,,, and for a specific frequency. The closed-form equations for the lumped elements of the differential filter of Fig. 1 are summarized as follows: (7) (8) (9) (10) (4) where is the differential line impedance at the input and output of the filter (100 for this design). Solving (4) gives the values of and [15] (11) (12) (13) (5) (6) is a negative capacitor, but it is always smaller than so they can be absorbed into each other and form the capacitor. Fig. 3 shows the implementation of the -inverter and parallel bandpass inductors using a practical transformer. The values of B. Design The response of the differential filter of Fig. 1 can be tuned over a wide frequency range by varying. The shape and relative bandwidth of the filter is approximately fixed due to the filter topology, i.e., capacitive tuning with an inductive inverter [1]. The input/output matching is maintained better than 16 db over the tuning range by varying. Table I presents the element values of a two-pole 5% (Chebyshev, equiripple bandwidth) GHz differential tunable filter. The filter is first designed at 10 GHz using (10) (13) with and. Next, and capacitors are varied to achieve a

3 ENTESARI AND REBEIZ: DIFFERENTIAL 4-BIT GHz RF MEMS TUNABLE FILTER 1105 TABLE I ELEMENT VALUES FOR THE TUNABLE LUMPED FILTER Fig. 5. Simulated values of C and C for different switch combinations. C TABLE II ELEMENT VALUES EXTRACTED FROM SONNET SIMULATION Fig. 4. (a) Simplified circuit model. (b) Layout for C as a capacitor bank. filter response down to 6.5 GHz using Agilent ADS. 2 To achieve a 40% tuning range or more, and. C. 3- and 4-Bit Digital MEMS Capacitors The resonant capacitor is substituted by a capacitor bank with four unit cells (Fig. 4). This results in 16 different filter responses using 16 different combinations of switches in the upand down-state positions.,, and represent MEMS switches, is half of the MAM capacitor in series with the MEMS switch, and is the parasitic capacitance coming from the layout implementation of. Each matching capacitor is composed of three unit cells, and provides enough capacitive variation to result in a well-matched circuit over the whole tuning range. Fig. 5 shows the different values of and for different combination of switches in the up- and down-states positions. These values are extracted from full-wave simulation of 2 ADS 2002, Agilent Technol. Inc., Palo Alto, CA. the capacitor bank layouts using Sonnet 3 and fitted to a model. The corresponding unit-cell values presented in Table II are calculated using, where is found from the full-wave simulations and includes all inductive (parasitic) effects. varies between 140 ff (State 0, all the switches are up) and 425 ff (State 15, all the switches are down), which is close to the range in Table I. varies from 95 (State 0) to 253 ff (State 7), which is higher than in Table I and provides a safe margin to match the return loss of the filter at lower frequencies. D. Effect of Bias Resistance on Capacitor The resonant capacitor bank quality factor has an important role in determining the unloaded of the resonators and the insertion loss of the filter. The MEMS and MAM capacitors are high- elements as measured in [18], and do not degrade the unloaded of the resonators. However, the bias line resistor has a very strong loading effect on the quality factor of (and the resonator). Fig. 6 shows the simulated quality factor of at an arbitrary frequency GHz for different bias line values. This is done using ADS, with the component values in Table II placed in the circuit model of Fig. 4. The simulated impedance is fitted to a simple circuit ( is negligible), and the quality factor is defined as [6] 3 Sonnet 8.52, Sonnet Software Inc., North Syracuse, NY. (14)

4 1106 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 3, MARCH 2005 Fig. 6. Simulated resonant capacitor quality factor for different switch combinations, and bias resistances. TABLE III SIMULATED CENTER FREQUENCIES AND THE CORRESPONDING 16 DIFFERENT STATES FOR C AND C.THE INDEXES U/L REFER TO THE UPPER (U) AND LOWER (L) MATCHING CAPACITORS IN FIG. 1 Fig. 7. Simulated: (a) insertion loss and (b) return loss of the tunable two-pole GHz filter. where is the operation frequency, is the total series resistance including MEMS switches, MAM capacitors, and bias line resistances, and is the equivalent capacitance. The quality factor doubles if the bias line resistance changes from 1 to 10 k sq. For higher bias line resistances (up to 1 M sq), the quality factor improvement is only 20%. E. Tunable Filter Simulations The resonant capacitor layout can change both the transformer coupling factor and inductor values. The amount of desired coupling and inductance is adjusted by changing the length of the inductors and the distance between the two inductors when the coupled resonators are simulated together in Sonnet. The simulated of the wide microstrip lines ( m, m) on the glass substrate is around 78 at 8 GHz. The whole filter response is then simulated in ADS by cascading -parameters of the matching capacitors and resonators. The simulated center frequencies and the corresponding states of and are presented in Table III. Fig. 7 shows the insertion loss for 16 different states with k sq and is 4 and 5.6 db at 9.8 and 6.5 GHz, respectively. The higher insertion loss at 6.5 GHz is due to the low-resistivity bias line, which has a strong loading effect when the switches are all in the down-state position. If the value of increases to 10 k sq, the insertion loss improves to 3 db at 9.8 GHz and 4 db at 6.5 GHz. The effective unloaded of the two-pole filter is 45 and 60 for k sq and 10 k sq, respectively, at 8 GHz. III. FABRICATION AND MEASUREMENT A. Fabrication, Implementation, and Biasing The tunable filter is fabricated on a 500- m glass substrate ( and ) using MEMS switches and microstrip lines using a standard RF MEMS process developed at The University of Michigan at Ann Arbor [18], [19]. The MEMS capacitive switch is based on a sputtered gold layer and is suspended m above the pull-down electrode. The dielectric Si N layer is thick and the bottom electrode thickness is 6000 (underneath the bridge). The MAM capacitors are suspended 1.5 above the first metal layer. The microstrip conductor, bridge anchor, and top plate of MAM capacitors are electroplated to 2- m thick using a low-stress gold solution. The bias lines are fabricated using a thick SiCr layer with a resistivity of 1 k sq.

5 ENTESARI AND REBEIZ: DIFFERENTIAL 4-BIT GHz RF MEMS TUNABLE FILTER 1107 Fig. 8. Photograph of a unit cell in a digital capacitor bank. Fig. 9. Photograph of the complete GHz filter. The center frequency of the tunable filter is directly related to the accuracy of the MAM capacitors (obtained versus designed values). Since the MAM capacitors are quite large and fully electroplated and the sacrificial layer underneath them is uniform, one can build these capacitors quite uniformly throughout the entire wafer with an accuracy around 5% in university laboratories. This can be clearly seen in the agreement between measured versus simulated center frequencies [see Fig. 12(a)]. The photograph of a unit cell in a digital capacitor bank is shown in Fig. 8. The width, length, and thickness of the bridge are 70, 280, and 0.8 m, respectively, and the gap is 1.5 m for the bridge and MAM capacitors. The bottom plate of one of the MAM capacitors is connected to the thin-film resistor to bias the bridge. The release height of the MEMS bridge and MAM capacitor is 1.5 m measured by a light-interferometer microscope. The measured pull-in voltage is V with a corresponding spring constant of N/m and a residual stress of MPa. The mechanical resonant frequency and quality factor of the switch are khz and, respectively [6]. The photograph of the complete GHz filter is shown in Fig. 9. It is composed of a transformer in the middle, two resonant capacitors, each with four unit cells, and four matching capacitors, each with three unit cells. Each switch has a separate SiCr dc-bias line for independent control. The center conductor of each capacitor bank is connected to the dc ground pad through the SiCr line. The filter is excited using differential input and output lines, which are compatible with Fig. 10. Measured: (a) insertion loss and (b) return loss of the tunable two-pole GHz filter. a ground signal ground signal ground (GSGSG) differential probe with a pitch of 150 m. B. Measurements The tunable filter is measured using a differential test setup: two hybrid couplers and two differential probes are used to provide differential excitation for the filters. As is well known, only the odd mode should be generated for differential circuits. The phase and amplitude imbalance for the couplers should be as small as possible. (Gain Bal. 2 db, Phase Bal. 10 ) for accurate differential measurements [20]. The measurement results are shown in Fig. 10 for 16 different states. The insertion loss [see Fig. 10(a)] is 4.1 and 5.6 db at 9.8 and 6.5 GHz respectively, and the relative bandwidth is approximately fixed for the whole tuning range, as expected from the simulation results. The return loss [see Fig. 10(b)] is always better than 16 db for the whole tuning range. Fig. 11 compares the measured and simulated insertion loss for two arbitrary states at 8.6 GHz (State 4) and 7.1 GHz (State 13). The simulated and measured responses agree very well. For the case of k sq, the simulated insertion loss is 1 db better at 8.6 GHz and 1.4 db better at 7.1 GHz, as compared with the measurements for k sq. The measured response of

6 1108 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 3, MARCH 2005 Fig. 13. Experimental setup for intermodulation measurements (1f = f 0 f ). Fig. 11. Comparison between the measured and simulated insertion loss for two arbitrary states at 7.1 (State 13) and 8.6 GHz (State 4). Fig. 14. Nonlinear measurements at V = 0 V. (a) The fundamental and intermodulation components versus the input power and the two-tone IM versus the beat frequency. (b) Filter insertion loss for different values of input power. Fig. 12. (a) Simulated and measured center frequency and loss. (b) Measured relative bandwidth of the 16 filter responses ( %). a fabricated filter without any bias lines also confirms that the insertion loss improves by 1 db at 9.8 GHz (all the switches are electroplated in the up-state position) to 1.6 db at 6.5 GHz (all the switches are electroplated in the down-state position). The simulated and measured center frequency and loss for each of the 16 different states is presented in Fig. 12(a). Fig. 12(b) shows the relative bandwidth variation for all responses, and is 5.5% at 9.8 GHz (State 0) and 4.7% at 6.5 GHz (State 15). IV. NONLINEAR CHARACTERIZATION The nonlinear analysis of MEMS switches, varactors, and tunable filters has been presented in [7]. This analysis shows that the mechanical force acting on the MEMS bridge is proportional to across the bridge. Under high RF drive conditions, the MEMS bridge capacitance variation results in a nonlinear behavior of the tunable filter. In the case of two-tone excitation, the mechanical force at the beat frequency changes the capacitance of the bridge and the presence of the input tones across the variable capacitor generates the third-order intermodulation. Fig. 13 shows the setup to measure the intermodulation components at the output of the tunable filter. Two tones generated by two synthesizers are combined, and the combined signal is delivered to the MEMS filter differentially through the baluns (3-dB couplers) and differential RF probes. The output signals are measured using a spectrum analyzer. Fig. 14(a) shows the measured output power for the fundamental and the intermodulation components for several

7 ENTESARI AND REBEIZ: DIFFERENTIAL 4-BIT GHz RF MEMS TUNABLE FILTER 1109 distorted. This filter can, therefore, handle an input power up to 24 dbm (250 mw). If a higher bias MEMS switch is used V, the RF power handling will increase to 1 W before self-biasing becomes an issue. The maximum power handling of the filter can be well predicted by studying the voltage and current across the MEMS switch for an input power of 200 mw (Fig. 15). Using a linear model in Agilent ADS, the rms voltage is calculated to be 12.5 V across, which is the switch with the largest loading MAM capacitor [State 0; ]. The corresponding height change in the MEMS switch is m, and results in a 0.97-fF change in the switch and a 0.7-fF change in the overall bit-4 capacitance. This is small enough that it has no effect on the frequency response [see Fig. 14(b)]. In the down-state position, the rms current in is 81 ma for m, and is well within the acceptable region [6]. However, the rms voltage across the is 3.5 V, which is just below the hold-down voltage of 5 V (the switch remains down even if the dc actuation voltage is removed). It is for these reasons that we predict a 250-mW power-handling capacity for hot switching (RF is never shut off). Fig. 15. Simulated rms V and I for the input capacitor banks (C, C ). The values for the output capacitive banks are lower by a factor of two due to the filter loss. values of. The measured is 45 dbm for khz. The measurement is in the up-state position since this state gives the worse products. Tunable filters with diode varactors have much lower values of (12 dbm in [3] and 28 dbm in [4]). Fig. 14(a) also shows the intermodulation component versus the difference frequency between input tones for dbm (no bias voltage on the bridges). The intermodulation component follows the mechanical response of the bridge, and the level drops by 40 db/decade for, which is in agreement with theory [7]. Fig. 14(b) shows the measured insertion loss of the filter for three different input power levels ( and dbm). Due to the spectrum analyzer (a noncalibrated system), the measured insertion loss value is not the same as Fig. 10. The center frequency is at 9.8 GHz and the bias voltage is zero [State 0; ]. For dbm, there is no self-actuation effect [6], [7], and the filter response does not change with increasing the level of. For dbm, the level of the RF voltage across the MEMS bridges is large enough to self-bias the bridges. As a result, the shape of the filter gets V. CONCLUSION This paper has demonstrated a wide-band tunable filter on a glass substrate from 6.5 to 9.8 GHz (44% tuning range). A novel lumped differential topology is used to miniaturize the filter structure. Resonant capacitor banks with four unit cells (MEMS capacitive switches in series with high- MAM capacitors) result in 16 different filter responses next to each other with very fine tuning resolution like a continuous tunable filter. Matching capacitor banks with three unit cells result in a return loss better than 16 db over the whole band. The measured results are very close to full-wave simulations. The loss of the filter can be improved by db using 10-k sq bias lines or higher instead of a 1-k sq fabricated SiCr line. We believe that the lumped capacitor design can be extended to 18 GHz on glass substrates. Above this frequency, the finite size of the MEMS bridge results in a large parasitic inductance, which limits the tuning performance of the 3- or 4-bit capacitors. REFERENCES [1] G. L. Matthaei, E. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures. Norwood, MA: Artech House, [2] W. J. Keane, YIG filters aid wide open receivers, Microwave J., vol. 17, no. 8, Sep [3] S. R. Chandler, I. C. Hunter, and J. C. Gardiner, Active varactor tunable bandpass filters, IEEE Microw. Guided Wave Lett., vol. 3, no. 3, pp , Mar [4] A. R. Brown and G. M. Rebeiz, A varactor-tuned RF filter, IEEE Trans. Microw. Theory Tech., vol. 48, no. 7, pp , Jul [5] I. C. Hunter and J. D. Rhodes, Electronically tunable microwave bandpass filters, IEEE Trans. Microw. Theory Tech., vol. 30, no. 9, pp , Sep [6] G. M. Rebeiz, RF MEMS Theory, Design, and Technology. New York: Wiley, [7] L. Dussopt and G. M. Rebeiz, Intermodulation distortion and power handling in RF MEMS switches, varactors and tunable filters, IEEE Trans. Microw. Theory Tech., vol. 51, no. 4, pp , Apr [8] H. T. Kim, J. H. Park, Y. K. Kim, and Y. Kwon, Millimeter-wave micromachined tunable filters, in IEEE MTT-S Int. Microwave Symp. Dig., vol. 3, Jun. 1999, pp

8 1110 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 3, MARCH 2005 [9], Low-loss and compact V -band MEMS-based analog tunable bandpass filters, IEEE Microw. Wireless Compon. Lett., vol. 12, no. 11, pp , Nov [10] A. Abbaspour-Tamijani, L. Dussopt, and G. M. Rebeiz, Miniature and tunable filters using MEMS capacitors, IEEE Trans. Microw. Theory Tech., vol. 51, no. 7, pp , Jul [11] A. Pothier, J. C. Orlianges, E. Zheng, C. Champeaux, A. Catherinot, D. Cross, P. Blondy, and J. Papapolymerou, Low loss 2-bit bandpass filters using MEMS DC contact switches, IEEE Trans. Microw. Theory Tech., vol. 53, no. 1, pp , Jan [12] D. Peroulis, S. Pacheco, K. Sarabandi, and L. P. B. Katehi, Tunable lumped components with applications to reconfigurable MEMS filters, in IEEE MTT-S Int. Microwave Symp. Dig., May 2001, pp [13] B. Lakshminarayanan and T. Weller, Tunable bandpass filter using distributed MEMS transmission lines, in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2003, pp [14] E. Fourn, A. Pothier, C. Champeaux, P. Tristant, A. Catherinot, P. Blondy, G. Tanne, E. Ruis, C. Person, and F. Huret, MEMS switchable interdigital coplanar filter, IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp , Jan [15] I. Hunter, Theory and Design of Microwave Filters. London, U.K.: IEE, [16] R. M. Young, J. D. Adam, C. R. Vale, T. T. Braggins, S. V. Krishnaswamy, C. E. Freidhoff, S. H. Talisa, E. Capelle, R. Tranchini, J. R. Fende, J. M. Lorthioir, and A. R. Torres, Low-loss bandpass RF filter using MEMS capacitance switches to achieve a one-octave tuning range and independently variable bandwidth, in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2003, pp [17] J. Brank, J. Yao, A. Malczewski, K. Varian, and C. L. Goldsmith, RF MEMS-based tunable filters, Int. J. RF Microwave Computer-Aided Eng., vol. 11, pp , Sep [18] J. S. Hayden and G. M. Rebeiz, Very low loss distributed X-band and Ka-band MEMS phase shifters using metal air metal capacitors, IEEE Trans. Microw. Theory Tech., vol. 551, no. 1, pp , Jan [19] J. B. Rizk and G. M. Rebeiz, W -band microstrip RF-MEMS switches and phase shifters, in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2003, pp [20] K. Entesari, T. Vaha-Heikkila, and G. M. Rebeiz, Miniaturized differential filters for C- and Ku-band applications, in Eur. Microwave Conf. Dig., Oct. 2003, pp Kamran Entesari (S 03) received the B.S. degree in electrical engineering from Sharif University of Technology, Tehran, Iran, in 1995, the M.S. degree in electrical engineering from Tehran Polytechnic University, Tehran, Iran, in 1999, and is currently working toward the Ph.D. degree in electrical engineering (with an emphasis on applied electromagnetics and RF circuits) at The University of Michigan at Ann Arbor. His research includes RF MEMS for microwaves and millimeter-wave applications, microwave tunable filters, and packaging structures. Gabriel M. Rebeiz (S 86 M 88 SM 93 F 97) received the Ph.D. degree in electrical engineering from the California Institute of Technology, Pasadena. He is a Full Professor of electrical engineering and computer science (EECS) with the University of California at San Diego, La Jolla. He authored RF MEMS: Theory, Design and Technology (New York: Wiley, 2003). His research interests include applying MEMS for the development of novel RF and microwave components and subsystems. He is also interested in SiGe RFIC design, and in the development of planar antennas and millimeter-wave front-end electronics for communication systems, automotive collision-avoidance sensors, and phased arrays. Prof. Rebeiz was the recipient of the 1991 National Science Foundation (NSF) Presidential Young Investigator Award and the 1993 International Scientific Radio Union (URSI) International Isaac Koga Gold Medal Award. He was selected by his students as the Eta Kappa Nu EECS Professor of the Year. In October 1998, he was the recipient of the Amoco Foundation Teaching Award, presented annually to one faculty member of The University of Michigan at Ann Arbor for excellence in undergraduate teaching. He was the corecipient of the IEEE 2000 Microwave Prize. In 2003, he was the recipient of the Outstanding Young Engineer Award of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). He is a Distinguished Lecturer for the IEEE MTT-S.