Implementation of Wideband Digital Beam Forming in the E-band

Transcription

1 Ipleentation of Wideband Digital Bea Foring in the E-band Val Dyadyuk, Xiaojing Huang, Leigh Stokes, Joseph Pathikulangara ICT Centre, CSIRO, PO Box 76, Epping, NSW, Sydney, Australia This paper reports the test results of a sall-scale prototype that ipleents a digitally bea-fored phased antenna array in the E-band. A four-channel dual-conversion receive RF odule for 7-76 GHz frequency band has been developed and integrated with a linear end-fire antenna array. Wideband frequency-doain angle-of-arrival estiation and bea foring algoriths were developed and ipleented using Gbps OFDM QPSK sybols. Measured perforance is very close to the siulated results and experiental data for an analogue-bea-fored array. This work is a stepping stone towards practical realization of larger hybrid arrays in the E-band. Keywords: Active Array Antennas and Coponents, Hybrid and Multi-Chip Modules, Millieter Wave Radio Counication Corresponding author: V. Dyadyuk; eail: val.dyadyuk@csiro.au; phone: I. INTRODUCTION High data rate illieter-wave counication systes are of growing iportance to the wireless industry. Future obile and ad-hoc counications networks will require higher bandwidth and longer range. An ad-hoc or obile (e.g. Inter-aircraft) network that relies on high gain antennas also requires bea scanning. With the advance in digital signal processing techniques, the adaptive antenna array is becoing an essential part of wireless counications systes. The use of adaptive antenna arrays for long-range illieter-wave ad-hoc counication networks is particularly critical due to increased free space loss and reduced level of practically achievable output power []. Although pure digital bea foring allows for producing output signals with the axiu SINR, ease of on-line calibration and generation of any antenna patterns siultaneously, it is ipractical for large wideband arrays due to two ajor reasons. Firstly, it is too costly since the cost of digital data processing is proportional to bandwidth and increases, at least, linearly with the nuber of eleents. Secondly, the sall separation of array eleents in the E-band (7-86 GHz) leaves little roo at the back of the array for connection of each RF chain associated with individual array eleents to a digital bea forer. A typical analogue RF chain is tightly packed behind

2 each antenna eleent and includes a low noise aplifier (or power aplifier), frequency converter, local oscillator (LO), as well as the interediate frequency () or baseband circuitry. Each of these chains would require a nuber of DC, and control circuit interfaces. Therefore, a hybrid approach [-3] where digital bea foring technique is applied to a saller nuber of units (analogue bea fored sub-arrays) is preferable. This provides a significant saving in both the aount of digital signal processing and the nuber of physical connections between the RF front end and digital bea forer. A sall scale E- band phased antenna array has been developed [] for an experiental verification of the hybrid bea foring concepts [, 3]. Analogue bea foring of this array using 6-bit phase shifters and attenuators at has been reported earlier in [4, 7]. In this paper, we report the test results of an E-band prototype that ipleents a digital bea-foring phased antenna array. II. E-BAND PHASED ANTENNA ARRAY PROTOTYPE The prototype has been developed to deonstrate a counications syste with gigabit per second data rates using an electronically steerable array as an initial step towards fully ad-hoc counications systes. The prototype configuration is flexible and can be used for experiental verification of both analogue and digital bea foring algoriths. Fig. shows the equipent configuration for digital bea foring experients. The steerable receive array is ounted on a rotator providing echanical steering in the aziuth plane for the array pattern easureent. Rotator 3 4 A/D converters Rx RF odule Rx Data acquisition (FPGA) Data processing (PC) LO sources Digital odulator Tx RF odule a) b) Fig.. a) Block diagra of the test setup; b) Photograph of the RF odule assebly where: is the antenna array; is the LO input; 3-6 are outputs. to baseband frequency conversion was ipleented in the receive (Rx ) and transit (Tx ) odules. The Rx odule has been developed in two versions. For digital bea foring (DBF), each of the analogue signals was digitised and streaed to a PC for data processing and bea foring as shown in Fig... For analogue bea foring (ABF) [4, 7], the phase and agnitude of each channel was controlled by 6-bit phase shifters and attenuators and a cobined signal was de-odulated using a ode reported in [5]. This Tx 5 6

3 replaces the digitizer shown in Fig..a for the DBF configuration. The deonstrator also includes a digital odulator reported in [5] and a single-channel transitter []. BB Sub-haronic down-converter _A MXR LO- Array Eleent A to A/D BB BB LO- LO- LO- Splitter LO- AMP Sub-haronic down-converter _A LO- MXR AMP Sub-haronic down-converter _A3 MXR LO- AMP Array Eleent A Array Eleent A3 BB Rx odule LO- Splitter _A4 LO- Sub-haronic down-converter MXR AMP Array Eleent A4 Rx RF odule. Fig.. Block diagra of the receiver configured for DBF experients. The transit odule has been built using the up-converter [8], a coercial low-pass filter, a ediu power aplifier and a corrugated horn antenna. Bench test results have been reported in []. The ain functional block of the prototype is a four-channel receive RF odule [, 4] integrated with a linear end-fire Quasi-Yagi antenna array [6, 7]. Array eleent spacing was (or.48 wavelengths at the carrier frequency) to suppress appearance of grating lobes for scanning angles up to ± 4 degrees. The RF odule uses sub-haronic frequency converters at the LO frequency of 38 GHz. For each channel we have used a cobination of CSIRO and coercial off-the-shelf MMICs siilar to those reported earlier for a single-channel receiver [8, 9]. The pre-aplifiers, interconnecting, atching, and group delay equalization circuits have been developed using a standard coercial thin-fil process on ceraic substrate. Fig..b shows a photograph of the assebled RF odule. It includes 6 MMICs, types of icrowave boards (on 7u Aluina substrate), 4 icrowave passives, and about 4 wire-bond connections. The receiver is usable over the frequency range of 7 to 76 GHz at the sub-haronic LO of 38 to 39 GHz and interediate frequency to 7 GHz. Typical conversion gain was 6 ± db over the operating RF and frequency range of GHz and GHz respectively. The axiu agnitude ibalance between each of four channels was below ±.5 db. Bench test results for the receive odule have been reported in []. The to baseband ( - GHz) frequency conversion was ipleented in the Rx odule [7].

4 The digitizer consists of an Analogue-to-Digital Converter (ADC) and a Field Prograable Gate Array (FPGA). The ADC is an EV8AQ6 (EV Seiconductors) 8-bit device. This converter is used in dual channel ode, each channel sapling at Giga saples per second. The ADC has an effective nuber of bits (ENOB) of 6 bits at the frequency of operation and input bandwidth of GHz. The FPGA for signal processing is a Xilinx V5SX95T. The experiental setup uses two such digitizers as shown in Fig. 3 giving a total of four ADCs and two FPGAs. The interconnections labelled 3 in the photograph Fig. have enough bandwidth to transport digital signals after pre-processing on one FPGA to the other for digital bea foring. In this experient, the saples are captured on the FPGA internal eory and uploaded to a PC for processing. Fig. 3. Photograph of the digitizer odule assebly where and are two identical digitizers, 3 is interconnecting cable and 4 is a sapling oscillator. All easureents were conducted in the CSIRO far field anechoic chaber as shown in Fig. 4. Transitter, digital odulator and control equipent were located on the outside of the chaber. The available signal to noise ratio was above 33dB for the easureent distance up to 6, but ost of the tests were conducted at the distance of. to iniize unwanted reflections fro the walls and ceiling of the chaber. Fig. 4. Syste test setup in the far field anechoic chaber where is the receive array RF odule, is the rotator and 3 is the transit antenna aperture, 4 is the receive antenna array asked with absorbers 5.

5 III. DIGITAL BEAM FORMING ALGORITHM A) Received Signal Model Consider a linear wideband array with M eleents and eleent spacing d. Denoting the received baseband signal fro the th eleent as x ( t), the frequency doain odel of the received signal can be expressed as X f + fc π j dsinθ fc λc ( f ) = S( f ) H ( f ) P( f θ ) e + Z ( f ),, for B f, =,,..., M, () where X ( f ) is the Fourier transfor of x ( t), S ( f ) is the inforation-bearing reference signal in the frequency doain, H ( f ) is the channel frequency response of the RF chain for the th eleent, ( f,θ ) P is the eleent radiation pattern, f c and λ c are the frequency and wavelength of the RF carrier respectively, Z ( f ) is zero-ean coplex Gaussian noise, and B is the bandwidth of the signal. For digital ipleentation, the received tie doain signals ( t) sapled at t = nt, i.e., x [ n] x ( nt ) x, =,,..., M, are = is the received signal sequence fro the th eleent, where T = is the sapling period, and divided into blocks of size N. Each block is B converted into the frequency doain by fast Fourier transfor (FFT). The discrete odel is then expressed as X f k + f c π j dsinθ f c λc [ k] = S[ k] H [ k] P( f θ ) e + Z [ k] where the index k represents a discrete frequency f k = k kb N,, for k =,, L, N, (), ( k N ) X [ k], S [ k], H [ k], and [ k] N B, N for k =,, L, N N for k =, +, L, N Z are the th received frequency-doain signal, reference signal, frequency response, and noise at discrete frequency f k. B) Frequency Doain Beaforing In order to achieve axiised signal to noise ratio, the frequency doain digital beaforing is perfored as follows. First, the signal sequence received at each eleent is synchronised, and the cobined channel frequency response together with the phase shift introduced by the incident angle (i.e., H [ k] P( f, θ ) k e f k + fc j fc π dsinθ λ c (3) as a whole) is estiated using the training sequence during the preable period. Then, the synchronised signal after

6 the preable is converted into the frequency doain by FFT, and the channel frequency response at each discrete frequency is copensated by the inverse of the channel frequency response. Finally, the copensated frequency doain signals for all eleents are sued up and the cobined signal is converted into tie doain by inverse FFT (FT). To obtain the angle of arrival (AoA) inforation or for a bea for a specified incident angle, the receiver array needs to be calibrated. This calibration can be done for each eleent after copensation of the channel frequency response [ k] H only. [ k] H can be estiated in advanced fro the signal preable received at degree incident angle for each eleent. The frequency doain beaforing at a specified incident angle is perfored as follows. First, the signal sequence received at each eleent is synchronised using the preable. Then, the synchronised signal after the preable is converted into the frequency doain by FFT followed by calibration. After calibration, the frequency doain signals are weighted with coefficients W ( f ) k f k + fc π j d sinθ fc λc = e (4) where θ is the incident angle at which a bea is to be fored. The weighted signals are finally sued up and converted into the tie doain by FT to obtain the beafored output signal. The structure of this frequency doain digital beaforer is shown in Fig. 5. x (t) Received signal fro eleent = x M- (t) Received signal fro eleent =M- A/D A/D FFT Synchronization and Calibration FFT X [] X [] X [N-] X - [] X - [] X - [N-] W (f ) W (f ) W (f N- ) W M- (f ) W M- (f ) W M- (f N- ) FT Beafored output Fig. 5. Frequency-doain digital bea forer structure.

7 IV. TEST RESULTS The DBF experients were conducted by transitting a Gbps OFDM test signal with N=8 subcarriers used in the B=.5 GHz bandwidth. QPSK is used for data odulation. To facilitate synchronization and channel estiation, a preable coposed of 5 predefined OFDM sybols is pre-pended before the inforation bearing reference sequence. The signal forat is shown in Fig. 6, where s denotes the predefined training sequence and s,, s n denote the reference sequence coposed of n OFDM sybols with pseudo-rando data odulated by QPSK. s s s - s - s s s n Preable Fig. 6. Transitted signal forat. To reduce hardware cost as well as overcoe the practical ipairents involved in frequency conversion, such as I/Q ibalance, the received RF signal is first converted into signal with centre frequency.5 GHz and thus only one A/D converter operating at GHz is necessary. Each OFDM sybol has 5 saples in the tie doain after A/D conversion at. The digital coplex baseband signal is then obtained through digital down-conversion. The receive array was rotated in the aziuth plane fro -75 to 75 in step. For each aziuth angle the received signals were digitized, captured on the FPGA internal eory, uploaded to a PC and processed as described in Section III.B. Fig. 7 shows the transitted digital signal with 5 saples. The first 5 OFDM sybols for the preable, followed by 5 rando reference sybols. The received signals fro the 4 channels in the array with incident angle degree, each having 6384 saples, are captured at the receiver FPGA after A/D conversion, which are shown in Fig Nuber of Saples Fig. 7. Transitted digital signal.

8 .5.5 Nuber of Saples x Nuber of Saples x Nuber of Saples x Nuber of Saples x 4 Fig. 8. Captured 4 channel received signals at aziuth Nuber of Subcarriers 5 5 Nuber of Subcarriers Nuber of Subcarriers 5 5 Nuber of Subcarriers Fig. 9. frequency response for each channel at aziuth.

9 3 3 Phase (radian) Phase (radian) 5 Nuber of Subcarriers 5 Nuber of Subcarriers 3 3 Phase (radian) Phase (radian) 5 Nuber of Subcarriers 5 Nuber of Subcarriers Fig.. Phase frequency response for each channel at aziuth. After synchronization, channel estiation is perfored for each channel using the fifth OFDM sybol in the preable. The agnitude frequency response and phase frequency response for each channel are shown in Fig. 9 and Fig. respectively. The average phases (represented as coplex nubers) for the 4 channels are calculated as j.8787, -.5+j.859, , and j.899, which are used for calibration. A separate ABF experient was also conducted at selected incident angles using the sae antenna array for coparison and verification of the DBF results. Fig. copares the E- plane co-polar array radiation patterns at aziuth coputed by the DBF and easured for the ABF [7]. In both experients the calibration was perfored at zero degree aziuth angle. For the ABF [7], the array has been calibrated by cancelling the ain bea to obtain a null at aziuth. The null ABF calibration reference is also shown in Fig.. The E-plane co-polar radiation patterns of the DBF array for a selection of other positive and negative aziuth angles are shown in Fig. with the agnitude noralized to the axiu of the ain bea at zero degree aziuth angle.

10 5 DBF to deg [this work] ABF to deg [7] Siulated at deg [7] ABF NULL refernce [7] -5 Mag, db Aziuth, deg Fig.. Coparison of the E-plane co-polar radiation patterns for digitally (DBF) and analogue bea fored (ABF) array [7] at aziuth. Measured E-plane co-polar radiation patterns for the ABF array [7] are appended to the coputed DBF results shown in Fig.. Measured array gain was 9.5 dbi for steering angles below and reduced to approxiately 8.4 dbi at the steering angle of ± 35. Obtained DBF results were in a very close agreeent with those easured for the ABF array and EM siulation predictions [7] for steering angles within ± 4. The agnitude and angle of the DBF and ABF ain bea patterns were within db and degree respectively. Bea steering accuracy of deg has been achieved by both DBF and ABF ethods. DBF to 5deg ABF to 5deg DBF to deg DBF to -5deg ABF to -5deg DBF to -deg ABF to deg DBF to 6deg ABF to 6deg ABF to -deg DBF to -6deg ABF to -6deg 5 DBF to deg ABF to deg DBF to 9deg ABF to 9deg DBF to 35deg ABF to 35deg 5 DBF to -deg ABF to -deg DBF to -9deg ABF to -9deg DBF to -35deg ABF to -35deg -5-5 Mag, db - -5 Mag, db Aziuth, deg Aziuth, deg Fig.. E-plane co-polar radiation patterns for digitally (DBF) and analogue bea fored (ABF) array [7] for a selection of positive and negative aziuth angles.

11 V. CONCLUSION A steerable E-band receive array deonstrator that ipleents a four-eleent linear antenna array has been tested using a wideband frequency-doain digital bea foring algorith. Obtained DBF array patterns were very close to those easured for the analogue bea fored array and EM siulated predictions. Bea steering accuracy of deg has been achieved for steering angles within ± 4. Deonstrated wideband adaptive digital bea foring along with validated phase-only ABF at can be used for hybrid bea foring of larger arrays. To our knowledge, this work represents the first experiental results on the digitally bea fored antenna array in the E-band. REFERENCES [] Dyadyuk, V.; Guo, Y.J.: Towards ulti-gigabit ad-hoc wireless networks in the E-band, Global Syposiu on Millieter Waves (GSMM9), Sendai, Japan, 9. [] Huang, X.; Guo, Y.J.; Bunton, J.D.: A hybrid adaptive antenna array. IEEE Tran. on Wireless Counications, vol. 9, no. 5 (), [3] Huang, X.; Guo, Y.J.; Bunton, J.D.: Adaptive AoA Estiation and Beaforing with Hybrid Antenna Arrays, IEEE VTC-Fall, Alaska, USA, 9. [4] Dyadyuk, V.; Stokes, L.: Wideband Adaptive Bea Foring in the E-band: Towards a Hybrid Array, Global Syposiu on Millieter Waves, Incheon, Korea,. [5] Dyadyuk, V.; Bunton, J.D.; Pathikulangara J.; et al.: A Multi-Gigabit M-Wave Counication Syste with Iproved Spectral Efficiency, IEEE Trans. on MTT, Vol. 55, Issue, Part (7), [6] Deal, W. R.; Kaneda, N.; Sor, J.; Qian, Q.Y.; Itoh, T.: A new quasi-vagi antenna for planar active antenna arrays, IEEE Trans. on MTT, Vol. 48, No.6 (), [7] Dyadyuk, V.; Stokes, L.; Nikolic, N.; Weily, A.R.: Deonstration of Adaptive Analogue Bea Foring in the E-band, Journal of the Korean Institute of Electroagnetic Engineering & Science, Vol., No. 3 (), [8] Dyadyuk, V.; Stokes, L.; Shen, M.: Integrated W-band GaAs MMIC Modules for Multi- Gigabit Wireless Counication Systes, Global Syposiu on Millieter Waves, Nanjing, China, 8. [9] Dyadyuk, V.; Archer, J.W.; Stokes, L.: W-Band GaAs Schottky Diode MMIC Mixers for Multi-Gigabit Wireless Counications, In: Advances in Broadband Counication and Networks, River Publishers, Denark, 73-3.

12 Val Dyadyuk received B.Sc. and M.Sc. degrees in electrical engineering in 968 and 97 respectively fro Kharkov Institute for Radio Electronics in Ukraine. He is currently the Research Tea Leader in Microwave Systes at CSIRO ICT Centre, Sydney. Previously he held positions of the Head of the Microwave Engineering Branch at Institute for Radio Physics and Electronics, NASU, Ukraine, and Director of Research at the SCAD Scientific & Industrial Group (Kharkov, Ukraine). Mr. Dyadyuk is a recipient of the Engineers Australia (Sydney Division) R&D Excellence Award 7, and a joint recipient of the CSIRO Chairan's Medal 7, and the Australian Engineering Excellence Award 7 for exceptional research in gigabit wireless counications. He has published three book chapters and over 4 journal and conference papers, and holds ten patents. His research interests include illieter-wave wireless counication systes, EM siulations and design of GaAs and InP MMIC s and various icrowave circuits. Xiaojing Huang received his Bachelor of Engineering, Master of Engineering, and Ph.D. in electronic engineering degrees fro Shanghai Jiao Tong University, China, in 983, 986, and 989 respectively. Fro 989 to 994, Dr. Huang worked in the Electronic Engineering Departent of Shanghai Jiao Tong University as a Lecturer since 989 and an Associate Professor since 99. For 994 to 997 he was the Chief Engineer with Shanghai Yang Tian Science and Technology Corporation Liited, China. In he was a Senior and Principal Research Engineer at Motorola Australian Research Centre in Sydney. Fro 4 to 9, Dr. Huang was an Associate Professor in the School of Electrical, Coputer and Telecounications Engineering, University of Wollongong, Australia. In March, 9, he joined the CSIRO ICT Centre, Australia, as a Principal Research Scientist. His research interests are in counications theory, digital signal processing, and wireless counications networks. Leigh Stokes received B.A. degree in econoics fro Macquarie University, Sydney, Australia in 978, Electronics Engineering Certificate fro North Sydney TAFE in 988, and Post-Graduate Certificate in project anageent fro University of Technology, Sydney, Australia in 3. He was the Principal Technical Officer in charge of aintenance (99-998) at Waverley Radio Terinal, for Telstra in Sydney, which housed digital radio links. He joined the CSIRO ICT Centre, Sydney, Australia in. Mr. Stokes is the Manager for the GHz Testing Facility for illieter wave easureents including on-wafer probing of MMICs. He is also responsible for integration of the research prototypes, test setup, and developent of software for autoated easureents. Mr. Stokes is a joint

13 recipient of a Millieter-Wave Best Paper Award fro the 9th Topical Syposiu on Millieter Waves (TSMMW 7) and a joint recipient of CSIRO Chairan s Medal in 7 for exceptional research in gigabit wireless counications. Joseph Pathikulangara received a B.E. degree in electronics and counication engineering fro Indian Institute of Science in 984 and M.Tech degree in coputer science and engineering fro IIT, Bobay in 99. He was with the Defense Research and Developent Organization in India developing coand, control and counication systes for issile and EW projects between 984 and 995. Since joining CSIRO in Sydney, Australia in 995, he has been developing signal processors and software radios in various fors for several application spaces. He is responsible in developing specialist expertise in leading edge digital techniques, signal processing and FPGA technologies and aintaining flexible and configurable building blocks that can be rapidly adapted to provide quick engineering solutions. Mr. Pathikulangara is a joint recipient of CSIRO Chairan s Medal in 7 for exceptional research in gigabit wireless counications. List of figures Fig.. a) Block diagra of the test setup; b) Photograph of the RF odule assebly where: is the antenna array; is the LO input; 3-6 are outputs. Fig.. Block diagra of the receiver configured for DBF experients. Fig. 3. Photograph of the digitizer odule assebly where and are two identical digitizers, 3 is interconnecting cable and 4 is a sapling oscillator. Fig. 4. Syste test setup in the far field anechoic chaber where is the receive array RF odule, is the rotator and 3 is the transit antenna aperture, 4 is the receive antenna array asked with absorbers 5. Fig. 5. Frequency-doain digital bea forer structure. Fig. 6. Transitted signal forat. Fig. 7. Transitted digital signal. Fig. 8. Captured 4 channel received signals at aziuth. Fig. 9. frequency response for each channel aziuth. Fig.. Phase frequency response for each channel aziuth. Fig.. Coparison of the E-plane co-polar radiation patterns for digitally (DBF) and analogue bea fored (ABF) array [7] at aziuth. Fig.. E-plane co-polar radiation patterns for digitally (DBF) and analogue bea fored (ABF) array [7] for a selection of positive and negative aziuth angles.