IMPULSE radiating antenna s (IRAs) have been used to radiate

Transcription

1 812 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 On the Characterization of a Reflector Impulse Radiating Antenna (IRA): Full-Wave Analysis and Measured Results Majid Manteghi, Member, IEEE, and Yahya Rahmat-Samii, Fellow, IEEE Abstract There is a growing demand for impulse radiating antennas (IRAs) to receive and transmit short pulses. The basic concepts of IRA are reviewed and the far-field pattern versus frequency of an ideal IRA is characterized based on the fundamental properties of IRA. It is shown that the transmitted pulse is ideally in the form of a time derivative of the input pulse. The physical optics simulation results show that the far-field characteristics of a parabolic reflector are very close to an ideal IRA if it is fed properly. The reflector IRA was constructed, analyzed and measured at UCLA. The near-field and far-field characteristics of the reflector IRA are studied using both the method of moments (MoM) full-wave simulations and the frequency domain measurements. In this paper, the radiation mechanism of the reflector IRA is studied using a detailed current distribution on the parabolic reflector and the feeding structure at different frequencies. Applying either the calculated current distribution on the reflector IRA or the measured near-field results, it is seen that the aperture field intensity of the parabolic reflector is not the same in the two principle planes and as a result the beam-widths in the two principle planes are different. The far-field patterns of the antenna are measured and the calculated far-field patterns support the measured results. The calculated current distribution results provide a guideline on how to properly change the feeding structure to achieve a more uniform aperture field and increase the antenna radiation efficiency. Index Terms Antenna measurements, frequency domain analysis, impulse radiating antenna (IRAs), method of moments (MoM), reflector antenna, ultrawide-band antenna. I. INTRODUCTION IMPULSE radiating antenna s (IRAs) have been used to radiate electromagnetic energy in a very short period of time [1], [2]. Various types of antennas are proposed and tested for different applications. The reflector IRA [3] [5] and the transverse electromagnetic (TEM) horn antenna [6], [7] have been used extensively for high power applications. The reflector IRA (Fig. 1) consists of a parabolic reflector fed by a self-reciprocal TEM transmission line [8], [9]. The spherical wave that propagates through the TEM feed is converted to the plane wave by the parabolic reflector. Before studying the reflector IRA properties a review of an ideal IRA will be presented. Manuscript received January 11, 2005; revised August 17, The authors are with the Antenna Research and Measurements Laboratory, Department of Electrical Engineering, University of California at Los Angeles, Los Angeles, CA USA ( majid@ee.ucla.edu; rahmat@ee.ucla.edu; Digital Object Identifier /TAP Fig. 1. The 57 cm diameter reflector IRA mounted in the spherical near-field chamber at UCLA. A magnified drawing of the feeding structure at the focal point is shown in the right corner of this figure. A. Far-Field Pattern of an Ideal IRA When an ideal IRA is illuminated with a short pulse plane wave in a specific direction and polarization, the open circuit voltage at the antenna port has the same signal shape as the incident wave. The electric field at a reference point of the incident plane wave,, is related to the open circuit voltage at the antenna port,, with the effective length of the antenna [10],,as To achieve a with the same wave form as the incident wave,, in (1) the effective length of the antenna has to have frequency independent amplitude and linear phase which represents a time delay in the open circuit voltage. Equation (A-6), in Appendix A, shows that for an antenna with ultrawide-band (UWB) matched input, the antenna gain is directly proportional to the square of the effective length of the antenna and square of frequency. Because the effective length of an ideal IRA has to be independent of frequency, gain would be proportional to (1) X/$ IEEE

2 MANTEGHI AND RAHMAT-SAMII: ON THE CHARACTERIZATION OF A REFLECTOR IRA 813 Fig. 2. A normalized far-field pattern of an ideal IRA when the H-plane beam-width and the E-plane beam-width are the same. 3 db beam-width is shown in dashed lines. the square of frequency, therefore, in the direction that is independent of frequency (usually boresight) the radiated electric field intensity has to be proportional to frequency The solid angle of the antenna is defined as [11] where is the maximum directivity of the antenna. For simplicity, assume that the antenna efficiency is 100%, so the antenna directivity is equal to the antenna gain. The IRA gain is proportional to the square of frequency, thus the solid angle is proportional to the inverse of the square of frequency. As a result, in an ideal case, the beam-width (BW) of the antenna can be approximately related to frequency as (2) Fourier transformation of (5) confirms the linear relationship between the far-zone field intensity and frequency as in (2). Now we can list the properties of an ideal IRA as: 1) Input reflection coefficient is low in an ultrawide frequency band; 2) Gain is proportional to the square of frequency; 3) The radiated far-field at boresight is proportional to the time derivative of the aperture field; 4) A linear phase relationship exists between the radiated far-field and the input signal in the entire frequency band. (all frequency components have the same time delay); 5) Direction of the main beam of the radiation pattern does not change over the whole frequency band and the beam width is proportional to 1/f. In Section II, first the parabolic reflector is illuminated by a frequency independent spherical wave ideal feed. The scattered fields are calculated using physical optics technique and it is shown that the far-field patterns demonstrate similar behavior as far-field patterns of an ideal IRA. Then, the reflector IRA with its TEM feeding structure is studied. A method of moments based software, Hybrid electric field integral equation (EFIE) and magnetic field integral equation (MFIE) Iterative (HEMI) [12], is employed to calculate the current distribution on the antenna body as well as the far-field patterns. The antenna is measured at the recently constructed spherical near-field measurement chamber at UCLA and the experimental results are presented in Section III. Aperture fields are calculated using the holographic back projection technique and the result are compared with the calculated result for the current distribution from Section II. Furthermore, the far-field patterns calculated using near-field measured data are compared with the ones calculated by full-wave analysis. Conclusion is presented in Section IV. (3) Equation (3) suggests an expected far-field pattern for an ideal IRA. For a typical antenna pattern, the anticipated normalized far-field pattern for an ideal IRA versus frequency is shown in Fig. 2. B. Far-Field in Time Domain For a planar aperture, the far-field at boresight ( or ) is related to the tangential electric field,, using (B-5) in Appendix B, i.e., where and are the aperture area and the speed of light respectively. For an antenna with a uniform aperture field the electric far-field at boresight is Equation (5) shows that the electric field in the far zone at boresight is proportional to the time derivative of the aperture field. (4) (5) II. PARABOLIC REFLECTOR A parabolic reflector antenna with an idealized feeding structure was initially analyzed to compare its performance to that of an ideal IRA. Due to the massive calculations associated with the method of moments to sweep in a wide frequency band, we used the frequency domain physical optics (PO) technique [13] for the idealized reflector IRA. Later, HEMI is employed to simulate the Reflector IRA with the actual feeding structure. A. Idealized Reflector IRA An infinitely small dipole was used to illuminate a 57 cm parabolic reflector with a focal length to diameter ratio of This configuration is simply a parabolic reflector fed by a frequency independent spherical wave illuminator with and radiation patterns in two principle planes. The frequency domain PO technique is used to calculate the scattered far-field of the parabolic reflector for a wide frequency band. The calculated far-field pattern in the E-plane is shown in Fig. 3. The far-field pattern in the H-plane is almost the same as the one in the E-plane. Comparing Fig. 3 with the expected far-field pattern for an ideal IRA (Fig. 2), one realizes that the far-field pattern of the parabolic reflector is very

3 814 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 Fig. 3. Calculated scattered E at E-plane (Co-pol.) for the reflector with a short dipole feed. The fields are normalized to their own maximum for each frequency. close to the far-field pattern of an ideal IRA when it is properly illuminated. B. Practical Reflector IRA A nondispersive TEM feeding structure is employed to illuminate a parabolic reflector to realize a reflector IRA. The parabolic reflector converts the spherical TEM mode of the feeding structure to a uniform phase plane wave at the aperture of the reflector. The conical coplanar plates produce a self-reciprocal structure with its circle of symmetry usually lying on the rim of the reflector [8]. It is shown theoretically that half of the power carried through the TEM mode by this structure propagates outside of the circle of symmetry and does not contribute to the highly directive radiated field by the parabolic reflector [1], [8]. In order to demonstrate the performance of the reflector IRA experimentally, an antenna was built at UCLA using a 57 cm parabolic reflector with a focal length to diameter ratio of 0.40 and two pairs of conical coplanar plates, which are placed perpendicular to each other, as the TEM feeding structure (Fig. 1). The size and angles for these plates are chosen in a way to reach characteristic impedance of 400 for each pair of conical transmission lines and all arms are terminated to the reflector with their own characteristic impedances loads [14], [15]. There is an analytical method introduced in [3] to analyze the reflector IRA which gives the radiation fields and the near-zone fields with a good approximation. In this simple model, the interaction between the TEM feeding structure and the parabolic reflector is not taken into account; therefore, the calculated radiation field does not contain the correct form of the tail waveform [16]. A method of moments based software, Hybrid EFIE and MFIE Iterative (HEMI), is employed to calculate the antenna surface currents and the far-field characteristics. Because the reflector and the feeding structure are included in the HEMI model (Fig. 4), the results include the blockage effects of the TEM feeding structure and the interaction between the reflector and the TEM feeding structure. Furthermore, all current distributions are calculated including the total, co-polarization, and cross-polarization currents. The two perpendicular conical coplanar plates for are connected to each other as well as the two in and the gap source Fig. 4. HEMI mesh model of the reflector IRA. The maximum edge length for the feeding arm is 4 mm and for the parabolic reflector is 10 mm. This only shows one quarter of the structure in Fig. 1 due to the symmetry. is placed between these two so we expect a -polarized radiation field. The reflector and its feeding structure have even and odd symmetry with respect to the plane and the plane, respectively. Therefore, one quarter of the antenna is used in the simulation. A PEC plane and a PMC plane are placed at and planes, respectively. This reduces the number of unknowns by a factor of four. HEMI uses the RWG [17] basis function. For the present application the HEMI mesh model of the reflector IRA, Fig. 4, the maximum edge length for the feeding arm is 4 mm and for the parabolic reflector is 10 mm. Fig. 5 shows the calculated current distribution on the reflector at three different frequencies. These figures show that the current distribution is not uniform and it is more concentrated around the plane (H-plane). Different current distributions along and axis generates different beam widths in the far zone for the two principle planes which reduces the aperture efficiency. The current distributions on the TEM feeding structure are calculated as well (Fig. 6). These figures demonstrate that the current density on the feeding arms decreases with distance from the focal point. Also, a standing wave effect at the end of the feeding arms can be seen. In the HEMI model, a wide variety of resistive load are tried between the feeding arms and the reflector to avoid the standing waves but they did not disappear. It means that there is some reactive energy stored in the junction area and a single resistive load cannot match the feeding arms to the surface of the reflector. One may need to use a combination of lumped impedances for termination, as mentioned in [18]. Furthermore, the calculated currents show that the current density is higher at the edges of the feeding arms and has lower density along the middle of each arm. The points with lowest current density on the feeding arms are the optimal places for the coaxial cable to detach from the arms. Thus, we can use each one of these arms as an UWB balun [19]. III. EXPERIMENTAL RESULTS To measure the time domain characteristics of an IRA, one can use either short pulses and a time-domain measurement

4 MANTEGHI AND RAHMAT-SAMII: ON THE CHARACTERIZATION OF A REFLECTOR IRA 815 Fig. 5. Calculated y-component (co-pol), and x-component (cross-po.) of the current distribution on the reflector at (a) 1 GHz, (b) 4 GHz, and (c) 6 GHz. setup or many frequencies in a wide frequency band and use an inverse Fourier transformation to calculate the time-domain results. In this work, we used the frequency domain measurement method. The recently constructed spherical near-field measurement chamber and the far-field anechoic chamber at UCLA are used to measure the radiation characteristics of the antenna. The measured results presented in the following sections are: A) input impedance, B) holographic images, and C) far-field patterns. A. Input Impedance An IRA, at first has to have a low reflection coefficient in a wide frequency band. The conical coplanar TEM transmission line is designed to achieve a 200 input impedance at the antenna port. Fig. 7(a) shows the real part and the imaginary part of the measured input impedance which is observed at the focal point. The associated scattering parameter in a 200 system is presented in Fig. 7(b). Fig. 7(a) shows that the input impedance of the reflector IRA has a low variation about 200 in the frequency band between 1.5 to 13 GHz. An HP-8510B vector network analyzer was used for all measurements and the sweep frequency start from 45 MHz to 13 GHz with 801 points. One can use the inverse Fourier transformation of the reflected Fig. 6. Calculated current distribution on the TEM feeding structure at different frequencies. (a) 1 GHz, (b) 2 GHz, (c) 4 GHz, (d) 6 GHz. signal from the antenna port to study some important properties of the TEM feeding structure [16]. Fig. 8(a) shows a schematic drawing of a quarter of the feeding structure with the parabolic reflector. Some of the important points in the reflected signal are indicated with the capital letters as A) the feeding point, B) reflection point at the reflector apex, C) the chip resistor point, and D) the connection point between the feeding arm and the reflector. The inverse Fourier transformation of the measured reflected signal for different terminations are presented in Fig. 8(b). In addition to short-circuit and open-circuit, 100 and 200 are examined to study the effect of the termination on the reflected signal tail. Because the inverse Fourier transformation is applied to the signals with low frequency components, the truncation error generates some variation between the low frequency behaviors of the reflected signal tails versus termination loads. This effect appears as a local DC shift in the signal waveform. There is a differentiated impulse associated with the feeding point at which is well-matched to the capacitive properties of the input impedance shown in Fig. 7(a). Since the input signal has not seen the termination load at this time, there is almost no difference between reflected signals for different terminations. To look at the reflected signal in more details, Fig. 8(b) is magnified with horizontal axis in time scale and distance scale in Fig. 8(c) and 8(d) respectively (time is measured for a round trip). The next important point in the reflected signal waveform

5 816 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 Fig. 7. (a) Measured real part and imaginary part of the input impedance of the reflector IRA at the focal point. (b) scattering parameter (s ) in a 200 system. Fig. 8. (a) Schematic drawing of a quarter of the reflector IRA with some of the important points in the reflected signal. (b) The time domain reflected signal for different chip-resistors. (c) A magnified part of the b which shows the reflected signal from different points for different loads. (d) The same figure as c but with the horizontal axis scaled in distance. indicates the first reflection from the apex of the parabolic reflector which is located at ns or. With the exception of the low frequency difference, the reflected signal waveform from point does not have an observable variation versus termination loads. Point, located at ns or, is associated with the termination loads. Different termination loads have different signature in the reflected signals from point. The short-circuit and open-circuit loads determine the upper and lower limits for the reflected signal from this point. The feeding arm is connected to the parabolic reflector at point located at ns or. The other important feature which is able to be identified is associated with the second reflection from point and is located at ns or. There is no feature in the schematic drawing that can explain the small spike at point ( ns or ). There is a small spike at the point which was first identified as an error in the inverse Fourier transformation. Since this signature can be seen in the entire measured data, there should be a physical explanation. There is a small patch on the surface of the parabolic reflector which is located exactly at m away from the focal point which seems to be associated with this small spike. The time domain reflected signals for different termination loads show that neither the short-circuit nor the open-circuit are the best termination loads. Furthermore, there is no significant difference observed between 100 and 200 loads. It means

6 MANTEGHI AND RAHMAT-SAMII: ON THE CHARACTERIZATION OF A REFLECTOR IRA 817 Fig. 9. Holographic images from measured at (a) 1 GHz, (b) 4 GHz, and (c) 6 GHz for amplitude and phase (unwrapped) of the co-polarization aperture fields. that there is some reactive energy stored in the arm-reflector connection area which can not be reduced using just the resistive loads. One has to use a combination of resistive and reactive loads or change the arm-reflector connection to make a better match. B. Holographic Images Using the measured spherical near-field data, the holographic images can be generated for different tangential field at the antenna aperture [20]. Fig. 9 shows the co-polarized tangential magnetic field (electric current) on a plane located at the reflector aperture. These figures indicate that the measured field intensity is not uniform on the antenna aperture. The measured field has higher intensity around the axis in comparison to the axis area. Also, the co-polarized components have either a reasonably constant phase or linear variation phase in those areas with higher field intensity. The linear phase variation observed in some of the holographic images is because of a slight misalignment of the reflector IRA with the measurement system. If the phase-center of the antenna is shifted in plane from the center phase of the measurement system a linear variation will appear in the measured phase in the aperture plane. These results have a good agreement with the calculated current distribution on the antenna surface using HEMI (Fig. 5). C. Far-Field Once the tangential near-field distribution over an antenna enclosure or the electric current distribution on the surface of the antenna is known, one can calculated the far-field patterns. The calculated far-field from the near-field measured data in azimuth-elevation plane and the calculated far-field from both the measured data and the HEMI results for two principle planes are presented in Fig. 10.

7 818 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 Fig. 10. H for the E-plane) of measured far-field patterns with the calculated ones at (a) 1 GHz, (b) 4 GHz, and (c) 6 GHz. Normalized elevation-azimuth measured far-field patterns and a comparison between two principle planes cuts (E and H for the H-plane and E and Both the calculated current distributions illustrated in Fig. 5 and the holographic images (Fig. 9) show that the aperture field is more concentrated around the -axis. So, the beam-width in the H-plane is expected to be narrower than the beam-width in the E-plane. The feeding structure is symmetric in the plane and the symmetry in the plane depends on performance of the balun. As seen from these figures, the measured far-field patterns at all frequencies are fairly symmetric in the both principle planes and this confirms the balanced feeding mechanism. The H-plane and E-plane cuts calculated from near-field measured data and full-wave analysis (HEMI) for different frequencies are also presented in Fig. 10. The calculated far-field from full-wave simulation has a good agreement with the one calculated from the near-field measured data. In Appendix A it is shown that the gain of an ideal IRA has to increase with square of frequency (A-6). The maximum gain of a parabolic reflector is obtained at boresight,so one has to measure and calculate the far-field of the reflector IRA at versus frequency. We used the three antenna calibration system to measure the antenna gain versus frequency. The measurement setup consists of two ultra-wideband horn antennas, the reflector IRA, and a HP8510B vector network analyzer. The far-field anechoic chamber of UCLA was used for the far-field measurements. Fig. 11 shows the measured and calculated far-zone frequency response of the antenna at boresight. For the convenience of this study, a ramp function (representing a delay in the time domain) has been subtracted from the phase of the calculated and measured far-field [Fig. 11(b)]. As one can see from this figure, the phase has a low variation in the entire

8 MANTEGHI AND RAHMAT-SAMII: ON THE CHARACTERIZATION OF A REFLECTOR IRA 819 Fig. 11. Calculated and measured far-zone frequency response of the reflector IRA. (a) Calculated and measured amplitude, (b) calculated and measured phase. Fig. 12. (a) Input differentiated Gaussian pulse. (b) Calculated and measured boresight radiation far-field. frequency band of operation. Low phase distortion is one of the requirements of the IRAs. The measured far-zone frequency response of the antenna deviates from the calculated one in some frequency windows. Some of the reasons can be listed as follows: 1) The frequency cutoff of the UWB standard horn antennas is 1.2 GHz so the low frequency results are not valid. 2) The far-field anechoic chamber is designed for C-band and X-band and it may affect the measured data at lower frequencies. 3) There are some mechanical errors due to the hand-made feeding structure. Also, the reflector surface has some irregularities that contribute to surface errors. In order to calculate the time-domain radiated far-field at boresight a differentiated Gaussian pulse is used as an example of the input pulse [Fig. 12(a)] and convolved with the inverse Fourier transformation of the calculated and measured frequency response of the antenna [Fig. 12(b)]. Clearly other forms of the input pulses can also be used [18]. The time domain radiated waveform has a small pulse similar to the input signal with a minus sign at [point in the Fig. 12(b)]. This early time signal is associated with the radiation from the TEM feed itself. It has been shown that half of the power accepted by the reflector IRA radiates directly through the TEM feeding structure [1], [2]. The current distribution on the reflector IRA and its feeding structure is calculated using HEMI at 4 GHz. Fig. 13. Far-field pattern of the feeding structure at 4 GHz. It is assumed that the feeding structure by itself carries the same current when it is connected to the parabolic reflector (Fig. 6 at 4 GHz). Then the radiation patterns of the feeding structure itself while it is carrying the same current as the complete antenna are calculated. Fig. 13 shows a 3-D radiation pattern of the TEM feed at 4 GHz without the parabolic reflector. This figure shows that the far-field pattern of the TEM feed has a wide beam-width.

9 820 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 The far-field pattern of the reflector IRA is the superposition of the visible part of this pattern and the directed far-field pattern of the parabolic reflector. Therefore, the early time radiated signal does not change rapidly with variation. It takes almost 1.53 ns [point in the Fig. 12(b)] for the excitation waveform to go through a round trip to the reflector and come back to the focal point. As discussed in the introduction, the radiated field from a good IRA aperture has to have a time derivative relationship with the excitation waveform. Fig. 12(b) shows that the far zone time domain signal calculated from both the measured results and the full-wave analysis (HEMI) at point is the time derivative of the input differentiated Gaussian pulse and it agrees well with the theoretical results. Fig. 14. Receiving antenna is connected to a transmission line with characteristic impedance Z which is terminated to its match load. time signal and the time derivative of the input signal can be identified in both results. APPENDIX IV. CONCLUSION Based on the definition of the gain and effective length of the antenna in the frequency domain, the far-field pattern of an ideal IRA were determined. It was shown that the far zone field strength and the inverse of the beam-width are proportional to frequency for an ideal IRA. So, the time domain signal has to have a time derivative relationship with the excitation waveform. Also it was shown that a parabolic reflector can function very close to an ideal IRA if it is fed properly. A MoM based software (HEMI) was employed to calculate the current distribution and the far-field characteristics of the reflector IRA. The reflector IRA was constructed and measured in the spherical near-field chamber and the far-field anechoic chamber at UCLA. By taking the inverse Fourier transform of the measured scattering parameter, the time domain reflected signal at the antenna port was calculated. It was shown that a resistive termination load cannot completely match the feeding arm to the parabolic reflector. To avoid the standing wave at the feeding arms one has to either change the end of the feeding arms in the way to reduce the reactance of the arm-reflector junction or use a combination of resistive and reactive termination loads. The full-wave analysis (HEMI) results at different frequencies shows that the current density around the axis is stronger than the current density around -axis. The holographic images were calculated from the near-field measured data confirm these results too. For that reason the H-plane has to have a narrower beam-width in comparison to the E-plane. The farfield patterns in the azimuth (H-plane) and elevation (E-plane) were calculated from the near-field measured data and presented for different frequencies. These figures have a good agreement with the calculated and measured near-field data. To obtain a symmetric far-field pattern (equal beam width at E-plane and H-plane) one has to change the feeding structure in the way to illuminate the parabolic reflector uniformly. The calculated and measured far zone frequency responses of the antenna at boresight were used to calculate the radiated field associated with a differentiated Gaussian pulse. The early A. Ideal Aperture in the Frequency Domain If the receiving antenna with input impedance is connected to a transmission line with characteristic impedance and is terminated to its own matching load (Fig. 14), the maximum delivered power by the receiving antenna can be calculated from the open circuit voltage,, and effective length,, as (A-1) The maximum delivered power by the receiving antenna also is related to the effective aperture by where is the free space impedance, and gain is defined as (A-2) (A-3) One can calculate the relation between the effective aperture versus effective length of the antenna using (A-1) and (A-2) (A-4) Finally, substituting from (A-4) into (A-3), gain of the antenna is calculated as a function of the effective length by which can be simplified to (A-5) (A-6)

10 MANTEGHI AND RAHMAT-SAMII: ON THE CHARACTERIZATION OF A REFLECTOR IRA 821 for an antenna with an input impedance matched to the characteristic impedance of the transmission line at entire frequency range where is the speed of light in free space. B. Ideal Aperture in the Time Domain The frequency domain radiated field of a set of arbitrary sources, and, which are limited in a closed volume in free space is given [10] by (B-1) where the observation point is located at, the source point at,, and. One can rewrite this formula for far-field by substituting by in (B-1) (B-2) The time domain electric field in the far zone and in its general form [21] can be written from (B-2) as where the retarded time,,is defined as (B-3) (B-4) For simplicity, assume the aperture as a planar surface which is located at. If a PEC is located right behind the aperture due to the image theory ( and where is the tangential electric field in the aperture) we have where is the area of the aperture. (B-5) REFERENCES [1] R. H. DuHamel et al., Frequency independent conical feeds for lens and reflectors, in Proc. IEEE Int. Antennas Propagation. Symp. Dig., vol. 6, Sep. 1968, pp [2] C. E. Baum, Radiation of impulse-like transient fields, Sensor and Simulation Notes #321, Nov [3] C. E. Baum and E. G. Farr, Impulse radiating antennas, in Ultra-Wideband, Short-Pulse Electromagnetics, H. L. Bertoni, L. Carin, and L. B. Felson, Eds. New York: Plenum, 1993, pp [4] E. G. Farr, C. E. Baum, and C. J. Buchenauer, Impulse radiating antennas, Part II, in Ultra Wideband/Short-Pulse Electromagnetics 2, L. Carin and L. B. Felsen, Eds. New York: Plenum Press, 1995, pp [5] E. G. Farr and C. E. Baum, Impulse radiating antennas, part III, in Ultra Wideband/Short-Pulse Electromagnetics 3, C. E. Baum, L. Carin, and A. P. Stone, Eds. New York: Plenum Press, 1997, pp [6] M. Kanda, Transients in a resistively loaded linear antenna compared with those in a conical antenna and a TEM horn, IEEE Trans. Antennas Propag., vol. 28, no. 1, pp , Jan [7] A. P. Lambert, S. M. Bookers, and P. D. Smith, Calculation of the characteristic impedance of TEM horn antennas using the conformal mapping approach, IEEE Trans. Antennas Propag., vol. 43, no. 1, pp , Jan [8] C. E. Baum, Radiation from self reciprocal aperture, Sensor and Simulation Notes #357, Apr [9] E. G. Farr and C. E. Baum, Radiation from self-reciprocal aperture, in Electromagnetic Symmetry, C. E. Baum and H. N. Kiritikos, Eds. Bristol, U.K.: Taylor and Francis, 1995, ch. 5. [10] Antenna Handbook, Y. T. Lo and S. W. Lee, Eds., Van Nostrand Reinford, New York, [11] W. L. Stutzman and G. A. Thiele, Antenna Theory and Design. New York: Wiley, [12] R. E. Hodges and Y. Rahmat-Samii, An iterative current-based hybrid method for complex structures, IEEE Trans. Antennas Propag., vol. 45, no. 2, pp , Feb [13] Y. Rahmat-Samii, Reflector antennas, in Antenna Handbook, Y. T. Lo and S. W. Lee, Eds. New York: Van Nostrand Reinford, [14] J. S. Tyo, Optimization of the TEM feed structure for four-arm reflector impulse radiating antennas, IEEE Trans. Antennas Propag., vol. 49, no. 4, pp , Apr [15] E. G. Farr and C. E. Baum, Prepulse associated with TEM feed of an impulse radiating antenna, Sensor and Simulation Notes #337, Mar [16] K. Kim and W. R. Scott, Numerical analysis of the impulse radiating antenna, Sensor and Simulation Notes #474, Jun [17] S. Rao, D. Wilton, and A. Glisson, Electromagnetic scattering by surfaces of arbitrary shape, IEEE Trans. Antennas Propag., vol. 30, no. 3, pp , May [18] D. V. Giri and C. E. Baum, Temporal and spectral radiation on boresight of a reflector type of Impulse Radiating Antenna (IRA), in Ultra Wideband/Short-Pulse Electromagnetics 3, C. E. Baum, L. Carin, and A. P. Stone, Eds. New York: Plenum Press, 1997, pp [19] M. Manteghi and Y. Rahmat-Samii, A novel UWB feeding mechanism for the TEM horn antenna, reflector IRA, and the Vivaldi antenna, IEEE Antennas Propag. Mag., vol. 46, no. 5, pp , Oct [20] Y. Rahmat-Samii and J. Lemanczyk, Application of spherical near-field measurements to microwave holographic diagnosis of antennas, IEEE Trans. Antennas Propag., vol. 36, no. 6, pp , Jun [21] A. Taflove, Ed., Advances in Computational Electrodynamics the Finite- Difference Time-Domain Method. New York: Artech House, 1998, ch. 7, p Majid Manteghi (S 01 M 05) received B.S. and M.S. degrees from the University of Tehran, Tehran, Iran, in 1994 and 1997, respectively, and the Ph.D. degree in electrical engineering from the University of California, Los Angeles (UCLA), in He worked as a Research Assistant in the Microwave Laboratory, University of Tehran, from 1994 to 1997, where he designed microstrip patch antennas, arrays, traveling wave antennas, handset antennas, Base Transceiver Station (BTS) single and dual polarized antennas, reflector antennas, and UHF transceiver circuits and systems. From 1997 to 2000, he worked in the telecommunication industry in Tehran where he served as the head of an RF group for a GSM BTS project. In fall 2000, he joined to the Antenna Research, Analysis, and Measurement Laboratory (ARAM) of the University of California, Los Angeles. He is currently a Research Engineer with the Electrical Engineering Department of UCLA. His research area has included ultrawide-band impulse radiating antennas, miniaturized patch antennas, multiport antennas, dual frequency dual polarized stacked patch array designs, and miniaturized multiband antenna for MIMO applications.

11 822 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 3, MARCH 2006 Yahya Rahmat-Samii (S 73 M 75 SM 79 F 85) received the M.S. and Ph.D. degrees in electrical engineering from the University of Illinois, Urbana-Champaign. He was a Guest Professor with the Technical University of Denmark (TUD) during summer He was a Senior Research Scientist at NASA s Jet Propulsion Laboratory, California Institute of Technology, Pasadena, before joining the University of California, Los Angeles (UCLA) in Currently, he is a Distinguished Professor and the Chairman of the Electrical Engineering Department, UCLA. He has also been a Consultant to many aerospace companies. He has been Editor and Guest Editor of many technical journals and book publication entities. He has authored and coauthored more than 660 technical journal articles and conference papers and has written 20 book chapters. He is the coauthor of Impedance Boundary Conditions in Electromagnetics (Washington, DC: Taylor & Francis, 1995) and Electromagnetic Optimization by Genetic Algorithms (New York: Wiley, 1999). He is also the holder of several patents. He has had pioneering research contributions in diverse areas of electromagnetics, antennas, measurement and diagnostics techniques, numerical and asymptotic methods, satellite and personal communications, human/antenna interactions, frequency selective surfaces, electromagnetic bandgap structures and the applications of the genetic algorithms. On several occasions, his work has made the cover of many magazines and has been featured on several television newscasts. Dr. Rahmat-Samii is a Member of Sigma Xi, Eta Kappa Nu, Commissions A, B, J, and K of the United States National Committee for the International Union for Radio Science (USNC/URSI), Antennas Measurement Techniques Association (AMTA), and the Electromagnetics Academy. He was elected as a Fellow of the Institute of Advances in Engineering (IAE) in Since 1987, he has been designated every three years as one of the Academy of Science s Research Council Representatives to the URSI General Assemblies held in various parts of the world. In 2001, he was elected as the Foreign Member of the Royal Academy of Belgium for Science and the Arts. He was also a member of UCLA s Graduate council for a period of three years. For his contributions, he has received numerous NASA and JPL Certificates of Recognition. In 1984, he received the coveted Henry Booker Award of the URSI which is given triennially to the Most Outstanding Young Radio Scientist in North America. In 1992 and 1995, he was the recipient of the Best Application Paper Prize Award (Wheeler Award) for papers published in the 1991 and 1994 IEEE ANTENNAS AND PROPAGATION. In 1999, he was the recipient of the University of Illinois ECE Distinguished Alumni Award. In 2000, he was the recipient of IEEE Third Millennium Medal and AMTA Distinguished Achievement Award. In 2001, he was the recipient of the Honorary Doctorate in physics from the University of Santiago de Compostela, Spain. In 2002, he received the Technical Excellence Award from JPL. He is the winner of the 2005 URSI Booker Gold Medal to be presented at the URSI General Assembly. He was also a Member of the Strategic Planning and Review Committee (SPARC) of the IEEE. He was the IEEE AP-S Los Angeles Chapter Chairman ( ) and his chapter won the Best Chapter Awards in two consecutive years. He was the elected 1995 President and 1994 Vice-President of the IEEE Antennas and Propagation Society. He was one of the Directors and Vice President of the Antennas Measurement Techniques Association (AMTA) for three years. He was appointed an IEEE Antennas and Propagation Society Distinguished Lecturer and presented lectures internationally. He is listed in Who s Who in America, Who s Who in Frontiers of Science and Technology, and Who s Who in Engineering. He is the designer of the IEEE Antennas and Propagation Society logo that is displayed on all IEEE ANTENNAS AND PROPAGATION publications.