WITH a widespread adaptation of radio frequency identification

Transcription

1 2620 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012 Dual-Band Long-Range Passive RFID Tag Antenna Using an AMC Ground Plane Dongho Kim, Member, IEEE, and Junho Yeo, Member, IEEE Abstract A dual-band passive radio frequency identification (RFID) tag antenna applicable for a recessed cavity in metallic objects such as heavy equipment, vehicles, aircraft, and containers with long read range is proposed by using an artificial magnetic conductor (AMC) ground plane. The proposed tag antenna consists of a bowtie antenna and a recessed cavity with the AMC ground plane installed on the bottom side of the cavity. The AMC ground plane is utilized to provide dual-band operation at European ( MHz) and Korean ( MHz) passive UHF RFID bands by replacing the bottom side of the metallic cavity of a PEC-like behavior and, therefore, changing the reflection phase of the ground plane. It is worthwhile to mention that the European and the Korean UHF RFID bands are allocated very closely, and the frequency separation ratio between the two bands is just about 0.045, which is very small. It is demonstrated by experiment that the maximum reading distance of the proposed tag antenna with optimized dimensions can be improved more than 3.1 times at the two RFID bands compared to a commercial RFID tag. Index Terms Antenna, artificial magnetic conductor (AMC), cavity, dual-band, radio frequency identification (RFID), tag. I. INTRODUCTION WITH a widespread adaptation of radio frequency identification (RFID) systems, possibilities of RFID tags being attached on various types of metallic objects have been increased and many researchers have extensively investigated for RFID tag antenna design for metallic objects in recent years. Although a simple design method for a low-cost tag antenna using a double-folded dipole antenna with T-matching and a foam spacer has been used in the past [1], most designs are based on a microstrip-patch-type antenna or a planar inverted-f-type antenna that has its own ground plane [2] [5]. Later, a somewhat different approach using a dipole antenna and an artificial magnetic conductor (AMC)-type ground plane has been studied by the authors for low-profile and platform-tolerant RFID tags [6], [7]. For multiband operation, dual-band planar inverted-f antenna (PIFA)-type RFID tag antennas mountable on metallic surfaces operating in the two frequency bands over MHz passive UHF bands have Manuscript received November 17, 2010; revised October 21, 2011; accepted November 26, Date of publication April 12, 2012; date of current version May 29, This work was supported by the Daegu University Research Grant. D. Kim is with Department of Electronic Engineering, Sejong University, Seoul , Korea ( dongkim@sejong.ac.kr). J. Yeo is with School of Computer and Communication Engineering, Daegu University, Gyeongbuk , Korea ( jyeo@daegu.ac.kr). Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TAP also been proposed [8], [9]. However, the read range of all the tag designs for metallic objects is limited to a few meters, which is similar to that of a common dipole-like label tag. In metallic objects such as vehicles, aircraft, ships, heavy equipment, and metallic containers, there exist various types of recessed volumes, and these recessed volumes can be used as cavities to form a cavity-backed antenna with a common label-type RFID tag. This type of RFID tag antenna can increase the read range considerably. By using this concept, an embedded circular patch-type RFID tag antenna in UHF band using ceramic material for identifying metallic objects has been proposed, and the effect of various positions of the embedded tag in a cavity of a metallic object on the maximum reading distance has been analyzed [10]. However, the read range has not been considerably enhanced in this case. Recently, both a longrange passive RFID tag antenna consisting of a bowtie-type antenna in a recessed rectangular cavity in metallic objects and an impedance matching technique for this configuration have been proposed by the authors [11]. Instead of modifying antenna geometry itself, the coupling between the bowtie antenna and the cavity has been used to match the antenna s input impedance to the tag s chip impedance. The maximum read range has been improved about 3.2 times compared to a commercial label-type RFIDtaginsingleband. In this paper, we propose a dual-band passive RFID tag antenna applicable for a recessed cavity in metallic objects with long read range by using an AMC ground plane. The proposed tag antenna is comprises a bowtie antenna and a recessed cavity with the AMC ground plane installed on the bottom side of the cavity. The AMC ground plane is utilized to provide dualband operation at European ( MHz) and Korean ( MHz) passive UHF RFID bands, which are allocated very closely, and the frequency separation ratio between the two bands is about 4.5%. By simply replacing the bottom side of the metallic cavity with the AMC ground plane, we can change the reflection phase of the ground plane, and therefore, we can obtain double resonance at the desired two frequency bands. The maximum reading distance of the proposed tag antenna with optimized dimensions is compared to that of a commercial ALN-9540-WR RFID tag from Alien Technology Co. [12], which is flexible and cheap and designed to provide performance in a general standalone environment. All simulation data are obtained using CST Microwave Studio [13]. II. ANTENNA DESIGN AND RESULTS A. Antenna Design With an AMC Ground Plane The geometries of the proposed RFID tag antenna and the AMC ground plane are shown in Figs. 1 and 2, which have X/$ IEEE

2 KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE 2621 Fig. 1. Geometry of the proposed dual-band bowtie-type RFID tag antenna embedded in a recessed volume of a metallic cavity. Fig. 3. Photograph of (a) the fabricated cavity with the AMC ground plane and (b) bowtie tag antenna with the attached Higgs-2 chip. Fig. 2. Geometry of the AMC substrate with mm, mm, and. The radius of the via is 0.5 mm. optimized design parameters of mm, mm, mm, mm, mm, mm, mm, mm, mm, mm, mm, mm,, mm, mm, mm, and mm. Basically, our tag is a dipole-type antenna. Each arm of the tag is composed of a modified bowtie-shaped loop. An RFID chip is attached in between the bowtie loops. The tag is etched only on the bottom side of a Taconic TRF-45 dielectric laminate. The thickness and the relative permittivity of the substrate is 1.63 mm and 4.5, respectively. In this paper, to show the easiness and effectiveness of impedance matching by using a coupling effect between the tag and the metallic cavity, we intentionally chose the bowtie-shaped tag antenna, which is very simple and provides little variety in its design parameters for a fine impedance tuning. All directions of the rectangular metallic cavity are enclosed with metallic walls except the top opening (see Fig. 1). The cavity measures,,and, respectively. To easily attach our tag on various platform structures, we intentionally included the metallic wing in our tag antenna design, which entirely encloses the top opening of the air-filled recessed volume of the cavity. This wing can alleviate the tag s input impedance variation from the possible change of a variety of shapes and material properties of candidate platform structures. The bowtie dipole antenna is placed inside the rectangular metallic cavity with a spacing of from the open top aperture of the cavity. As shown in Fig. 1, to prevent possible damage caused by any physical impact or weathering, we deliberately have placed the antenna and the chip facing the bottom of the cavity. For the tag chip, we have selected a strap-type Higgs-2 chip from Alien Technology Co., which is much easier to directly attach on the bowtie dipole antenna than a bare-type chip without additional strap. The input impedance of the chip is about 11-j130 at 910 MHz. The top view of a compact AMC ground plane in the cavity consisting of an array of an -directed narrow rectangular patch pair with offset vias is depicted in Fig. 2. The total number of AMC unit cells used for the proposed tag is 23. Each narrow rectangular copper patch is etched on 1.63-mm-thick Taconic TRF-45 dielectric laminate. The offset via posts are connecting the rectangular AMC patches to the metallic bottom side of the cavity, and by using these posts, a high-impedance frequency band of the AMC can be considerably lowered maintaining the same length of unit cell as indicated in [6]. Fig. 3 presents the fabricated cavity with the AMC ground plane and the bowtie antenna. To examine the effect of a tag s depth from the top opening of the cavity on the variations of the input reflection coefficient ( ) and the induced power of the tag chip, we have changed from6to30mm,whichisshowninfig.4. To compute the input reflection coefficient, we have replaced the RFID chip with a discrete port and two identical capacitors. Two capacitors of 2.68 pf are connected at the two ends of the

3 2622 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012 TABLE I PERFORMANCE COMPARISON BETWEEN THE ALIEN S ALN-9540 SQUIGGLE TAG AND THE PROPOSED BOWTIE TAG EMBEDDED IN THE CAVITY Maximum realized gain Maximum induced power Maximum increased reading distance ALN-9540 shows an omnidirectional beam pattern in an H-plane. the antenna are most well matched to complex conjugate values of the chip impedance at both frequency bands as shown in Fig. 4(a) and (b), and these optimized values are used for the final design and fabrication. The realized gain of the proposed antenna is 6.74 dbi at 869 MHz and 6.46 dbi at 913 MHz, respectively, and these are more than 4.36 db higher compared to that of a commercial tag ALN-9540 from Alien Technology Co. [12]. Maximally induced power on the chip is 1.35 dbm at 869 MHz and 1.05 dbm at 913 MHz, respectively, which is more than 4.72 db larger than that of ALN-9540 (seetablei). For the comparison of RFID tags recognition performance, the maximum reading distance is one important parameter. Thus, we have compared the reading distance between the proposed tag and the commercial ALN-9540 tag, which can be computed by (1) Fig. 4. Effect of the distance on an antenna performance (a) and (b) power induced in the RFID chip of the tag embedded inside the cavity ( -polarized 1-V/m plane-wave incidence is assumed). port of 11, which gives impedance of about 11-j130 at 910 MHz. To calculate the induced power on the chip, we need a reader antenna. Of course, we can model the reader antenna and include it in our computer simulation together with the proposed tag antenna. However, this kind of approach demands extremely large computation cost not only for computer memories, but for a simulation time. Hence, instead of directly model the reader antenna, we have supposed that an -polarized plane wave is an incoming signal toward the tag from the reader antenna. We assume that the plane wave has an electric field strength of 1 V/m. It is observed that two different resonant frequency bands can be produced by placing the bowtie antenna inside the recessed cavity with the AMC ground plane, and the locations of the two frequency bands and the matching characteristic of the antenna can be adjusted by changing the spacing of the antenna inside the cavity. In fact, as the spacing decreases, the first resonant frequency moves toward higher frequencies, while the second one shifts toward lower frequencies, and the antenna s matching characteristic improves. At mm, the input impedances of where is a difference of maximum reading distances between tag1 and tag2, is the induced power at each chip of RFID tag antennas, and is antenna gain of each tag. The detailed derivation of (1) is given in [11]. Based on the computed gain and induced power, we could determine that our antenna shows much better performance in the reading distance than the commercial ALN-9540 squiggle tag, which is more than at least 2.9 times longer at both frequency bands. The detailed comparison between the proposed antenna and ALN-9540 is summarized in Table I. Now, we show Fig. 5 to explain how dual-band operation can be achieved from a single-band cavity antenna simply by installing the AMC structure. Fig. 5(a) shows the characteristic of the proposed antenna without the AMC ground plane. In this case, we see that single resonant frequency band is produced at around 890 MHz, and the input reflection coefficient of 20 db and the realized gain of 6.9 db are achieved. Note that these values are very similar to those obtained at dual bands when the AMC ground plane is used. Fig. 5(b) shows the total phase behaviors of total fields computed at mm (at the open top of the cavity) and at mm (near the bottom of the cavity) with and without the AMC ground plane. First, when we do not install the AMC ground plane, the total phase at the resonant frequency of 890 MHz is 90. In other words, the phase of about 90 can be considered as a best phase value resulting in the best impedance matching and the highest realized gain characteristics for the proposed tag environment. At this point, it is worth noting that the key idea of impedance matching in

4 KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE 2623 Fig. 5. (a) and realized gain of the proposed RFID tag antenna with and without the AMC ground plane. (b) Total phase responses of total fields with and without the AMC ground plane. Fig. 6. Effects of a via position on (a) reflection phase of the AMC and (b) input reflection coefficients ( ) of the tag antenna. this paper is none other than utilizing coupling effect between the tag and cavity. Therefore, we can come to a conclusion that the phase of 90 is the result of an optimized effect of coupling for the best impedance matching. A similar phase response of an AMC ground plane was also reported in [14]. Next, we have examined the phase value with the AMC ground plane. When the AMC ground plane is installed on the bottom side of the cavity, we can find the total phase of 90 at two frequencies of 869 and 913 MHz, which is different from the resonant frequency without the AMC. The reason for this can be explained by observing the total phase at the top opening of the cavity mm with the AMC ground plane. At 869 MHz, the total phase at mm and mm are almost the same to be about 90. However, the total phase at mm is about at 913 MHz, and this phase value is complemented by the pass difference of 48 mm from the AMC to the cavity open top ( -plane at ). Therefore, the total phase at mm again becomes around 90 at 913 MHz. Consequently, we can say that the first resonance at around 869 MHz comes from the cavity structure itself, while the second resonance at 913 MHz is from the negative reflection phase of the AMC ground plane. To further investigate the characteristics of the compact AMC ground plane, the variation of the reflection phase of the unit cell of the AMC ground plane when the position of the offset via post changes is plotted in Fig. 6(a). For this purpose, the distance between the via post and the center of the unit cell is varied from 40.3 to 42.3 mm. We observe that the zero reflection phase frequency position of the AMC unit cell moves toward lower frequency as the distance increases, which means the offset in the via posts is increased. Fig. 6(b) shows the effect of varying on the input reflection coefficient of the antenna. It is seen that the two resonant frequency bands can be controlled by the offset distance of the AMC ground plane. Fig. 7. Effect of a cavity size on antenna s impedance matching properties. Similar to the behavior of the reflection phase characteristic of the AMC unit cell, the input reflection coefficient of the antenna shifts toward lower frequency as the offset distance increases. It is important to note that any two impedance-matched frequency bands for each value of correspond to the frequencies of total phase of about 90, which is a starting point of the proposed dual-band tag antenna design. Next, we have investigated the effect of a cavity size on an input reflection coefficient of our antenna, which is shown in Fig. 7. In the figure, and denote the length and width of the recessed rectangular cavity (see Fig. 1). From the figure, we can find a resonant frequency variation tendency of the input reflection coefficient caused by changing the dimension of the cavity, which is another important antenna tuning parameter together with the via offset distance showninfig.6(b). There are two ways to move resonant frequency bands toward higher frequencies: One is to increase, and the other is to decrease. That is to say, increasing or decreasing of and results in an opposite effect, which can be explained by analyzing induced surface current distribution on the antenna and

5 2624 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012 Fig. 8. Fig. 1. Equivalent circuit to model the proposed RFID tag antenna shown in the internal walls of the cavity. The detailed explanation for the reason of variation in the input reflection coefficient shown in Fig. 7 can be found in [11]. B. Equivalent Circuit Representation and Experimental Results Fig. 8 shows an equivalent-circuit model of our antenna, which consists of some lumped elements [15], [16]. To model the reactance of the bowtie antenna at a very low-frequency region, a series capacitor is introduced, and simply to negate the effect of at higher frequencies, is connected. A parallel resonance network of,,and creates resonance at frequencies lower than,where denotes a radiation resistance. To describe the parasitic coupling effect between the tag and the cavity, the shunt capacitance and inductance are also inserted [11]. After that, the antenna s input impedance seen from an antenna feed point can be written as Fig. 9. Comparison of antenna s input impedance ( in Fig. 8) computed from 3-D simulation of the antenna structure and from an equivalent circuit shown in Fig. 8. (a) Resistance. (b) Reactance. (2) Fig. 10. Surface current density distribution induced on the AMC substrate at 913 MHz. To complete circuit representation, we computed best values for each lumped element by using a hybrid genetic algorithm (HGA) with a population of 12. The target fitness function that should be minimized is set by where is the total number of frequency points within a target frequency band, and are real and imaginary parts of the input impedance computed by CST MWS, and and are real and imaginary parts of the input impedance obtained from the circuit elements by using the HGA. We compared the simulated input impedance calculated from a three-dimensional field computation with the optimized circuit impedance (see Fig. 8), which is given in Fig. 9. After (3) the HGA optimization, each circuit component value is determined like the following: pf, nh, pf, nh, pf, nh, and k. We can see that the HGA result agrees very well with the simulated antenna s input impedance, which verifies that the proposed equivalent-circuit model and the computed lumped elements values are fairly accurate. Fig. 10 presents the vector surface current distribution induced on the AMC ground plane at 913 MHz. It is clearly observed that most energy is concentrated between the via posts on the AMC, and this makes the offset via lower the zero reflection phase frequency of the AMC. The input impedance variations for some RFID tag antennas are depicted in a Smith chart normalized with 50,whichis given in Fig. 11. In the figure, the solid line with square symbols denoted as corresponds to the proposed tag antenna, and and are for the same antenna without the AMC ground plane and the standalone bowtie antenna in free space, respectively.

6 KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE 2625 Fig. 11. Comparison of the impedance matching behaviors for some tags: the proposed tag antenna embedded in the cavity with the AMC substrate; the same antenna used in without the AMC substrate; the same bowtie antenna used in in free space, where. Fig. 12. Radiation pattern (realized gain) of the proposed RFID bowtie-tag antenna embedded in the large metallic body at MHz. Fig. 13. Comparison of measured sensitivity and reading distances for the proposed and the ALN-9540 tag. The radiated realized gain properties of the proposed antenna on E- and H-planes at 913 MHz are plotted in Fig. 12. We intentionally, omitted the radiation pattern at 869 MHz because it is very similar to that shown in Fig. 12. The 3-dB beamwidths in E- and H-planes are 81 and 93 at 869 MHz, and 77 and 90, respectively, at 913 MHz. Fig. 13 shows the measured minimum tag sensitivity and the maximum reading distance of the proposed tag from 860 to 940 MHz. Minimum tag sensitivity, which is the same as a minimum reader transmission power enabling communication with the tag, and the maximum reading distance of the proposed tag are measured by using a commercial TESCOM TC-2600A RFID tester with ALR-9800 reader. The experiment was carried out in a fully anechoic chamber at the RFID/USN Center, Incheon, Korea. During the experiment, we set the transmission power of the reader antenna as 36 dbm. Compared to ALN-9540, we can clearly see the dual-band characteristic of the proposed tag, and the minimum tag sensitivity is about 10 db smaller at both around 869 and 913 MHz. The maximum reading distances are more than 3.1 times longer at both frequency bands. Finally, we summarize the theoretical and experimental performance of the proposed tag and the ALN-9540, which is compared in Table I. The measured maximum reading distance of the proposed tag is m at 864 MHz and m at 910 MHz, respectively, while it is 7.23 m at 869 MHz and 6.54 m at 910 MHz, respectively, for ALN Note that the frequencies chosen in Table I are those that provide maximum reading distances. Therefore, the distance increase at both bandsismorethan3.1times,which very well agrees with the theoretical reading distance derived by using (1). This is a result of trading off a 3-dB gain bandwidth because ALN-9540 is omnidirectional on the H-plane and its 3-dB gain bandwidth is broader than the proposed tag covering MHz. Consequently, we can confirm that the proposed design method is greatly successful to improve the tag performance for various platform materials. III. CONCLUSION We have proposed a long-range dual-band passive RFID tag antenna applicable for a recessed volume in metallic objects such as heavy equipment, vehicles, aircraft, and large-sized containers by using an AMC ground plane. The proposed tag antenna consists of a bowtie antenna and a recessed cavity with the AMC ground plane installed on the bottom side of the cavity. The AMC ground plane is utilized to provide dual-band operation at European ( MHz) and Korean ( MHz) passive UHF RFID bands by replacing the bottom side of the metallic cavity with a PEC-like behavior and, therefore, changing the reflection phase of the ground plane. The frequency separation ratio between the European and the Korean UHF RFID bands is very small, and the dual-band operation for this kind of very low-frequency separation ratio can be achieved by using the AMC structure. In general, RFID tags covering both UHF frequency bands (European and Korean) have good impedance matching property from 869 to 920 MHz including non-rfid frequency band in between the two RFID bands. However, our antenna shows good impedance matching behavior only in the two RFID bands. Therefore, we can say that our antenna itself is equipped with a kind of a guard-band filter, which filters out signals existing in non-rfid frequencies, and is therefore helpful to prevent possible hazardous external interference. Although the aperture size of our tag is with a cavity height of, which is larger than the commercial ALN-9540 RFID tag ( with negligible height), the main purpose of the proposed tag is on special identification of large metallic bodies such as heavy vehicles, containers, etc., on which our tag can be considered relatively small in comparison to the size of those platform bodies. Consequently, in spite of not using any kind of active power source (like batteries), we can obtain more than 3.1 times longer reading distance than

7 2626 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012 the commercial ALN-9540 RFID tag at the target RFID bands, which is fairly desirable to identify large-sized metallic bodies. This proves the validity and usefulness of our design approach from the theoretical and practical points of view. For the future work, we are focusing on a miniaturization of the proposed tag for a smaller size tag for long-distance applications. REFERENCES [1] C.Cho,H.Choo,andI.Park, DesignofplanarRFIDtagantennafor metallic objects, Electron. Lett., vol. 44, no. 3, pp , [2] S. L. Chen, A miniature RFID tag antenna design for metallic objects application, IEEE Antennas Wireless Propag. Lett., vol. 8, pp , [3] T.Deleruyelle,P.Pannier,M.Egels,andE.Bergeret, AnRFIDtag antenna tolerant to mounting on materials, IEEE Antennas Propag. Mag., vol. 52, no. 4, pp , Aug [4] C. Occhiuzzi, S. Cippitelli, and G. Marrocco, Modeling, design and experimentation of wearable RFID sensor tag, IEEE Trans. Antennas Propag., vol. 58, no. 8, pp , Aug [5] M. Lingfei and Q. Chunfang, Planar UHF RFID tag antenna with open stub feed for metallic objects, IEEE Trans. Antennas Propag., vol. 58, no. 9, pp , Sep [6]D.KimandJ.Yeo, Low-profile RFID tag antenna using compact AMC substrate for metallic objects, IEEE Antennas Wireless Propag. Lett., vol. 7, pp , [7] D.Kim,J.Yeo,andJ.I.Choi, Low-profile platform-tolerant RFID tag with artificial magnetic conductor (AMC), Microw. Opt. Technol. Lett., vol. 50, no. 9, pp , [8] M. Hirvonen, K. Jaakkola, P. Pursula, and J. Saily, Dual-band platform tolerant antennas for radio-frequency identification, IEEE Trans. Antennas Propag., vol. 54, no. 9, pp , Sep [9] J.-Y. Park and J.-M. Woo, Miniaturised dual-band S-shaped RFID tag antenna mountable on metallic surface, Electron. Lett., vol. 44, no. 23, pp , [10] J.-S. Kim, W. Choi, and G.-Y. Choi, UHF RFID tag antenna using two PIFAs embedded in metallic objects, Electron. Lett., vol. 44, no. 20, pp , [11] D. Kim and J. Yeo, A passive RFID tag antenna installed in a recessed cavity in a metallic platform, IEEE Trans. Antennas Propag.,vol.58, no. 12, pp , Dec [12] Alien Technology, Morgan Hill, CA, Product overview: ALN-9540 squiggle inlay, Aug [13] CST-GmbH, Darmstadt, Germany, CST Microwave Studio: Workflow and solver overview, CST Studio Suite 2010, [14] F. Yang and Y. R. Samii, Reflection phase characterizations of the EBG ground plane for low profile wire antenna applications, IEEE Trans. Antennas Propag., vol. 51, no. 10, pp , Oct [15] M. Hamid and R. Hamid, Equivalent circuit of dipole antennas of arbitrary length, IEEE Trans. Antennas Propag., vol. 45, no. 11, pp , Nov [16] K. Rambabu, M. Ramesh, and A. T. Kalghatgi, Broadband equivalent circuit of a dipole antenna, Inst. Elect. Eng. Proc. Antennas Propag., vol. 146, no. 6, pp , Dec Dongho Kim (M 08) received the B.S. and M.S. degrees in electronic engineering from Kyungpook National University, Daegu, Korea, in 1998 and 2000, respectively, and the Ph.D. degree in electrical and electronic engineering from Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in From 2000 to 2011, he was a Senior Researcher with the Electronics and Telecommunications Research Institute (ETRI), Daejeon, Korea, where he was involved with the development of various antennas including RFID, mobile communication and high-gain Fabry Perot resonance antennas, and artificially engineered structures such as electromagnetic band-gap (EBG) structures, frequency selective surfaces (FSSs), and artificial magnetic conductors (AMCs), etc. In 2011, he joined the Department of Electronic Engineering, Sejong University, Seoul, Korea, where he is now an Assistant Professor. His research interests include advanced electromagnetic wave theory and applications, design of highly efficient and miniaturized antennas using artificially engineered materials, design of EBG structures, FSSs, and AMCs, platform-tolerant special RFID antenna design, and development of a variety of metamaterials with negative permittivity and permeability. Prof. Kim is a life-member of the Korean Institute of Electromagnetic Engineering and Science (KIEES). Junho Yeo (S 01 M 08) received the Bachelor s and Master s degrees in electronics engineering from Kyungpook National University, Daegu, Korea, in 1992 and 1994, respectively, and the Ph.D. degree in electrical engineering from Pennsylvania State University, University Park, in During 1994 and 1999, he was a Researcher with the Republic of Korea Agency for Defense Development (ROKADD), Daejeon, Korea, where he was involved with the development of missile telemetry systems, especially the design and fabrication of lowprofile transmitting and ground-station receiving antennas. From 1999 to 2003, he was a Graduate Research Assistant with the Electromagnetic Communication Laboratory (ECL), Pennsylvania State University. From September 2003 to June 2004, he was a Postdoctoral Research Scholar in the same laboratory. In August 2004, he joined Radio Frequency Identification (RFID) technology research team, Electronics and Telecommunications Research Institute (ETRI), Daejeon, Korea, as a Senior Researcher. Since March 2007, he has been an Assistant Professor with the School of Computer and Communication Engineering, Daegu University, Gyeongsan, Korea. His research interests include computational electromagnetics, design of a class of antennas using electromagnetic band-gap (EBG) and artificial magnetic conductor (AMC) structures for RFID and mobile applications, portable wideband directive antenna design, and development of RFID sensor tags and long-range passive RFID tags. Prof. Yeo is a member of the IEEE Wave Propagation Standards Committee and a reviewer for the IEEE TRANSACTION ON ANTENNAS AND PROPAGATION, IET Microwaves, Antennas and Propagation, Progress in Electromagnetic Research, and the ETRI Journal.