Reduced Complexity MUD-MLSE Receiver for Partially-Overlapping WLAN-Like Interference

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1 Author manuscrit, ublished in "IEEE VTC Sring 2007 (2007)" Reduced Comlexity MUD-MLSE Receiver for Partially-Overlaing WLAN-Like Interference P. Mary 1,2,J.M.Gorce 2, G. Villemaud 2, M. Dohler 1, M. Arndt 1 1 France Télécom R&D, 28 Chemin du Vieux Chêne, Meylan Cedex, France 2 CITI, INSA-Lyon, bat. Léonard de Vinci, 21 Avenue Jean Caelle, Villeurbanne Cedex, France hilie1.mary@orange-ftgrou.com Abstract The roll-out density of wireless local area networks (WLANs) has recently witnessed a dramatic increase and is currently reaching saturation levels. The frequency bands designated to WLANs do thus not suffice anymore to rovide nonoverlaing, and hence interference-free, communication bands. A large body of research has been dedicated to a wide variety of otimum maximum likelihood sequence estimation (MLSE) and sub-otimum in-band interference mitigation techniques. Our contribution lies in a reduction of the state-sace of a MLSE detector in the case of a desired WLAN receiver exeriencing delayed interference from some other transmitters oerating in artially overlaing sectral bands and over indeendent frequency-selecting block-fading channels. Based on the formulation of the otimum receiver, we derive a sub-otimum receiver of reduced comlexity and demonstrate its satisfactory erformance in the context of strong interference. Index Terms Interference Cancellation, Wireless Local Area Networks, Multi-User Detector, MLSE, Partial Sectral Overla I. INTRODUCTION Wireless local area networks (WLANs), oerating at around 2.4 GHz, have been widely deloyed during the last decade. For instance, France Telecom alone has deloyed more than two Million Livebox WLAN systems for domestic and industrial use in the last decade in France. Victim of its success, the IEEE WLAN standard suffers now a lack of radio resources. Although internationally u to 14 mutually overlaing 22 MHz channels are available for IEEE b WLANs occuying a total bandwidth of u to 83.5 MHz at around 2.4 GHz, roll-out Engineers nowadays confine themselves to using a few channels only, e.g. the channels numbered 1, 5, 9 and 13 in Euroe, so as to limit interference between adjacent cells. Since four non-overlaing channels clearly do not suffice for the envisaged WLAN deloyment scenarios, other means have to be sought to mitigate the limited WLAN system caacity. In this context, two cases of interference can clearly be distinguished [1]: First, if an aggressive roll-out attern of four with strong co-channel interference (CCI) is being deloyed, then (second tier) interfering frequency bands fully overla with the frequency band of interest. Second, if a less stringent roll-out attern is being ertained, then (already first tier) interfering frequency bands artially overla with the band of interest. This late case is referred to as artial channel interference (PCI). The imortance of interference mitigation has been well recognized by the research community and a lethora of milestone contributions has emerged in recent years. Most notably, Sergio Verdú ioneered interference cancelation (IC) methods based on maximum likelihood multi-user detection (MUD) techniques for CDMA-based systems in the early 80s [2], which exhibited rohibitive comlexity. Thereuon, a large body of work concentrated on reducing this comlexity [3] [5] and most notably [6], [7]. Resultant and alternative techniques have then mainly been alied to interference mitigation in the context of cellular systems but also WLANs, some imortant of which are briefly touched uon below. In many cases, CCI is considered as the main cause of interference [10] [13]. In [7], the authors suggest to reduce the memory of the CDMA channel in an iterative multiuser detector. However, there are only a few interfering scenarios considered and the reduced state method is not clearly indicated. Although most of the works are related to cellular alications, a linear minimum mean square error (LMMSE) detector is roosed in [9] to deal with the time-disersive channel and CCI occurring in the ISM band. The roosed detector has been shown to outerform more traditional detectors, e.g. a Rake receiver. CCI rejection is also accomlished by means of blind and semi-blind IC methods [10], [11], when little or no a riori information is available on individual interferers. Multile Antenna Interference Cancelation (MAIC) techniques have been introduced to avoid/mitigate strong CCI from a given satial direction [12] [14]. Such an aroach, however, has its limitations in highly cluttered indoor environments where interference generated by one source often arrives from multile directions. Furthermore, sace-time MUD methods are investigated for the next generation MIMO OFDM-based WLAN systems, i.e. IEEE n; see, e.g., [15] and [16]. In comarison, less works have been devoted to the case of ACI or PCI mitigation. As already mentioned, PCI lays a crucial role in the limited caacity of WLAN networks. PCI cancelation would allow to increase the reuse factor, thus increasing the global throughut. A maximum-likelihood sequence estimation (MLSE) based on the Ungerboeck formulation has been extended for ACI mitigation in [17] for GSM-tye systems. They show that, even for high levels of ACI, significant erformance imrovements can be achieved with a highly comlex MLSE detector, as well as a successive interference canceler of reduced comlexity /$ IEEE 1876

2 Following the develoment roosed in [17], this aer assesses the efficiency of PCI mitigation in strongly overlaing channels over severely frequency-selective multiath channels. Due to the differing channel imulse resonses (CIRs) for each received signal, we focus on the general case of interfering signals having an asymmetrical cross-correlation (ACC) function. The aer is structured as follows. The system model with an aroriate MLSE of [17] is described in Section II, where we have augmented the branch metrics to facilitate causal imlementation. In Section III, we introduce and discuss the channel synchronization roviding a MLSE of reduced comlexity. In Section IV, we resent and discuss simulation results for our MUD-MLSE alied to WLAN-like systems. Finally, in Section V, conclusions are drawn. II. STANDARD MUD-MLSE RECEPTION For the sake of comleteness, we outline in this section the otimal synchronous MUD-MLSE receiver as artially derived in [2], [17]. We commence with the system model, followed by the augmented derivation of the MUD-MLSE algorithm, which facilitates causal imlementation. A. System model We will deal with binary hase shift keying (BPSK) modulation schemes only; an extension to higher order modulations as well as differential formulations is achieved using a similar aroach. The BPSK symbol sequence of the k th user, denoted by {d k (n),n=1,...,n} with N being the sequencelength, is shaed by a root raised cosine (RRC) filter with given roll-off factor and u-converted to the k th frequency band. In time-continuous formulation, the transmitted signal is hence exressed as: x k (t) = n d k (n)h f (t nt s ), (1) where T s is the symbol duration and h f (t) is the time-resonse of the RRC filter. We assume that there is more than one oerational transmitter, where the bands of user k and l are searated by a sectral distance f kl. Note that if there is an equal bandsacing, then f kl = f (l k), where f is the minimum band-sacing. The signal sequence of the active user roagates through a multiath roagation channel with channel imulse resonse h k (t). At the receiver, additive white Gaussian noise (AWGN), n(t), with a one-sided ower sectral density N 0 is added to the received signal, after which it is amlified, samled, er user RRC matched filtered, and rocessed. Assuming user i to be the baseband user of interest, the total baseband signal can be exressed as: y(t) = d i (n)g i (t nt s )+ (2) n e j2π fikt d k (n)g k (t nt s )+n(t), k k =i n where g k (t) =h f (t) h k (t) and reresents the convolution oerator. The otimum detection of the desired user in the resence of PCI is achieved by means of a MLSE. B. MUD-MLSE and Viterbi Branch-Metric Maximizing the log-likelihood function of the received data is achieved by maximizing a sum of branch metrics [17]: J H = M k (n), (3) k n where M k (n) is given in (4) on to of the next age. In there, Re{ } denotes the real art and * the comlex conjugate. Furthermore, ψ k (n) = y(t)e j2π fikt gk(t nt s )dt, (5) and the generalised correlation term [17] is given as: s kl () = g l (v)e j2π fklv gk(v t)dv t=ts. (6) The cross-sequence interference is given by the last term in (4). Note, however, that the metric formulation (4) cannot be used directly in a Viterbi algorithm, because the cross-sequence terms are not causal; indeed, the summation with resect to is done for all. The causal formulation may be easily obtained by moving the non-causal terms in M k (n), involving d l (n ) with <0, intom l (n ) leading to a modified branch metric given in (7) on to of the next age and referred to as Mk c (n). This detail is rovided because it will be used in the next section for ACC interference rejection. III. REDUCED STATE-SPACE MUD-MLSE RECEPTION A. Viterbi metric derivation for reduced comlexity The above derived MUD-MLSE is otimum in the minimum error robability sense and hence may exhibit a very large comlexity in frequency-selective fading channels requiring a high number of states in the Viterbi decoder. Since the Magnitude of Auto and Cross Correlation Function s kk s ll s kl s lk Symbol Number Fig. 1. Generalized auto- and cross-correlation terms of bands k and l with instantaneous channel realizations given in eq. (8). 1877

3 M k (n) =Re { d k(n) [ 2ψ k (n) d k (n)s kk (0) 2 >0 d k (n )s kk () l l =k ]} d l (n ) e j2π f kl(n )T s s kl () (4) { [ Mk(n) c =Re d k(n) 2ψ k (n) l d l (n) e j2π f klnt s s kl (0) 2 l ]} d l (n ) e j2π f kl(n )T s s kl () >0 (7) signal of interest and the interfering signals do not undergo the same wideband fading channel, the instantaneous realizations of the CIR may introduce a significant asymmetry in the crosscorrelation function s kl (). As an illustrative examle, let us take the instantaneous amlitudes of two symbol-saced channels: h k = [ ] h l = [ ] Clearly, in this examle, the maximum ower tas are not ositioned at the same time. Consequently, the n th symbol of the artially overlaing band is not the main interfering symbol at time n. In Figure 1, the absolute values of the generalized correlation terms s kk,s ll,s kl and s lk given in (6) are deicted. The maximum energy of the cross-correlation functions are shown to be shifted in oosite directions. The joint MLSE detector based on (7) takes into account all non zero coefficients of auto and cross correlation function from the 0 th delay. This MLSE receiver exhibits a high comlexity, the associated state-sace increasing exonentially with the time-lag. To decrease the comlexity, we roose to limit the state-sace of the decoder whilst otimizing the selection of the strongest cross-correlation terms. For instance, with reference to Figure 1, a full-comlexity MUD-MLSE detector ought to have a state-sace caturing all auto-interfering aths and also all cross-interfering aths, i.e. L =5. A sub-otimal MUD-MLSE would aim at taking into account only the most significant terms. Thus, L =4aears to be a good trade-off if only auto-interference is considered, but fails to remove significant cross-interference. We hence roose to modify the branch metric (7) so as to synchronize the estimated symbols of each signal in a Viterbi imlementation such that the maximum of the intercorrelation terms shifts towards zero. This can be achieved starting from (4) by considering symbol n + ν k of interfering channel k simultaneously with symbol n in the user channel i with ν k Z in the metric decision. The branch metric is obtained under a causality constraint with resect to this new synchronization, leading to the new branch metric given by (9). This formulation facilitates erformance to be traded against comlexity. B. Finding the otimal shift In a subotimal situation, i.e not all cross-correlation coefficients are taken into account, the estimation error is minimised if the shift ν k is chosen such as to retain the maximum (8) energy of cross-correlation terms. The otimal shift is found by maximizing the energy of cross-correlation terms according to: ν k = arg max ν k Z (L 1) = (L 1) s ik ( + ν k ), (10) with ν k Z being the considered shift. Related to this, in [6], the authors suggest an algorithm to reduce the number of states of the trellis diagram based on the knowledge of the CIR at the receiver. The algorithm quantifies the differences between the robability density functions of the correct and incorrect branch metrics in the trellis and allows to reduce significantly the comutational comlexity. However, this aroach is largely more comlicated than the herein roosed one, which is simly based on the energy of the CIRs. Furthermore, in [7], the authors suggest to reduce the memory of the CDMA channel in an iterative multiuser detector. The current aer differs from [7], because they reduce the number of states of the Viterbi without considering interference. IV. PERFORMANCE RESULTS &DISCUSSION For all simulations, we have used two different ower delay rofiles. The first channel is a 5-ta symbol-saced CIR with average owers [0, -2, -5, -10, -15]dB and simulates a line of sight (LOS) communication between the mobile station (MS) and the access oint (AP). A second communication is started between another MS and the same AP on the next channel number. From the receiver oint of view, this communication is characterized by a non line of sight (NLOS) roagation with the ower delay rofile [ ]dB. Both are blockfading channels, having a total average ower normalized to unity. The RRC roll-off factor is fixed at α =0.33 and the acket size to 1000 symbols. The symbol duration is assumed to be 90ns as a reference to the chi duration in WLAN. Figure 2 shows the average bit error rate () versus Eb/No obtained with different state-saces, labelled L, and with/without the otimal shift. In this simulation, the signal to interference ratio (SIR) is fixed to 0dB and the sectral overla to about 65% ( f T s /(1 + α) =0.35). In all grahs, the curve related to MLSE with PCI, is the erformance of a receiver which does not take into account the interfering adjacent channel but only its own inter-symbol interference (ISI). This curve shows that without interference rejection, the system cannot work. The adative shift allows to imrove the results for any sub-otimal value of L (L < 5). For instance, 1878

4 M c k(n) = Re { d k(n + ν k ) [ 2ψ k (n + ν k ) d l (n + ν l ) e j2π f kl(n+ν l )T s s kl (ν k ν l ) (9) l 2 ]} d l (n + ν l ) e j2π f kl(n+ν l )T s s kl ( + ν k ν l ) l > Emirical CDF F() Eb/No (db) Fig. 2. Average versus Eb/No, labelled on Viterbi state-sace L, with and without comlexity reduction; channel overla = 65%, interference ower = signal ower. for Eb/No =15dB and L =4, the robability of error is reduced by a factor of 10 thanks to the shift. Figure 3 shows the cumulative density function (CDF) of the at Eb/No =15dB. Again, for each L, the erformance with and without otimal shift is rovided. The CDF is of interest in the case of block-fading channels, as often occurring in WLAN settings. In such channels, we are not only interested in the average data throughut, but also in the likelihood of having a given error rate at a given Eb/No. For the reduced state-sace of L = 3, i.e. only 16 Viterbi states, yields a sufficient outage robability at low s. Indeed, in this case, the robability to yield a of is 68% with the otimal shift while only 40% for the standard aroach with L =3. If we consider L =4, the robability to have a less than is more than 95% with the shifted metric and again less than 85% with the standard one. When the Viterbi state-sace (L) is sufficient, the erformance is otimum in the error rate sense. Figure 4 shows the average versus the carrier sacing. The SIR equals 0dB and Eb/No =15dB. Here, increasing the sectral overla degrades the erformance. But with a carrier sacing of 3MHz only (80% channel overla since the bandwidth is aroximatively 15MHz) a large gain is achieved. Indeed, for L =4the obtained with the metric in (9) is by an order of magnitude less: 5 against 4 without metric adjustment. We also observe that the of a subotimal Viterbi state-sace with and without metric adjustment, i.e. L =3and L =4, converge when the Fig. 3. CDF of at Eb/No = 15dB, labelled on Viterbi state-sace L, with and without comlexity reduction; channel overla = 65%, interference ower = signal ower. sacing between bands increases. This is due to less interfering signal when the channel overla decreases, so both metrics are equivalent. We also include the single-user MLSE erformance curve. This curve converges towards the erformance curve of the MUD-MLSE receiver with L =5without floor. The erformance enhancement of a receiver based on (9) is ointed out in strong interference conditions (the left art of the grah). Finally, Fig. 5 shows the average bit error rate versus the average interfering ower, with a sectral overla of 65% and Eb/No =15dB. We observe a erformance enhancement for L =4with the shifted metric. A reduction of about 10 is achieved when the interfering signal is 5dB weaker than the desired signal. When the SIR increases, the augmented metric and the classical one converge. The best erformance imrovement is achieved at low SIR. For a SIR = -5dB and L = 4 the is about 4 for the metric in (9) against only without shift. Moreover, when the signal to interference ratio increases, the MUD-MLSE receiver with L =5converges towards the single-user detector without floor. V. CONCLUSIONS The aim of this aer has been to demonstrate the erformance of a reduced-comlexity MLSE detector in the context of heterogeneous WLAN-like channels. The asynchronous behavior is due to the indeendent instantaneous channel imulse resonse realizations of the desired and interfering users, 1879

5 Carrier sacing (MHz) SIR (db) Fig. 4. Average versus channel overla, labelled on Viterbi statesace L, with and without comlexity reduction; Eb/No = 15dB, interference ower = signal ower. where the strongest aths of the interferer is time-shifted comared to the strongest ath of the desired user. The interference cross-correlation terms hence yield their maximum at a nonzero time-lag. Based on this observation, we have roosed to shift the MLSE branch metrics so as to comensate for this asynchronism. This facilitates the construction of MLSE ath metrics of reduced state-sace, thereby significantly reducing the comlexity of the detector. The simulation results oint out how the shifted metric enhances the efficiency of a subotimal MUD-MLSE detector. Note that the high observed herein does not reflect the of a b based WLAN system because neither DSSS neither CCK were considered. As future work, this technique will be introduced in a comlete b receiver. We exect, however, that the qualitative behavior will be the same as the one exosed herein. The roosed MUD-MLSE detector of reduced comlexity can be alied to other systems, which suffer a artial sectral overla and asynchronous interference due to a strong frequency selective channel. We exect that with the marriage of emerging cognitive radios [18] and forthcoming sectrum liberalization [19], interference cancelation methods will lay a vital role for future cellular, WLAN and ad hoc systems. REFERENCES [1] R. Diestel, Grah Theory, Sringer-Verlag, [2] S. Verdu, Otimum Multiuser Signal Detection, Ph.D diss., Univ. of IL., Urbana-Chamaign,Aug [3] S.Moshavi, Multi-user detection for DS-CDMA communications, IEEE Commun. Mag., vol. 34, , Oct [4] Laster J.D, Reed J.H, Interference Rejection in Digital Wireless Communications, IEEE Signal Processing Magazine, vol. 14, Issue: 3, May [5] Andrews J.G, Interference Cancellation for Cellular Systems: A Contemorary Overview, IEEE Wireless Communications Magazine, vol. 12, Issue: 2, Aril [6] H. Zamiri-Jafarian, S. Pasuathy, Comlexity Reduction of the MLSD/MLSDE Receiver Using the Adative State Allocation Algorithm, IEEE Transactions on Wireless Communications, vol. 1, no 1, January Fig. 5. Average versus interfering ower, labelled on Viterbi state-sace L, with and without comlexity reduction; Eb/No = 15dB, channel overla = 65%. [7] Z. Qin, et al., Iterative Reduced-State Multiuser Detection for Asynchronous Coded CDMA, IEEE Transactions on Communications, vol. 50, no: 12, December [8] P.B. Raajic, B. Vucetic, Adative receiver structures for asynchronous CDMA systems, IEEE J. Select. Areas Commun., vol. 12, no. 4, , May [9] I. Oermann, Extending the scoe of WLAN through LMMSE CDMA receiver structures, IEEE PIMRC 2002, Volume 2, Set. 2002, [10] A.M. Kuzminskiy, Y.I. Abramovich, Adative second-order asynchronous CCI cancellation: Maximum likelihood benchmark for regularized semi-blind technique, in Proc. ICASSP, vol. IV, , May [11] A.M. Kuzminskiy, C.B. Paadias, Re-configurable semi-blind cancellation of asynchronous interference with an antenna array, in Proc. ICASSP, vol. IV, , Ar [12] L.J. Pesik, M.A. Beach, D.P. McNamara, P.N. Fletcher, Performance analysis of smart antenna systems for indoor wireless LANs, Third International Conference on 3G Mobile Communication Technologies 2002, Conf. Publ. No. 489, 8-10 May 2002, [13] T.W. Nuteson, G.S. Mitchell, J.S. Clark, D.S. Haque, Smart antenna systems for wireless alications, IEEE International Symosium on Antennas and Proagation 2004, Volume 3, June 2004, [14] M. Ahn, D. Kim, J.S. Kenney, Throughut imrovement in interference limited multiath environments using a smart antenna for IEEE b WLAN, IEEE Radio and Wireless Conference 2004, Set. 2004, [15] S. Suthaharan, A. Nallanathan, B. Kannan, Joint interference cancellation and decoding scheme for next generation wireless LAN systems, IEEE SPAWC 2003, June 2003, [16] B. Xu, C. Yang, S. Mao, Multiuser sace-time code for OFDM/SDMA systems [WLAN alications], VTC 2004-Sring, Volume 2, May 2004, [17] H. Arslan, S. C. Guta, G. E. Bottomley, S. Chennakeshu, New Aroaches to Adjacent Channel Interference Suression in FDMA/TDMA Mobile Radio Systems, IEEE Transactions on Vehicular Technology, Vol. 49, No. 4, July 2000, [18] Joseh Mitola III, Cognitive Radio: An Integrated Agent Architecture for Software Defined Radio, Ph.D diss., Royal Institute of Technology (KTH) Stockholm, Sweden, 8 May, [19] M. Cave, Indeendent Audit of Sectrum Holdings - Cave Reort, Her Majestys Treasury, Ofcom, Dec